Wideband Digital Receivers for
`Multi-Standard Software Radios
`
`Reza Karimi’, Bernd Friedrichs”
`‘Motorola GSM Products Division, Swindon, UK
`*Bosch Telecom, Backnang, Germany
`e-mail: karimir@ecid.cig-mot.com, friedr@bk.bosch.de
`
`Abstract: Multi-standard software-definable radios which are capable of operation according to a
`variety of different mobile radio standards represent an extremely powerful tool for evolution towards
`future third-generation cellular systems. This is particularly the case in Europe where the emergence of
`advanced UMTS air-interfaces needs to be accompanied with some degree of backward compatibility
`with the well-established GSM/DCSsystems. This paper examines a number of the architectural issues
`and trade-offs involved in the design ofwideband multi-standard GSM/UMTSdigital radios and presents
`an examination ofthe filtering and ADC technology requirements for their implementation.
`
`This work has been undertaken in the context of the FIRST project (Flexible Integrated Radio System
`Technology) as part ofthe ACTS mobile line.
`
`1. Introduction
`
`The ability to process signals corresponding to a wide range of frequency bands and channel
`bandwidthsis a critical feature of 3rd generation cellular multi-standard radios and impacts heavily
`on the design of both analogue and digital segments of the radio. In Europe, cellular frequency bands
`are concentrated in the neighbourhoodsof 1 and 2 GHz, while channel bandwidths can vary from 200
`kHz (GSM/DCS)to around 1.6 MHz (DECT, FRAMES mode-! UMTSproposal) and even upto 6.4
`MHz and beyond (FRAMESmode-2 and other wideband UMTSproposals) [1]. A primitive approach
`for the implementation of a multi-standard radio is characterized by the use of distinct transceiver
`chains, each optimized to support one of the above radio standards. However, such a “stacked-radio”
`or “velcro” approach is inflexible and increasingly infeasible for the support of more than two
`standards. It should however be noted that depending on the frequency coverage of the radio, some
`degree of duplication in the RF components(e.g. preselect filter, PA, LNA...) may be inevitable.
`
`A more advanced approach involves the use of a single wideband transceiver whichis sufficiently
`flexible for the support of multiple standards [2]. One possible option involves the use of
`programmable analogue selectivity whereby the bandwidth of the analogue segment of the
`transceiver
`is adapted to accommodate a
`single channel of the target air-interface. The
`programmability of the analogue filtering may be at a coarse level, followed by fine digital channel-
`selection filtering. In this paper, however, we focus on the more challenging option of using fixed
`analogue selectivity whereby the fixed bandwidth of the analogue front-end equals the width of the
`widest channel of interest. This latter option is also of interest in the context of advanced base
`stations capable of digitizing entire operator frequency bands. The implications of this approach with
`respect to current and emerging technologies in the context of wideband multi-standard GSM/UMTS
`receivers are investigated.
`
`2. Characteristics of Wideband Receivers
`
`An important feature of a wideband muli-standard radio is that the passband B, of the analoguefront-
`end needs to be sufficiently large to accommodate the air-interface with the widest channel
`bandwidth. This implies that, unlike traditional narrowbanddesigns, the latter stages of the receiver
`and the ADC can be potentially exposed to a large numberof carriers when processing signals from
`
`1
`
`SAMSUNG 1026
`
`1
`
`SAMSUNG 1026
`
`

`

`narrowband standards. The situation may be readily quantified for the case of a GSM/UMTS
`wideband receiver. The large channel bandwidths of the proposed UMTSair-interfaces (>1.6 MHz)
`imply that the dynamic range ofthe receiver needs to cope with the multiple GSM carriers as well as
`possible blocking signals (caused by sources external to the network) which can be present within the
`bandwidth B, . The powerlevels of these blockers are detailed in the GSM specifications 5.05 [3] and
`are summarized in Table (1). The figures are relative to a wanted carrier at +3dB above the receiver
`sensitivity level (-100 dBm for DCS mobiles and —104 dBm for other radio types).
`
`Blocker Offset df
`MHz
`0.6 - 0.8
`0.8-1.6
`1.6 - 3.0
`
`DCS-MS
`[dBc]
`
`DCS-BTS
`[dBc]
`
`GSM-MS
`[dBo]
`
`GSM-BTS
`[dBc]
`
`
`
`Table (1) - GSM/DCSblocker specifications.
`
`The abovespecifications are used in the following sections in order to evaluate the filterimg and ADC
`requirements of wideband GSM/UMTSreceivers.
`
`2.1. AnalogueFiltering
`
`The objective offiltering in the analogue domainis not only to isolate the channelofinterest but also
`to suppress adjacent channels which mayalias as co-channel interferers duc to the ADC sampling
`action. Thesituation is depicted in Figure (1) for a GSM receiver with an analogue passband of B, =
`1.6 Mhz.
`
`Spectrum
`prior to ADC
`
`Rejected by
`Rejected by
`digitalfiltering
`analogfilters
`> oe
`
`oF
`_
`
`71...88 dB
`54...85 dB
`
`Ps
`N
`
`Blocking ~
`signals
`
`a
`
`@ Baseband Sampling
`ret
`Po bg
`e@ Passband Subsampling
`Fo=IF =F /2
`Fg = ADC sampling rate
`IF = Intermediate frequency
`
`Pg= Powerof blocker
`
`1 co
`Anti-aljas P \
`filter
`x
`
`
`
`
`
`
`
`Py = Powerof wanted signal
`
`
`
`
`
`_
`Stopband
`NN
`attenuation A
`Wanted
`
`widebanda~
`a
`signal
`
`
`NL
`
`Figure (1): Rejection of blocking signals
`
`The diagram applies irrespective of whether the receiver architecture deploys baseband or passband
`digitization (sce Section 3). While blocking signals which appcar outside the analoguc pass/transition
`bands must be rejected by analogue filtering, those appearing within the analogue passband can be
`rejected by digital filtering following analogue-to-digital conversion. The extent of stopband
`attenuation is determined by the amountof tolerable co-channel interference caused by the aliasing of
`large blockers. Co-channel interference poweris typically specified to be at a level of about 10 to 20
`dB below that of the wanted signal depending on the modulation scheme (SNReo.cxanver, > 9 dB for
`GSM). Consequently, depending on the radio type and the sampling rate uscd, the required stopband
`attenuation is given by:
`
`A = Paap — Pra + SNRco-CHANNEL-dB = [54...88] + [10...20] = 64...108 dB
`
`qd)
`
`The above values represent the combined effect of all analoguefilters. The large values of stopband
`attenuation, in conjunction with wide passbands, demand very steep transitions in the frequency
`response, therefore requiring additional stages of filtering compared with narrowband designs. Apart
`
`2
`
`

`

`from size and cost issues, an increase in the numberof stages also contributes to the receiver noise
`figure and non-linearities. Filter complexity may be traded off against power consumption and DSP
`load (for digital decimation and channelselection) via an increase in the ADC samplingrate F,.
`
`2.2. Analogue to Digital Conversion
`
`Asin the case of analoguefiltering, the ADC requirements for a GSM/UMTSmulti-standard radio
`are heavily influenced by the GSM blocker specifications. Two scenarios considered for the
`evaluation of ADC parameters are depicted in Figure (2) below:
`Blocker
`Pp
`CW carrier
`
`Pe
`Pp
`
`
`Wanted
`
`
`GSM: -43 dBm
`
`Px
`DCs: 49 dBm
`Sensitivity
`
`+3dB
`
`Noise PSD
`
`
`
`1+__orntt——___——
`
`fF —> , <
`f
`0.8 MHz
`0.8 MHz
`f
`ce
`B, = 200 kHz
`\
`cen ;
`Quantization Noise
`SINNRaF = 20dB
`
`Scenario #1
`
`Moculated
`
`cw
`.
`Sinewave
`
`Scenario #2
`
`Wanted
`Signal
`
`Px
`Sensitivity
`
`GSM/ DCSSpecification:
`CI=94B
`
`GSM/ DCSSpecification
`C/ = 9dB
`
`Pima
`3rd-Order Intermod Component
`C/1M3 = 200B
`
`Figure (2): Scenarios for evaluation of ADC requirements.
`
`In the first scenario, a large CW carrier at an offset frequency of of (see Table 1) is considered to
`cause blocking of the wanted carrier (at 3dB above sensitivity) [3]. Since df S$ B, for a wideband
`receiver, the ADC is exposed to the full amplitude of the blocker, 1.e. the dynamic range of the ADC
`needs to simultancously accommodate the blocker (as well as other adjacent-channel carricrs) and
`provide adequate signal-to-noise ratio within the bandwidth of the wanted carrier. The ADC full-scale
`range X,, needs to be sufficiently high to prevent clipping when the signals add in phase and may be
`written as:
`
`(2)
`Xp = {2Pp
`where the sinusoidal blocker is assumed dominant and no allowance is made for headroom. The
`
`required numberof bits 6 (or quantization step-size A) can then be found for a specific value of
`quantization SVRo- (followingdigital filtering) where:
`
`2xB
`22
`O#
`
`AN
`=——-x
`12°
`
`Pop
`
`Or
`
`3a
`
`Ga)
`
`SNRop
`
`OF
`
`
`P
`=—- = 3x 27?
`Por
`
`o»
`
`PE
`x = x
`XxX?
`
`
`FE.
`5
`2xB. GP)
`
`3b
`
`The last ratio in Equation (3b) represents the processing gain which results from digital filtering
`under the assumption of a white quantization noise spectrum [4]. To ensure that the quantization
`noise poweris negligible compared to that of interferers and other sources of thermal and device
`noise, a value of SVRox= 20 dB may be assumed. The spurious-free dynamic range SFDRis defined
`as the ratio of X,,
`to the rms amplitude of the largest spurious component over the entire Nyquist
`band. SI/’DR represents all sources of noise and distortion (including integral and differential non-
`linearities, sampling jitter etc.) caused by the ADC. Assuming that the largest spurious component
`falls within the bandwidth of the wanted carrier and assuming a total SNR of 20 dB, the SkFDR may
`be computed as:
`
`SF,DR;=20lo,
`
` xX
`2B.
`
`= | +20
`
`dBFS
`
`(relativetofull-scale)
`
`(4)
`
`In the second scenario, two large carriers at an offset of 800 kHz are considered to cause intermod
`products which fall within the bandwidth of the wantedcarrier [3]. If B, is sufficiently large to pass
`the two large carriers, this scenario dictates the two-tone linearity requirements of the ADC. The two-
`
`3
`
`

`

`tone SFDRis defined here as the ratio of the amplitude sum of the two large carriers (nominally equal
`to full scale X,, ) over that of the third-order intermod component. Again assuming a 20 dB margin:
`
`SEDR, =20og, +20
`
`2x.,/2P,
`2P.
`
`dBFS
`
`(relativetofull-scale)
`
`(5)
`
`The ADC parameter values required for receivers intended for two different UMTS channel
`bandwidths of 1.6 MHz and 6.4 MHzare presented in Tables (2a) and (2b) where 5, = 200 kHz.
`
`POCSMOBILETT
`pOcsBTsS18
`PGSMMOBILEJ12-713
`GSM BTS
`15 - 16 p05
`e Analogue channel bandwidth B, = 1.6 MHz as specified in the FRAMES mode-1 proposal.
`
`e Sampling Frequency F, = (2x2B,)= 6.4 MHz .
`
`The ADC dynamic rangeis dictated by the GSM blocker at an offset of 0.8 MHz from the wanted
`carrier. Note that SFDR; figures are not quoted since one of the intermodtest carriers (1.6 MHz away
`from the wantedcarrier) falls outside the analogue passband.
`
`pOcsBTS218
`Pp|988
`GSM BTS
`14-15 P08|
`e Analogue channel bandwidth B, = 6.4 MHz as specified in the FRAMES mode-2 proposal.
`
`e Sampling FrequencyF, = (2x2B,) = 25.6 MHz .
`
`The ADC dynamic range is now dictated by the blocker at an offset of 3 MHz from the wanted
`carrier. Note that despite an increase in the blocker level, the quantization resolution requirements
`have fallen compared to Table (2a). This is due to the increased processing gain resulting from the
`higher sampling rate (see Equation 3). The SFDR figures are not subject to any processing gain and
`have accordingly increased. The current state of the art in commercially available ADCs (6=11,
`SFDR=80 dB and F,=40 MHz) falls significantly short of the values presented above for GSM base
`stations but approaches those required for DCS radios. It should be emphasized that the analysis
`presented here involves worst-case test scenarios. It has been argued, based on experience in
`commercial systems that a moderate relaxation in the stringent GSM blocking specifications may be
`feasible without a significant impact on system performance [5]. Furthermore, measures such as
`frequency hopping, adaptive beamforming, and improved detection algorithms may somewhat
`alleviate the demands on the ADC.
`
`3. Survey of Receiver Architectures with Fixed Analogue Bandwidth
`
`the merits of a number of receiver
`Having considered the filtering and ADC requirements,
`architectures for wideband multi-standard radios will be examined in this section. Figure (3)
`illustrates four different architectures, showing the path from antennato the digital demodulation unit
`(includes decimation, low-pass channel selection filtering and complex phase rotation for carrier
`synchronisation). Several methods for sampling at IF followed by digital down-conversion are listed
`as types A, B, C. The architecture of type D employs analogue quadrature down-conversion and
`sampling at baseband, usually known as direct down-conversion. For a multi-standard terminal
`capable of processing standards like DCS or UMTSland mobile segments 3/6, the entire frequency
`bandpresentat the preselect filter is about 500 MHz wide spanning from 1710 to 2170 MHz.
`
`4
`
`

`

`DLPF
`Preselect 1st Mixer
`BP
`2nd Mixer AA
`Decimation
`
`
`r=H(2)|INR
`
`
`.
`—
`4
`
`
`
`
`FI=Rx+IF1=Bg F2=IF1+IF2 Bg Fs 3 IF2 Digital Quad Digital Carrier
`
`Downconversion
`Sync
`A) DoubleIF with digital subsampling
`DLPF
`Decimation
`AA
`Preselect
`Mixer
`
`
`
`LQ»
`x
`=| H(z)
`
`yLAs
`
`
`FI=RxtlF1
`By
`FS=ZIF1
`Digital Quad
`Digital Carrier
`yee
`wy
`.
`:
`Downconversion
`Sync
`B) Single IF with digital subsampling
`
`DLPF
`Decimation
`
`
`=fH]INR
`
`Digital Quad
`Digital Carrier
`Downconversion
`Syne
`
`Preselect
`
`
`Mixer
`
`AA
`
`FI=Rx+IF1
`
`Bg
`
`Fs<<IF1
`
`Preselect Mixer
`
`FI=Rx+IF1
`
`IF1
`Ba
`
`
`DLPF
`
`Decimation
`
`
`> (5a) H(z) LN Q=>
`yLlSs
`
`Digital Quad
`Digital Carrier
`
`
`Downconversion
`sync
`D) Sampling at baseband (direct down conversion / single IT / double IT)
`
`
`
`Figure (3): Alternative analogue front-end architectures
`
`Architecture A (double IF with digital subsampling). This approach was described in [2]. See also
`Figure (4) for a spectral representation of the signals in the receiver chain. Sampling is performed at
`an appropriate //’, and the down-conversion processis realized completely via DSP. Subsampling in
`the second Nyquist zone given by therelation F,=(4/3)x/F, implies a sampling rate F, which is less
`than twice the highest frequency component in the sampled signal. The presclect filter rejects the
`image frequencies of the first mixer. The transition bandwidth ofthe preselect filler can be made
`broad if JI’; is chosen to be sufficiently large (i.e. at least a couple of hundred MHz). The bandpass
`(BP) and anti-alias (AA) filters at J7"; and IJ’, respectively are designed for bandwidth B, (eg. By =
`1.6 MHz). The BP filter rejects the image frequencies of the second mixer and requiresa transition
`bandwidth of (2x/F2) — Bg = (3/2)xF, — Bg. The AA filter suppresses the components which can be
`aliased due to sampling and requires a transition bandwidth of F,/2 — Bz. A high F, relaxes the
`required steepness of the analogue BP and AA [filter frequency responses, bul places additional
`demands on the ADC and the subsequent DSP. This important trade-off between the analogue and
`digital domains needs to be carefully considered, particularly with respect to the expected radio
`channel conditions and the typical levels of interferers and blockers. The choice of /F’ is more or less
`independent of LF, and can be optimized with respect to implementation of the BP filter. A drawback
`of architecture A, is that two analogue filters with demanding requirements (see Section 2) and two
`analogue mixers contributing to intermodulation distortions are required. Furthermore, F, and JF, are
`related according to the subsampling equation. Finally, apart from the subsampling, the architecture
`is very similar to that of conventional receivers.
`
`Architecture B (single LF with digital subsampling). This is similar to architecture A with the
`difference that only a single intermediate frequency JF; is employed. This allows a reduction in the
`analogue component count as well as a reduction in nonlinear distortions. However, according to the
`subsampling equation F,, = (4/3)x/F, from architecture A, /F; is a direct function of F, and thusit is
`difficult (in comparision to architecture A) to achieve a compromise between a high JJ’; (to ease
`analogue filtering, particularly image rejection) and a low JF, (to ease ADC and DSP complexity and
`improve attenuation of nearby blocking signals). However, since the preselect filter cannot achieve a
`steep frequency response, a low JF, design is not really feasible, and a high JF, will cause infeasible
`DSP requirements. Consequently, architecture B seems unrealistic.
`
`5
`
`

`

`Architecture C (single IF with extreme subsampling). The concept of extreme subsampling implies
`the use of a sampling rate F, which is considerably lower than JF. This corresponds to subsampling
`in the Mth Nyquist zone (i.e. F, = [4/(2M-1)|x/F\) where M>>1. An important implication is that
`extreme subsampling requires a fast ADC with a broad bandwidth (such ADC technology is now
`becoming available, eg. bandwidth of 450 MHz for F, =20 MHz from Analog Devices). This
`technology is in contrast to that of conventional ADCs where the bandwidth is usually limited to
`2xF,. Architecture C is similar to architecture A with respect to the two separate design tradeoffs for
`iF, and #: High JF, to widen the preselect transition bandwidth, low /F| to ease AA. High F, to
`widen AA andto achieve high stopbandattenuation, low F, to ease ADC and DSP. As an advantage,
`the analogue componentcountis reducedto that of architecture B. However, the requirements on the
`stopbandattenuation of the analoguefilter(s) are similar for architectures A,B and C.
`FL
`AntennaFilter |—— mage
`
`
`
`Spectrum beforefirst mixer
`
`0
`
`2200
`
`Rx
`
`f [MHz]
`
`1700
`-1700
`-2200
`BP Channel Filter
`Spectrum afterfirst mixer with F1 = Rx+/F7
`| Ds
`
`| _
`- A >
`F1
`OL IFA
`Me
`RxtIFt
`f
`-Rx4F1
`“BP Channel Filter“.
`Image”
`(enlarged) _.”
`
`Spectrum after second mixer with F2 = /F1+/F2
`
`
`
`IF1| F2
`|
`
`Ba
`Anti-alias Filter
`
`2IF2— Bg = (32)Fs— By
`
`-IF2
`AF1IAF2
`_
`Spectrum after ADC with Fs = (4/3) IF2
`
`0
`
`|
`
`IF2|
`By
`
`f
`IF1+IF2
`L_ (2)3)IF2 -By = Fs/2 -Bg
`
`7
`Fs
`
`-5
`Fs
`
`3
`Fs
`
`-s 0
`a
`
`Fs
`rr
`
`3
`aFs
`
`5
`aFs
`
`7
`aFs
`
`f
`
`Figure (4): Spectral representation of signals in doublc-IF recciver (type A).
`
`Architecture D (sampling at baseband). Sampling performed at baseband is an alternative to
`passband subsampling and represents a traditional approach. Sampling can be preceded by either
`direct-
`(zero IF), single-IF or double-IF downconversion stages. The main advantage of this
`architecture is that the AA filtering operation can be performed at baseband (instead of passband)
`with an analogue low-passfilter (ALPF)transition bandwidth of F, — Bg (instead of F,/2 — Bg). Filter
`requirements are thus relaxed. The analogue componentcount is reduced, particularly for zero-IF. A
`drawback is the need for an analogue quadrature down-converter which introduces the associated
`problems of gain/phase mismatch in the I and Q branches. This problem could be significant when
`dealing with multi-level modulation schemes proposed for pico-cellular modes of UMTS.
`
`the use of analogue down-conversion may be argued to be
`From a conceptual point of view,
`contradictory to the spirit of an ideal softwarc-dcfinable digital radio. However, from a practical point
`of view, architecture D can be seen to provide a number of advantages in the context of multi-
`standard radios. The most important advantage is that programmable analogue selectivity can be
`readily implemented through the use of tunable active low-pass AA filters (bandpass IF
`implementations using SAW technology is unrealistic). In addition to alleviating the dynamic range
`problemsof the ADC, this approach would allow a reduction in the sampling rate F, in accordanceto
`the channel bandwidth of the target standard, hence achieving significant savings in power
`consumption.
`
`6
`
`

`

`4. Digital Carrier Synchronisation and Related Tradeoffs
`
`For a multi-standard terminal capable of processing standards like DCS or UMTS,the entire
`frequency band can be about 500 MHz wide. According to the GSM specifications, the transmit
`frequency must be generated with an error less than 90 Hz (0.1 ppm) and this demandstheability to
`generate around five million distinct frequencics. In order to enable a simple and cost-cffective
`synthesizer design, it is possible to increase the frequency quantization step-size by several orders of
`magnitude,
`if complex phase rotation is performed at baseband in the digital segment of the
`transceiver (depicted in Figure 3). A total frequency error F, representing all imperfections due to
`control algorithms and Doppler shifts would require passband extensions of 2x|F;| for analogue AA
`filtering and 4x|F;| for digital low-passfiltering respectively. Frequency errors of the order of some
`percentage of B, are not critical and thus the quantization step-size of the local oscillator can be
`increased to 10...100 kHz, thereby allowing considerable simplifications compared to the case of
`analogue carrier control.
`
`5. Conclusions
`
`The concept of fixed-bandwidth analogue selectivity for the design of wideband multi-standard
`GSM/UMTSdigital radios was presented and the implications of this approach with regards to
`current and emergingfiltering and ADC technologies were examined. Also the merits of a numberof
`receiver architectures for the implementation of such radios were discussed. It was seen that the
`performance of current ADC technology falls significantly short of that required for widcband
`GSM/DCS basestations, although the issue is less severe for the case of mobiles. From the receiver
`architectures considered, it was seen that while double-IF subsampling is readily feasible, the more
`compact single-IF subsampling architecture is only feasible with extreme subsampling and is the
`preferred option once fast ADC technology becomes available (the same applies to subsampling
`directly at RF). Bascband sampling architectures arc most suitable for
`implementation of
`programmable-bandwidth multi-standard radios.
`
`Acknowledgements: The contributions by our colleagues in the FIRST project are gratefully
`acknowledged.
`
`References
`
`[1] ACTS FRAMESdeliverable AC090/NOK/AI0/DS/R/0087/b1: “Basic “Description of Multiple
`Access Scheme, Nov. 1996.
`
`[2]
`
` Friedrichs,B.; Karimi,R.: “Flexible Multi-Standard Terminals for 2nd and 3rd Generation
`Mobile Systems”, ACTS Mobile Telecommunications Summit, Granada (Spain), Nov. 1996,
`p.750-761.
`
`[3] GSM Recommendation 05.05.: “Radio Transmission and Reception”, March 1996.
`
`[4] Kester,W.; et. al.: “High Speed Design Techniques”, Analog Devices, 1996.
`
`[5] Hedberg,B.: “Technical Challenges in Introducing Software Radio for Mobile Telephony Base
`Stations”. ACTS Software Radio Workshop,Brussels, May 1997.
`
`7
`
`

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`»1984-1992
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`Ausgewahite Veroffentlichungen (in Bearbeitung)
`1981
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`Friedrichs,B.: Zur Realisierung von Empfangernfiir die Basisband-Dateniibertragung im IDN. Vortrag bei
`der Professoren-Besprechung bei AEG-Telefunken Backnang, Oktober 1981.
`
`1985
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`Friedrichs,B.: Zur Dimensionierung des Echolischers fiir den ISDN-Netzabschluf. Vortrag beim
`Elektrotechnischen Kolloquium, Universitat Erlangen-Niirnberg, November 1985.
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`1988
`
`Friedrichs,B.: Analyse und Dimensionierung wertdiskreter adaptiver Kompensationsfilter. ANT
`Nachrichtentechnische Berichte Heft 5, Seiten 30-39, November 1988.
`
`1992
`
`Friedrichs,B.: Buchbesprechung,,J.R.Johuson: Digitale Signalverarbeitung*. Frequenz 46(1992)3-4, Seite 116.
`
`Friedrichs.B.: Analysis of Finite-Precision Adaptive Filters. Part I: Computation of the Residual Signal
`
` Selected Publications (under construction)
`
`Friedrichs,B.: Analysis of Finite-Precision Adaptive Filters. Part Il: Computation of the Residual Signal
`Distribution. Frequenz,46(1992) 11-12, Seiten 262-267. (503 KB)
`
`Friedrichs,B.: Book Reviewof,,J.C.Bic, D.Duponteil, J.C.Imbeaux: Elements of Digital Communication“.
`Frequenz 46(1992)11-12, Seiten 299-300.
`
`1993
`
`Friedrichs,B.: Dateniibertragungsverfahren fiir sicherheitsrelevante Dienste in offenen
`Kommunikationsnetzen. Vortrag beim Fakultatskolloquium, Universitat Karlsruhe, Februar 1993.
`
`Friedrichs.B.: Verfahren der Codierungstheorie und Kryptographie fiir sicherheits-relevante Kommunikation.
`Seminarvortrag beim Arbeitsbereich Digitale Kommunikationssysteme, Universitit Hamburg-Harburg, Juli
`1993.
`
`Friedrichs,B.: Sicherungsverfahren fiir die Dateniibertragungiiber ,,Bedrohte Kandle* bei
`sicherheitsrelevanten Diensten. ANT Nachrichtentechnische Berichte Heft 10, Seiten 47-63, August 1993.
`
`Friedrichs,B.: Verfahren der Codierungstheorie und Kryptographie fiir sicherheits-relevante Kommunikation.
`Vortrag bei der Professoren-Besprechung bei ANT Nachrichtentechnik Backnang, Oktober 1993.
`
`1994
`
`Friedrichs.B.: Sicherungsverfahren fiir die Dateniibertragung iiber Bedrohte Kanale bei sicherheitsrelevanten
`Diensten. Teil I. Funkspiegel, Heft 1/94, 8. 3-14. (reprint from August 2003)
`
`Friedrichs,B.: Sicherungsverfahren fiir die Dateniibertragung tiber Bedrohte Kandle bei sicherheitsrelevanten
`Diensten. Teil I]. Funkspiegel, Heft 2/94, S. 3-11. (reprint from August 2003)
`
`Friedrichs.B.: Verfahren zur Kanalcodierung und Authentizitat fiir sicherheits-relevante
`Mobilkommunikation. §.Aachener Kolloquium Signaltheorie, Aachen, Marz 1994, Tagungsband, S. 39-92.
`
`Friedrichs,B.: Zur Fehlererkennungsfahigkeit von Random Codes am Beispiel des Message Authentication
`
`
`Code. ITG-Fachtagung Codierung fiir Quelle, Kanal und Ubertragung, Miinchen, Oktober 1994. ITG-
`Fachbericht 130, S. 145-152. (330 KB)
`
`Friedrichs,B.: Codierung und Kryptographiefiir sicherheitsrelevante mobile Kommunikation.
`Kolloquiumsvortrag, Universitat Chemnitz-Zwickau, Dezember 1994.
`
`1995
`
`Friedrichs,B.: Authentische und zuverlassige Mobilkommunikationfiir sicherhetts-relevante Anwendungen.
`Teil I: Sicherheitsanforderungen und grundlegende Verfahren. Frequenz, 49(1995)1-2, S. 17-27. (1270 KB)
`
`Friedrichs.B.: Authentische und zuverlissige Mobilkommunikationfiir sicherheits-relevante Anwendungen.
`
`Teil 1: Systemarchitektur und Eimbettung m GSM. Frequenz, 49(1995)3-4, S. 48-57. (1240 KB)
`
`1996
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`9
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`Friedrichs,B.: Ein Multistandard-Terminal fiir Mobilfunksysteme der heutigen und zukiinftigen Generation.
`Kolloquiumsvortrag, Universitat des Saarlandes, Juni 1996.
`
`
`
`Friedrichs,B.; Karimi,R.(Motorola/UK): A Flexible Multi-Mode Terminal for 2nd and 3rd Generation Mobile
`Standards.ACTS Mobile Communication Summit '96. Granada (Spanien), November 1996. Tagungsband,
`VoLIT, S. 750-761. (81 KB)
`
`1997
`
`Friedrichs,B.: Flexible und adaptive Multistandard-Terminals fiir zukiinftige Mobilfunksysteme.
`Kolloguiumsvortrag, Universitat Kaiserslautern, Januar 1997.
`
`Software-Radio Architektur. IEEE Communications
`Friedrichs,B.: Multi-Standard Mobilfunk-Empfainger mit
`
`Society German Chapter, Workshop Kommunikationstechnik. Ginzburg, Januar 1997. Universitatsverlag
`Uhm,TagungsbandS. 107-114. (161 KB)
`
`Friedrichs.B.: Multistandard-Terminals mit Software-Radio Architektur fiir zukiinftige Mobilfunksysteme.
`Vortrag beim Elektrotechnischen Kolloquium, Universitat Erlangen-Niirnberg, Februar 1997.
`
`Karimi,R.(Motorola/UK); Friedrichs,B.: WidebandDigital Receivers for Multi-Standard Software Radios.
`ACTS Mobile Communication Summit '97. Aalborg(Danemark), Oktober 1997. Tagungsband, VoLI, S. 392-
`398. (85 KB)
`
`Friedrichs,B.: Architektur flexibler und adaptiver Breitband-Terminals fiir zukiinftige Mobilfunksysteme.
`Kolloquiumsvortrag, Technische Universitat Hamburg-Harburg, Oktober 1997.
`
`Friedrichs.B.: Multistandard-Transceiver mit Software-Radio Architekturfiir zukiinftige Mobilfunksysteme.
`Kolloguiumsvortrag, Universitat Kiel, November 1997.
`
`1998
`
`
`
`Friedrichs.B.; Hespelt.V.: Comparison of FDD and TDD for HIPERACCESSIncluding Cellular Aspects.
`Contribution to ETSI BRAN#12, Sophia Antipolis (France). December 1998. (65 KB)
`
`1999
`
`Friedrichs,B.: Ubertragungstechnik fiir breithandige Funk-Zugangsnetze. Kolloquiumsvortrag, Universitit
`Darmstadt, Mai 1999.
`
`Friedrichs,B.: On the channel bandwidth of Hiperaccess Systems. Contribution to ETS] BRAN#15, Eindhoven
`(The Netherlands). September 1999. (177 KB)
`
`
`Friedrichs,B.: On the statistical multiplex of HA systems. Contribution to ETSI BRAN#16, Athens (Greece).
`November 1999. (82 KB)
`
`Friedrichs,B.: Principles of FrequencyAllocation for HA Systems. Contribution to ETSI BRAN#16, Athens
`(Greece). November 1999, (18 KB)
`
`Friedrichs,B.; Fazel,K.; Hespelt,V.: Transmit Power for TDMA Systems. Contribution to ETSI BRAN#16,
`Athens (Greece). November 1999. (22 KB)
`
`2000
`
`Friedrichs,B.: Zur Architektur breitbandiger drahtloser Zugangssysteme. Technische Universitat IImenau,
`Januar 2000.
`
`Friedrichs,B.: Architektur und Ubertragungstechnik breithandiger Funk-Zugangsnetze. Universitat Ulm,
`Januar 2000.
`
`Friedrichs,B.: Sichere und zuverlassige Kommunikation in ATM-basierten drahtlosen Zugangsnetzen.
`Kolloquiumsvortrag. Februar 2000, Ruhr-Universitat Bochum (Germany).
`
`Friedrichs,B.: Architektur von Point-to-Multipoint Systemen. Kolloquiumsvortrag, Hochschule Bremen, Marz
`2000. (1900 KB)
`
`
`Friedrichs.B; Fazel.K.: Drahtlose breitbandige IP/ATM-gestiitzte Zugangsnetze; Stand bei ETSI BRAN
`HiperAccess. ITG-Diskussionssitzung..IP itiber Funk“. Miinchen, Institut fiir Rundfunktechnik, Marz 2000.
`(1275 KB)
`
`Friedrichs,B.; Fazel.K.: Efficient Multiple Access Schemes for Wireless Broadband Point-to-Multipoint Access
`
`Networks. 7th Europ. Conf.on Fixed Radio Systems and Networks (ECRR 2000),Sept. 2000, Dresden.(271
`KB)
`
`
`Fazel,K.; Friedrichs,B.: Adaptive Channel Coding & Modulation for TDMA-Based Wireless Broadband Fixed
`
`Cellular Systems. 7th Europ. Conf.on Fixed Radio Systems and Networks (ECRR 2000),Sept. 2000, Dresden.
`
`10
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`10
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`

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`(420 KB)
`
`Friedrichs,B.: Adaptive Modulation und Codierungfiir breitbandige drahtlose ATM-basierte Punkt-zu-
`Mehrpunkt Zugangsnetze. Kolloquiumsvortrag. November 2000, Universitat Wien (Austria). (1600 KB)
`2002
`
`Friedrichs,B.; Buscato,M.(Ericsson/Italy); Cavalli,G.(Siemens/Italy);Emanuelsson,P.(Telia/Sweden);
`Salokannel,J.(Nokia/Finland); Stanwood,K.(Ensemble/USA); WahLS.(Alcatel/Germany); Zoric.M.(ETSI
`PTCC/France): The Data Link Control Layer of the ETSI BRAN HiperAccess Standard. Europ. Conf.on
`Fixed Radio Systems and Networks (ECRR 2002), September 2002, Lisbon (Portugal). Conference cancelled
`after acceptance of submissions. (226 KB)
`
`2003
`
`Friedrichs,B:: Evaluating the Benefits of UsingHiperAccess as a Backhauling Solution for 2G and 3G Mobile
`
`Networks. ITR’s 5th Annual Forum on Transmission Network Solutions for Mobile
`
`Operators. 20-22.5.2003,
`Berlin (Germany). (3090 KB)
`
`Friedrichs,B.: ETSI Standards for Broadband Radio Access Networks (BRAN Project). Joint AHCIET
`(Asociacion Hispanoamericana de Centros de Investigacion y Empresas de Telecomunicaciones) - CITEL
`(Comision Interamericama de Telecomunicaciones) Seminar on Broadband Wireless Access. 20-21 October
`2003, San Salvador (El Salwador). (2860 KB)
`
`Friedrichs,B.: ETSI BRAN (Broadband Radio Access Networks) Standardisierungfiir breithandige drahtlose
`PMP Systeme. ITG FA 7.2 Diskussionssitzung Fixed Wireless Access — Neue Technologien, Anwendungen und
`Standardisierung fiir Punkt-zu-Punkt und Punkt-zu-Mehrpunkt Funksysteme. 13.11.2003, Backnang
`
`(Germany). (1380 KB)
`:
`
`2004
`
`Friedrichs,B.: ETSIStandards for Broadband Radio Access Networks (BRAN). WCA 10th AnnualTechnical
`
`Symposium. January 2004, San Jose, CA (USA). (1240 KB)
`
`Friedrichs,B.: ETSI Standards for Broadband Radio Access Networks (BRAN Project). ITU-D rapporteur’s
`group meeting Q20/2. January 2004, Geneva (Switzerland). Paper was presented by Molly Gavin
`(QualcommUSA)
`
`Friedrichs.B.: Statistical Multiplex Gain in Cellular Communication Systems. Karlsruher Stochastik-Tage
`2004 (6th German Open Conference on Probability and Statistics).Tagungsband Abstracts. March 2004,