United States Patent (19)
`Shenoi
`
`USOO5764704A
`Patent Number:
`11
`(45) Date of Patent:
`
`5,764,704
`Jun. 9, 1998
`
`54 DSP MPLEMENTATION OF A CELLULAR
`BASE STATION RECEIVER
`
`75) Inventor: Kishan Shenoi. Saratoga, Calif.
`73) Assignee: SymmetriCom, Inc., San Jose, Calif.
`
`21 Appl. No.: 664,705
`22 Filed:
`Jun. 17, 1996
`(51) Int. Cl. ...
`... H04L27/14: HO3D 3/00
`52 U.S. Cl. .............
`... 375/324; 375/326; 329/327
`58) Field of Search ..................................... 375/229, 260,
`375/27.316,322.324, 326; 329/315,
`327; 455/205
`
`56)
`
`References Cited
`U.S. PATENT DOCUMENTS
`1/1972 Kobayashi .............................. 329/311
`3,634,773
`4,803,739 2/1989 Daikoku .......
`... 455/47
`5,177,611
`1/1993 Gibsonb et al.
`... 348/611
`5.434,577
`7/1995 Baghdady ............................... 342/380
`
`Primary Examiner-Chi H. Pham
`Assistant Examiner-Emmanuel Bayard
`Attorney, Agent, or Firm-Wilson Sonsini Goodrich &
`Rosati
`ABSTRACT
`57)
`Demodulating FM signals using digital signal processing
`extracts a carrier signal from digitized channel signals,
`multiplies the digital channel signal with this extracted
`carrier signal, and further filters out the carrier signal to
`produce the demodulated signal. The DSP technique first
`down converts a group of channels to baseband which are
`then processed through an A/D converter to produce a
`digitized composite signal. A bank of bandpass filters,
`typically based on FFT processors, applied to the composite
`signal produce (a group of) digitized channel signal(s). The
`digitized channel signal is then demodulated by recovering
`a carrier signal by digitally filtering, for example, using a
`Hilbert bandpass filter, the channel signal and digitally
`filtering the product of the carrier signal and the channel
`signal to recover the modulating voice signals.
`15 Claims, 8 Drawing Sheets
`
`AM DEMODULATOR
`
`122
`DIGITAL -
`CHANNEL
`SIGNAL
`P (nt)
`
`
`
`124 HILBERT TRANSFORMING
`BANDPASS FILTER s = 30kHz
`
`Ex.1018
`APPLE INC. / Page 1 of 14
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`

`

`U.S. Patent
`
`Jun. 9, 1998
`
`Sheet 1 of 8
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`5,764,704
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`APPLE INC. / Page 2 of 14
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`

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`U.S. Patent
`
`Jun. 9, 1998
`
`Sheet 2 of 8
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`5,764,704
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`Ex.1018
`APPLE INC. / Page 3 of 14
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`

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`U.S. Patent
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`Jun. 9, 1998
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`Ex.1018
`APPLE INC. / Page 4 of 14
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`U.S. Patent
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`Jun. 9, 1998
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`U.S. Patent
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`Jun. 9, 1998
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`Sheet 5 of 8
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`Ex.1018
`APPLE INC. / Page 6 of 14
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`

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`U.S. Patent
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`Jun. 9, 1998
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`Sheet 6 of 8
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`5,764,704
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`APPLE INC. / Page 7 of 14
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`

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`U.S. Patent
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`Jun. 9, 1998
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`Sheet 7 of 8
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`5,764,704
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`Ex.1018
`APPLE INC. / Page 8 of 14
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`

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`U.S. Patent
`
`Jun. 9, 1998
`
`Sheet 8 of 8
`
`5,764,704
`
`BANDPASS FILTER
`OVER ENTRE BAND
`
`SELECT A GROUP OF
`CHANNELS
`
`SHIFT THE SELECTED
`GROUP OF CHANNELS TO F
`
`201
`
`202
`
`204
`
`BANDPASS OVER THE
`ENERE SELECTED
`GROUP OF CHANNELS
`
`SHIFT THE SELECTED
`GROUP TO BASEBAND
`
`LOW PASS FILTER OVER
`THE SELECTED GROUP
`
`DIGITAL SAMPLING
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`206
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`208
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`210
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`BANDPASS FTER
`FOR CHANNELIZATION
`DCTA --
`CHANNEL SIGNA
`216
`
`HILBERT BANDPASS FILTER
`
`DIGITALLY MULTIPLY THE
`DIGITAL CHANNEL SIGNAL WITH
`THE OUTPUT OF HILBERT BANDPASS FILTER
`
`214
`212 2
`
`-
`
`MULTIPLE CHANNES
`CAN BE PROCESSED
`IN PARALE
`
`218
`
`DIGITALLY ELIMINATE CARRIER
`SIGNAL TO RECOVER
`MODULATING SIGNAL
`
`
`
`220
`
`SHIFT CENTER FREQUENCY
`OF BANDPASS FILTER
`
`(F SEQUENTIAL)
`
`PROCESSED
`YES
`
`222
`
`FIG.8
`
`Ex.1018
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`5,764,704
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`10
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`15
`
`1
`DSP MPLEMENTATION OFA CELLULAR
`BASE STATION RECEIVER
`BACKGROUND OF THE INVENTION
`1. Field of the Invention
`This invention relates to the demodulation of signals that
`are transmitted in cellular radio telephone systems, and more
`particularly to demodulating FM signals using digital signal
`processing (DSP).
`2. Description of Related Art
`Conventional methods for FM demodulation are typically
`based on analog systems. While analog FM demodulation
`techniques are inexpensive, they are not generally flexible to
`accommodate different types of FM demodulation tech
`niques for a variety of different types of signals such as a
`cellular environment. As in other areas, the use of digital
`signal processing (DSP) creates the potential for greater
`flexibility using FM signals, such as in cellular telephony.
`In a cellular telephone system, the channel assignments to
`a given base station is somewhat arbitrary, since any com
`bination of channels can be assigned to any one base station.
`This causes problems when dealing with demodulation of
`the received or incoming signal using analog FM demodul
`lation technique, since a typical FM demodulator, which is
`generally a phase lock loop (PLL) in an analog form, is only
`responsive to a certain frequency range. The problem arises
`when there are multiple channels which need to be demodu
`lated. Since the assignment of those channels may be
`arbitrary to any given base station, a base station has to be
`designed to handle any possible combination of channels or
`frequencies. In addition, the channels assigned to a given
`base station may change over time, based upon various types
`of studies of the cellular network to accommodate the
`changes in channel demand. With standard FM techniques,
`one can use a bank of standard analog PLLs that are tunable
`across the entire range of FM spectrum. This range is very
`broad, and thus, implementation of such analog PLLs
`becomes expensive and complicated. Moreover, analog cir
`cuits used for FM demodulation include phase or frequency
`locked loops and many varieties of discriminator circuits.
`Generally, analog FM demodulation is not flexible when
`several channels have to be demodulated, such as in cellular
`communications, given that the FM signal in a cellular
`system may be "standard cellular" (AMPS) or narrow band
`AMPS (NAMPS).
`SUMMARY OF THE DISCLOSURE
`It is an object of the present invention to provide a digital
`FM demodulation method and apparatus by extracting a
`carrier signal, preferably using a digital Hilbert transforma
`tion for subcarrier recovery and preferably using a digital
`AM demodulation with multiplication and integration to
`extract voice signals.
`Rather than using an analog implementation of FM
`demodulation, one can use a DSP implementation. The DSP
`55
`implementation of FM demodulation using AM demodula
`tion techniques will provide flexibility, because adaptation
`to different signals and different frequencies require minimal
`changes to the device, such as changing certain constants
`and subroutines in the DSP software. The DSP implemen
`tation of an FM demodulation enhances flexibility not
`available with analog FM demodulation technique. For
`example, the digital FM demodulation can provide the
`freedom of easily changing parameters to accommodate
`various frequencies and can be readily adapted to a cell site
`to accommodate TDMA, NAMPS, GSM and AMPS tech
`niques.
`
`65
`
`2
`According to one embodiment of the present invention, a
`method of demodulating FM signals using digital signal
`processing (DSP) includes the steps of selecting in the
`analog domain a band of channels by using a bandpass filter.
`The band of channels from, for example, a cellular or other
`wireless network received at the base station are then shifted
`down to IF. A group of adjacent channels selected from the
`band from the band of channels are selected from IF and are
`shifted down to base band, sampled at twice the upper band
`limitation to shift into the digital domain. Each group of
`channels is separately processed at base band with a DSP
`implementation. For example, the channel at the base band
`is sampled to produce a digital channel signal having
`in-phase and quadrature components, in which the in-phase
`component has a carrier component and the quadrature
`component has the modulating signal (speech) component.
`Each of the digitally sampled channels is separately pro
`cessed. Using a narrow frequency bandpass algorithm, such
`as a Hilbert transform bandpass algorithm, the digital chan
`nel signal is filtered to extract the carrier signal. The carrier
`signal is then multiplied digitally with the digital channel
`signal to form a mixed signal which contains the modulating
`signal. Finally, the carrier and higher frequency signals are
`filtered out to produce only the modulating signal. In the
`preferred embodiment of the present invention, the Hilbert
`bandpass filter algorithm has a pass band of less than 600 Hz
`to extract a carrier component at the center frequency of the
`filter,
`Other features and advantages of the invention will
`become apparent from the following detailed description,
`taken in conjunction with the accompanying drawings which
`illustrate, by way of example, various features of embodi
`ments of the invention.
`BRIEF DESCRIPTION OF THE DRAWINGS
`A detailed description of embodiments of the invention
`will be made with reference to the accompanying drawings,
`wherein like numerals designate corresponding parts in the
`several figures.
`FIG. 1 is a block diagram showing equivalence of phase
`and frequency modulation;
`FIG. 2 is a block diagram of Armstrong modulator for
`narrow band angle modulation (phase modulation);
`FIGS. 3A-3C show a conversion of group of channels to
`digital format;
`FIG. 4 is a bandpass filter bank for channelization;
`FIG. 5 is a per channel frequency domain representation;
`FIG. 6 is a block diagram showing principle of DSP based
`narrow band angle modulator;
`FIG. 7 is a frequency domain representation of a channel
`signal; and
`FIG. 8 is a flow diagram for demodulating FM signals
`using digital signal processing.
`DETALED DESCRIPTION OF THE
`PREFERRED EMBODMENTS
`Amethod of demodulating FM signals using DSP accord
`ing to an embodiment of the invention is shown in the
`drawings for purposes of illustration. In particular, a pre
`ferred embodiment of the present invention focuses on the
`DSP implementation of AMPS/NAMPS/TDMA. Also, as an
`illustrative example, the following description focuses on
`the demodulation of a single voice channel which is super
`imposed on a carrier using a narrow band FM.
`Cellular telephone systems in the U.S. operate in the
`frequency band between 824 MHZ and 894MHz. The lower
`
`25
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`45
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`3
`portion, between 824MHz and 849 MHz, is allocated for the
`"reverse" direction of transmission, from the mobile to the
`base station. The upper portion, between 869 MHz and 894
`MHz, is allocated for the "forward" direction of
`transmission, from the base station to the mobile. Typically,
`a nominal channel spacing is 30 KHz. that is, the frequency
`band is subdivided into "blocks" with a center frequency
`spacing of 30 KHz and nominal bandwidth of 30 KHz (15
`KHZ on either side of the center or carrier frequency). Each
`channel, uses carrier using angle modulation, specifically
`frequency modulation (FM), with a maximum frequency
`deviation of 12 KHz. Since the voice channel bandwidth is
`nominally 3 KHz, the voice channel would thus occupy a
`bandwidth of 2*(3+12)=30 Khz by applying Carlson's rule.
`The prescribed frequency allocation permits up to 832
`blocks, usually considered as two “bands." A and B, each
`having a capacity of 416 blocks.
`The frequency spectrum is considered to be subdivided in
`a frequency division multiplex (FDM) manner, with each
`block referred to as a channel. Each channel represents one
`telephone conversation. Each mobile unit is assigned a
`specific carrier frequency for the reverse direction, typically
`45 MHz below the carrier frequency used in the forward
`direction by the base station. Such a scheme is often referred
`to as "AMPS" (Advanced Mobile Phone Service).
`A fundamental principle of cellular telephony is that any
`geographical area can be subdivided into "cells," with
`adjacent cells using nonoverlapping frequency blocks.
`Consequently, each cell, and thus base station, can use only
`a subset of these 416 channels. This limits the number of
`simultaneous users in an area served by a base station. To
`increase the number of subscribers served, other modes of
`operation have been implemented. The principal idea is to
`use each channel to serve more than one subscriber. The
`35
`premise of NAMPS is to subdivide each 30 KHZ block into
`3 sub-blocks of 10 KHZ each and use very narrow band FM
`for each voice channel. This method increases the number of
`subscribers that can be served simultaneously by each base
`station by a factor of 3.
`A second approach to increase the number of simulta
`neous subscribers. again by a factor of 3, is to use the
`technique of TDMA (Time Division Multiple Access).
`TDMA presumes that the voice channel is digitized and
`encoded so as to use approximately 8 kbits/sec for a voice
`channel. The 30 KHz bandwidth used in cellular telephony
`can support a modulated digital bit-stream. By imposing a
`frame structure on this digital bit-stream, multiple subscrib
`ers can be accommodated within the block by assigning
`different "time-slots" to different subscribers. Nominally,
`each time slot of a full channel transports the signal for one
`subscriber. By maintaining the FDM hierarchy of 30 KHz
`blocks, TDMA and analog cellular, such as AMPS, can
`coexist in the same cell site. Thus, any 30 KHZ block can be
`assigned to digital (TDMA) or to analog transmission.
`As shown in FIG. 1, a frequency modulation 28 may be
`achieved by using a phase modulator 27 with an integrator
`26 which integrates the signal 24 prior to phase modulation.
`The integration is equivalent to a filter providing a 6
`dB/octave response, which is the opposite of the preempha
`sis filter. Thus, by using a phase modulator 27, a preempha
`sis circuit can be eliminated.
`Unlike amplitude modulation (AM), which is considered
`"linear,” angle modulation is inherently nonlinear. This
`means that in AM, the bandwidth of the modulating signal
`is preserved. The composite signal is achieved by translating
`the modulating signal in frequency by an amount equal to
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`4
`the carrier frequency. In angle modulation, the bandwidth is
`several times that of the modulating signal. Nonetheless, for
`low modulation indices, the phase modulation may be
`treated approximately similar to the amplitude modulation
`for purposes of DSP
`The above-described relationship between PM and AM is
`the basis for the Armstrong Modulator, as shown in FIG. 2.
`Note that the resultant (composite) signal can be viewed as
`the combination of an AM signal (double sideband sup
`pressed carrier, DSB-SC) to which a carrier component has
`been added in the quadrature component. In a conventional
`DSB-AM signal (non suppressed carrier), the additive car
`rier component is an in-phase component, in which the
`implied carrier is used to generate the DSB-SC signal. Thus,
`if a DSB-SC signal is added with an in-phase carrier
`component, a regular DSB-AM signal is produced. If the
`additive carrier component is in the quadrature component.
`the resultant signal is equivalent to a narrow band PM. The
`above description can be mathematically expressed as fol
`lows:
`If |m(t) <<1 (modulating signal is small), then the com
`posite PM signal, vpm(t). can be expressed as:
`
`vpm(t) = A cos(27tft + m(t))
`s A cos(2ft) - Am(t) sin (2nft)
`
`For conventional demodulators using analog
`implementations, the processing is done either directly at the
`carrier frequency or at an intermediate frequency (IF). The
`processing may include limiting (hard or soft) to minimize
`any extraneous AM component. The demodulation is imple
`mented using either a PLL (phase lock loop) method or a
`frequency discriminator method. None of these processors
`(clipping. PLL, frequency discriminator) is suitable when
`the channel signal is in a digital form at baseband. Clipping.
`for example, would generate harmonics which, in an analog
`implementation, can be filtered out. In a DSP
`implementation, the clipping has the effect of smearing,
`introducing inband spurious components via aliasing. Thus,
`emulating an analog processing is not appropriate in all
`CaSS.
`Since DSP techniques are applicable in the case of AM, a
`digital demodulator may be devised by taking the logical
`inverse of the Armstrong modulator described above. This is
`an underlying principle of the preferred embodiment of the
`present invention. All the processes are well suited for
`implementation in a DSP processor and may be imple
`mented in hardware, firmware or software.
`The process of converting a group of channels to digital
`signals, as a way of example, will now be discussed with
`regard to FIGS. 3A-3C. The analog front-end circuits
`choose a group of preferably in FM channels shown in FIG.
`4A as fourteen 30 KHz channels plus two guard bands and
`translate the group down to base band. An FM modulated
`signal of the n channels is bandpass filtered with a pre
`selected bandpass filter 42 over the entire band, which
`reduces interference at IF. Next, the group of channels is
`translated to IF by multiplying the output from the bandpass
`filter 42 with a first fosc 41. The output from the multiplier
`43 is then passed through a sharp bandpass filter 44 which
`completes the selection of the group of channels. Abandpass
`filter bandpass filter 44 have the frequency response of FIG.
`3B is applied over the entire band of channels 30 of FIG.3A.
`The selection of channels 1-14, with a sharp roll off of
`preferably 6 dB per octave in guard channels 0 and 15.
`reduces interference. A second multiplier 45 of FIG. 3C
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`5
`multiplies the output from the sharp bandpass filter 44 with
`a second fosc 49 to translate the group of channels to
`baseband. The output from the second multiplier 26 is
`filtered by an anti-aliasing low pass filter 46 before being
`converted to the analog to digital (AWD) converter 47. The
`output from the low pass filter 46 has a total of 16 channels
`from band 0 to band 15, each channel being about 30 KHz
`wide. As required by the Nyquist's Theory, the A/D con
`verter 47 has a sampling rate of at least which is twice (e.g.,
`960 KHZ) the frequency range of the selected group of
`channels (e.g., 480 KHz) at the base band. For example.
`when n=16 (a number of channels selected for DSP), the
`nominal sampling rate for the A/D converter 47 is 960 KHz.
`Not shown in FIG. 3C is an automatic gain control (AGC)
`function performed prior to AND conversion to maximize
`the resolution provided by the A/D operation at full range.
`Once the selected group of channels are shifted to base
`band, each channel therein can be processed separately and
`in parallel. FIG. 4 shows a bandpass filter bank for chan
`nelization which is typically based on 32-point DFT
`20
`(discrete Fourier transformation). The bandpass filter center
`frequencies, for the purposes of illustration, are 30 n KHz
`where n=0 to 15. In addition, the filter transition band is 15
`KHZ on each side of each bandpass filter. The bandpass filter
`separates the channels which, after undersampling. can be
`viewed as a single channel stream at a sampling rate of 120
`KHz. It is important to recognize that the channel sampling
`rate, for example, 120 KHz, is twice the Nyquist sampling
`rate. In the preferred implementation of the bandpass filter
`bank, with implied undersampling, the channel frequency
`spectrum to be processed is repositioned, as shown in FIG.
`5 by techniques known to those of skill in the art as
`described in Chapter 7 of digital Signal Processing in
`Telecommunications (Prentice Hall, 1995) by Kishan
`Shenoi (which is incorporated herein by reference). The
`effect of frequency translation and undersampling is to
`position the carrier frequency of the channel at 30 KHz
`(nominal), which is one-fourth the sampling frequency. The
`carrier frequency at baseband (or at the IF frequency used
`for processing) should be an integer submultiple of the
`sample rate (e.g. /2, 1/3, 1/4, etc.) so that the processing is
`easier. By using a higher sampling rate (by about a factor of
`2) than the minimum sampling rate, the impact of aliasing is
`reduced and the complexity of the filters used in the filter
`bank is reduced as well.
`For each channel, the 30 KHz spectrum containing the
`signal is relatively clean, since there is a 15KHz guard band
`on both sides of the signal band to prevent interference
`components from corrupting the signal band. In a preferred
`embodiment of the present invention, the bandpass filter
`50
`bank may be implemented by using a DFT (e.g., an FFT
`implementation). The output signal of the DFT processing
`provides in-phase and quadrature components (i.e., complex
`signals) of each channel. The complex signal is useful in the
`case of TDMA, since the digital demodulation is simplified
`when the in-phase and quadrature components are available
`simultaneously.
`As shown in FIG. 6, the digital channel signal 122 in the
`time domain is Xe(nT) with the sampling rate of f=120
`KHz. The digital channel signal 122 is the composite
`channel signal obtained from the bandpass filter bank which
`follows the A/D converter 47 of FIG. 3C. The frequency
`domain representation of x(nT) is X(f), and both are
`mathematically expressed respectively as follows:
`
`35
`
`6
`The frequency domain representation of X(f) is shown in
`FIG. 7. A carrier component 141 is represented by impulses
`142 and 143 (i.e., delta functions) at-30 KHZ and 30 KHz.
`The signal components 146 and 147 are the quadrature
`component, as evidenced by the "j" in the above expression.
`of the composite signal. The signal components 146 and 147
`are in an AM signal form, namely, double sideband sup
`pressed carrier (DSB-SC) AM. Since the voice signal has no
`dc component, a narrow frequency band pass around 30
`KHz of the digital channel signal will provide just the carrier
`component 143.
`A Hilbert transform filter 124 commonly used in DSP may
`be used for filtering to obtain the 30 KHz carrier. The Hilbert
`transform filter 124 is a digital filter with the frequency
`response:
`
`and H(f) is a narrowband magnitude response centered at 30
`KHZ For example, in an FIR (finite impulse response) filter
`having constrained coefficients to exhibit a negative sym
`metry (odd function) about the center, the frequency
`response is guaranteed to be purely imaginary, which cor
`responds to a 90 degree phase shift. In particular, an odd
`length, odd-symmetry filter (N the length of the filter being
`an odd integer) will have the phase shift of 90 degrees and
`a flat delay of (N-1)/2 samples. Since N is an odd integer.
`the flat delay will be an integer number of samples. Further.
`if N is chosen so that (N-1)/2 is a multiple of 4, an added
`benefit may be obtained. Since the carrier is nominally 30
`KHz, the period of the carrier is 4 samples with a 120 KHz
`Sampling rate. Consequently, if the flat delay is chosen to be
`a multiple of 4. the phasing at the output of the Hilbert
`transforming filter 124 will be accurate,
`The bandwidth of the Hilbert transform filter 124 (or a
`processor that implements the Hilbert transform filter) is set
`to be no more than about 300 Hz wide on each side of a
`carrier signal. This is because there is no audio signal within
`300 Hz of the carrier due to the audio signal filtering in the
`transmitter, Thus, the total band width of the Hilbert trans
`form filter 124 is preferably less than about 600 Hz. The
`output from the Hilbert transform filter 124 is a recovered
`carrier signal for the received digital signal. The carrier
`signal, which is orthogonal to the digital signal, is then
`multiplied with the digital signal to further separate the
`carrier signal from the voice signal.
`Portions 126 and 128 of the processing shown in FIG. 6
`correspond to a digital implementation of a synchronous
`demodulator for AM. Conventionally, the difficulty in syn
`chronous AM demodulation (as opposed to common enve
`lope detection) is generation of a local carrier y(nt) that is
`synchronous in phase and frequency to the carrier used for
`creating the AM composite signal x(nt). However, accord
`ing to the preferred embodiment of the present invention, by
`design, it is guaranteed that the "local oscillator" for gen
`erating the carrier y(nt) will be substantially of the correct
`frequency and the correct phase. As discussed above, the
`carrier signal is obtained by detecting the in-phase compo
`nent 143 (FIG. 7) of the digital channel signal 122 by using
`a digital Hilbert transform bandpass filter. The output 129
`m(nt) of the demodulator 126 will be a scaled (in amplitude)
`digital version of the modulation signal m(t) at the trans
`mitter provided that the magnitude of the modulation index
`used in the modulator (not shown) is substantially less than
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`7
`1. The software DSP based narrow band angle demodulator
`126 multiplies the recovered digital channel signal:
`X(nT-Acos(2 font-A m(nTsin(27tfont)
`by the carrier:
`y(nT-Acos(2tfon T-7tl2)=Asin(2rtfnT)
`yielding a demodulator output:
`
`8
`such modifications as would fall within the true scope and
`spirit of the present invention.
`The presently disclosed embodiments are therefore to be
`considered in all respects as illustrative and not restrictive,
`the scope of the invention being indicated by the appended
`claims, rather than the foregoing description, and all changes
`which come within the meaning and range of equivalency of
`the claims are therefore intended to be embraced therein.
`What is claimed is:
`1. A method of digitally demodulating an FM signal at one
`of an IF and base band frequency created by frequency
`modulating a carrier signal with a modulating signal, the FM
`signal having in-phase and quadrature components, wherein
`the in-phase component has the carrier signal and the
`quadrature component has the modulating signal, the
`method comprising the steps of:
`sampling the FM signal at an appropriate sample rate to
`produce a digital channel signal;
`digitally filtering the digital channel signal to extract the
`carrier signal using a narrow frequency bandpass algo
`rithm;
`digitally multiplying the carrier signal with the digital
`channel signal to form a mixed signal which comprises
`the modulating signal and the carrier signal shifted to a
`different frequency; and
`digitally filtering the mixed signal to recover the modu
`lating signal.
`2. A method according to claim 1, wherein the narrow
`frequency bandpass algorithm is a Hilbert bandpass algo
`rithm.
`3. A method according to claim 1, wherein the narrow
`frequency bandpass algorithm has a band pass range of
`approximately 300 Hz on each side of the carrier signal.
`4. A method according to claim 2, wherein the Hilbert
`bandpass filter has a pass band centered about 30 KHz, the
`sampling rate chosen such that the baseband FM carrier is at
`30 KHZ.
`5. A method according to claim 2, wherein the Hilbert
`bandpass filter is an odd-length and odd-symmetry, FIR
`filter.
`6. A method according to claim 2, wherein the Hilbert
`bandpass algorithm comprises a phase shift of about 90
`degrees and a flat delay of (N-1)/2, in which N is an odd
`integer.
`7. A method according to claim 6, wherein the ratio of FM
`carrier frequency at the one of the IF and baseband to the
`sample rate, being an integer.
`8. A method according to claim 6, wherein (N-1)/2 is a
`multiple of 4.
`9. A method according to claim 1, wherein the filtering of
`the mixed signal to produce the modulating signal is per
`formed by filtering out high frequency signals.
`10. A method according to claim 1, wherein the sampling
`rate for sampling the FM signal is about twice the upper
`limit frequency of the FM signal.
`11. A method according to claim 1, further comprising the
`implementation of the steps of claim 1 for other digital
`channel signals in the FM signal in parallel.
`12. A method of digitally demodulating a composite
`signal, having a plurality of channels, each channel created
`by frequency modulating a carrier signal with a modulating
`signal, the FM signal having in-phase and quadrature
`components, wherein the in-phase component has the carrier
`component and the quadrature component has the modulat
`ing signal, the method comprising the steps of:
`shifting a plurality of channels of the FM signal to base
`band;
`
`The recovery of the scaled demodulated modulation signal
`is obtained just by low pass filtering with filter 128:
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`15
`
`Knowledge of y(nT) 125, the implied carrier, is, therefore,
`extremely useful. While it is not shown in the figures, this
`signal may also be used for controlling the AGC (Automatic
`Gain Control) function for sampling with AID converter 47
`the signal and/or a per-channel AGC function. In particular,
`the channel signal can be amplified (or attenuated) such that
`the signal y(nT) 125 has a known prescribed amplitude.
`The above-described implementation of DSP is for one
`channel out of n channels selected at the IF stage. The
`undersampling in the digital bandpass filter bank ensures
`that all n channels are alike in that the carrier component is
`at 30 Khz as described in Chapter VII of Kishan Shenoi's
`book. Therefore, the same steps can be implemented to
`demodulate all n Channels. This permits one DSP or a bank
`of DSP's to perform the base station functions using much
`of the same software modules.
`The DSP FM demodulation technique described above
`can be summarized using a flow diagram 200 shown in FIG.
`8. First, a received FM signal is bandpass filtered 201 over
`the entire frequency band. A group of channels are selected
`202 from the FM signal and are shifted down to IF 204 and
`then preferably to base band 206. The selected group of
`channels which are now at base band are low pass filtered
`208 to remove any high frequency signals above the upper
`band limit. The selected group of channels are then sampled
`at approximately twice its upper limit 210. One channel
`from the selected group of channels is digitally filtered
`(channelization) 212 to form a digital channel signal which
`can be digitally processed to extract a modulating signal. At
`this point, the process may be done in parallel by additional
`processors 214 for each of the channels. A carrier signal for
`the digital channel signal is extracted by performing a digital
`narrow band bandpass filtering, such as using a Hilbert
`transform filter algorithm, on the channel signal 216. The
`recovered carrier signal is then digitally multiplied with the
`digital channel signal 218 to recover the modulating signal.
`A digital low pass filter algorithm filters out the shifted
`carrier signal to recover only the modulating signal 220. If
`multiple DSP's are not being used for parallel processing
`(step 214), a single DSP may repeat the process for addi
`tional channels in the group 222. The bandpass filter algo
`rithm for the channelization provides all n channels in
`parallel so that the other channels can be digitally processed
`to extract a modulating signal in parallel.
`While the description above refers to particular embodi
`ments of the present invention, it will be understood that
`many modifications may be made without departing from
`the spirit thereof. For example, the disclosed methods of
`demodulation may be done at IF instead of at the base band
`frequency. The accompanying claims are intended to cover
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`sampling the FM band signal at a rate at least twice the
`bandwidth of the FM band signal to produce a digital
`channel signal;
`digital bandpass filtering to produce a plurality of digital
`channel signals;
`then for each digital channel;
`digitally filtering the digital channel signal to extract the
`carrier signal using a Hilbert bandpass