`
`Samsung, EX1012, p. 1
`
`Samsung, EX1012, p. 1
`
`

`

`
`
`BIPOLAR AND MOS
`ANALOG INTEGRATED
`CIRCUIT DESIGN
`
`ALAN B. GREBENE
`
`MICRO-LINEAR CORPORATION
`SUNNYVALE, CALIFORNIA
`
`A Wiley-Interscience Publication
`John Wiley & Sons
`
`New York Chichester Brisbane Toronto Singapore
`
`Samsung, EX1012, p. 2
`
`Samsung, EX1012, p. 2
`
`

`

`
`
`Copyright © 1984 by John Wiley & Sons, Inc.
`
`All rights reserved. Published simultaneously in Canada.
`
`Reproduction or translation of any part of this work
`beyond that permitted by Section 107 or 108 of the
`1976 United States Copyright Act without the permission
`of the copyright owner is unlawful. Requests for
`permission or further information should be addressed to
`the Permissions Department, John Wilcy & Sons, Inc.
`
`Library of Congress Cataloging in Publication Data:
`Grebene, Alan B., 1939-
`Bipolar and MOSanalog integrated circuit design.
`
`“A Wiley-Interscience publication.”
`Includes index.
`1. Integrated circuits. 2, Electronic circuit
`design. 3. Metal oxide semiconductors. 4. Bipolar
`transistors. I. Title. II. Title: Bipolar and M.Q.S,
`analog integrated circuit design.
`TK7874.G693
`1983
`621.381°73
`ISBN 0-471-08529-4
`
`83-6563
`
`Printed in the United States of America
`
`10
`
`9
`
`8 76 5
`
`Samsung, EX1012, p. 3
`
`Samsung, EX1012, p. 3
`
`

`

`
`
`PREFACE
`
`The contents and organization of this book are primarily aimed at the practicing
`engineer in the field of solid-state electronics. It is intended as a valuable reference
`for the IC designer and user alike. For the analog IC designer, it provides rigorous
`design guidelines and examples, while for the user, it offers a detailed analysis of
`various classes of analog circuits, points out their design philosophy, capabilities,
`and limitations, and presents application examples and guidelines.
`It is intended to be an easy and smooth reading book on a rapidly evolving,
`high-technology subject. To this end, the lengthy and detailed mathematical treat-
`ment of the subject matter is minimized. Long derivations of device or circuit
`equations are avoided wheneverpossible; instead, the emphasis is placed on the end
`result, and the basic design philosophy leading up to it, with a clear understanding
`of the underlying assumptions and trade-offs. Whenever possible, each new design
`idea or concept is also demonstrated with a practical example.
`The adventof integrated circuit technology has altered many of the established
`circuit design techniques and principles. This is particularly evident in the field of
`analog integrated circuits where the designer is faced with a new set of design
`constraints and ground rules. In writing this book, it is my intention to educate the
`practicing electronics engineer in the fundamental design principles, capabilities,
`and applications of monolithic analog circuits. However, the subject matteris treated
`rigorously and from a fundamental viewpoint, to make this book suitable as a text
`for graduate study in semiconductor circuits.
`This book is an updated sequelto an earlier book by the author, Analog Integrated
`Circuit Design (published by Van Nostrand Reinhold, 1972) which covered the
`analog IC design technology of the 1960’s. Since then, many significant changes
`have occurredin the world of microelectronics. Perhaps the most important of these
`has been the “microprocessor revolution,” which hasresulted in a truly revolutionary
`growth of digital signal-processing techniques. In turn,
`this has led to a rapid
`evolution and advancementof analog circuit methods, particularly in the areas that
`interface with digital techniques and technologies. As a result, complete LSI systems
`have evolved which combine complex analog and digital functions on the samechip.
`v
`
`Samsung, EX1012, p. 4
`
`Samsung, EX1012, p. 4
`
`

`

`vi
`
`PREFACE
`
`A great deal of this development has been possible by extending the capabilities of
`MOS devices and process technology to cover analog functions. Consequently,
`analog IC design using MOStechnology has rapidly evolved into a major area of
`growth, These developments ofrecent years are profoundly reflected in the contents
`and the organization of this book.
`In the preparation of the text, it is assumed that the reader is familiar with the
`basic theory and principles ofsolid-state devices. Therefore, the solid-state device
`theory, which is already well covered elsewhere in theliterature, is reviewed only
`briefly, and almost all of the space is devoted to circuit approaches unique to
`monolithic integrated circuits. Hybrid integrated circuits, which represent an area of
`overlap between discrete and monolithic circuits, are not covered explicitly.
`The text of the book is comprised of fifteen chapters which follow a logical
`sequence in the form of three “sections.” The first section of the book, comprised
`of Chapters 1-3, reviews the basic “tools” of analog IC design and fabrication,
`namely, process technology,
`IC components, and techniques for placing these
`components on the chip,
`that is,
`the chip layout. These chapters are intended to
`familiarize the designer with the physical structures, advantages, and limitations of
`monolithic components. This knowledge is imperative to an analog IC design
`engineer since a successful design is one that efficiently utilizes the advantages of
`monolithic devices while avoiding their shortcomings.
`The second section of the text, made up of Chapters 4—6, covers the basic
`“building blocks,” or subcircuits, of analog IC design. One important chapter in this
`section, Chapter 6, deals with the use of MOStechnology in analog or combined
`analog/digital LSI design. All the subcircuits coveredin this section serve as essen-
`tial building blocks ofthe complex IC designs that are covered in the remainder of
`the book.
`The third and main section of the book, comprised of Chapters 7-15, covers the
`entire field of analog integrated circuits by dividing them into functional categories
`and then examining cach category separately. Thus, for example, circuit classes
`such as operational amplifiers, multipliers, oscillators, phase-locked loops, filters,
`and data conversion circuits are examined separately.
`In this section, particular
`emphasis is given to the recent developments in the field of analog circuits, partic-
`ularly in the areas of switched-capacitor filters, switching regulators, voltage-
`controlled oscillators, high-resolution data conversion circuits, and the precision
`reference circuitry associated with them.
`Part of the material in this book is patterned after a sequence of graduate level
`coursesin integrated electronics which I taught at Santa Clara University. Therefore,
`when preceded by courses on solid-state circuits and semiconductorelectronics, this
`book will be well suited for a senior or graduate level course.
`I am grateful to many people who have contributed directly or indirectly to the
`preparation of this book. In particular, I wouldlike to thank my wife, Karen, who
`has been a constant source of encouragementfor me during the long years of effort
`that have goneinto this book. I would also like to extend my appreciation to many
`colleagues and associates in the IC industry for their.assistance and guidancein the
`
`Samsung, EX1012, p. 5
`
`Samsung, EX1012, p. 5
`
`

`

`PREFACE
`
`vii
`
`| am particularly grateful to the
`organization and technical accuracy of the text.
`management of Exar Integrated Systems, Inc., for providing me the time to work
`on this book, and to Ms. Sue Wooldridge who has patiently typed and retyped the
`draft of the manuscript several times over.
`
`ALAN B. GREBENE
`
`Saratoga, California
`August 1983
`
`Samsung, EX1012, p. 6
`
`Samsung, EX1012, p. 6
`
`

`

`CONTENTS
`
`CHAPTER 1.
`
`INTEGRATED-CIRCUIT FABRICATION
`
`1
`
`1.1
`
`1.2
`1.3
`1.4
`1.5
`1.6
`1.7.
`1.8
`
`4
`
`The Planar Process
`
`1
`
`Electrical Resistivity of Silicon
`Solid-State Diffusion
`5
`Epitaxial Deposition
`12
`Oxidation of Silicon
`14
`Photomasking
`17
`20
`Ion Implantation
`Thin-Film Processes
`
`22
`
`Bipolar Integrated-Circuit Fabrication Steps
`1.9
`1.10 Modifications of Basic Process
`31
`
`26
`
`1.11 Assembly and Packaging
`1.12
`Integrated-Circuit Packages
`1.13 Testing of Integrated Circuits
`1.14 Reliability Considerations
`47
`
`38
`
`41
`46
`
`CHAPTER 2. ACTIVE DEVICES IN INTEGRATED CIRCUITS
`
`53
`
`2.1
`2.2
`2.3.
`2.4
`2.5
`
`54
`npn Transistors
`npn Transistors for Special Applications
`pnp Transistors
`83
`95
`Junction Field-Effect Transistors
`MOSField-Effect Transistors
`106
`
`75
`
`PASSIVE COMPONENTS: DIODES, RESISTORS,
`CHAPTER 3.
`AND CAPACITORS
`
`Part |
`3.1
`
`Integrated Diodes
`Junction Diodes
`122
`
`121
`
`122
`
`Samsung, EX1012, p. 7
`
`Samsung, EX1012, p. 7
`
`

`

`Cee
`
`x
`
`CONTENTS
`
`3.2
`3.3
`
`126
`Schottky Diodes
`Zener Diodes
`130
`
`Integrated Resistors
`Part Il
`136
`Diffused Resistors
`3.4
`144
`Pinched Resistors
`3.5
`147
`Epitaxial Resistors
`3.6
`lon-Implanted Resistors
`3.7
`Thin-Film Resistors
`154
`3.8
`Trimming of Resistors
`155
`3.9
`Integrated Capacitors
`Part Ill
`Junction Capacitors
`160
`3.10
`3.11 MOS Capacitors
`164
`
`150
`
`CHAPTER 4. BIAS CIRCUITS
`
`4.1
`4.2.
`4.3
`4.4
`4.5
`4.6
`4.7
`4.8
`
`170
`
`Constant-Current Stages
`pnp Current Sources
`183
`Voltage-Controlled Current Sources
`Supply-Independent Biasing
`189
`Voltage Sources
`193
`197
`DC Level-Shift Stages
`Temperature-Independent Biasing
`Stabilization of Chip Temperature
`
`187
`
`204
`210
`
`135
`
`160
`
`169
`
`CHAPTER 5. BASIC GAIN STAGES
`
`215
`
`5.1
`5.2
`5.3
`
`215
`Differential Gain Stages
`Gain Stages with Active Loads
`Output Stages
`246
`
`233
`
`CHAPTER 6. ANALOG DESIGN WITH MOS TECHNOLOGY
`
`263
`
`264
`Basic Characteristics of MOS Transistors
`6.1.
`271
`Building Blocks of NMOS Analog Design
`6.2
`Analog Design with Depletion-Mode Load Devices
`6.3
`Analog Design with CMOS Technology
`290
`6.4
`6.5 MOS Voltage References
`299
`6.6 MOS Transistor as an Analog Switch
`
`303
`
`284
`
`Samsung, EX1012, p. 8
`
`Samsung, EX1012, p. 8
`
`

`

`CONTENTS
`
`.
`
`CHAPTER 7. OPERATIONAL AMPLIFIERS
`
`310
`Fundamentals of Operational Amplifiers
`7.1
`Circuit Configurations for Monolithic Operational Amplifiers
`7.2
`Frequency Compensation
`325
`7.3.
`Large-Signal Operation
`333
`7.4
`Input Stage Design
`339
`7.5
`Practical Op Amp Circuits
`7.6
`7.7 MOS Operational Amplifiers
`7.8
`Special-Purpose Op Amp
`375
`7.9
`Other Operational Amplifier-Based Circuits: Buffers and
`Comparators
`383
`
`350
`368
`
`xi
`
`309
`
`320
`
`CHAPTER 8 WIDEBAND AMPLIFIERS
`
`397
`
`398
`General Design Considerations
`8.1
`High-Frequency Transistors
`399
`8.2
`401
`High-Frequency Device Models
`8.3.
`Frequency Response of Single-Transistor Gain Stages
`8.4
`Compound Devices
`410
`8.5
`Neutralization of Collector-Base Capacitance
`8.6
`Amplifier Circuits Using Local Feedback
`417
`8.7
`Amplifier Circuits Using Overall Feedback
`423
`8.8
`Dual-Loop Feedback Amplifiers
`429
`8.9
`8.10 Root-Locus Techniques
`433
`8.11 Current Amplifiers: The Gilbert Gain Cell
`8.12 Electronic Gain Control
`443
`
`437
`
`415
`
`403
`
`CHAPTER 9. ANALOG MULTIPLIERS AND MODULATORS
`
`451
`
`9.1
`9.2
`9.3.
`9.4
`9.5
`9.6
`9.7.
`9.8
`
`451
`
`A Classification of Modulators and Multipliers
`Properties of an Analog Multiplier
`452
`Applications of an Analog Multiplier
`454
`Variable-Transconductance Multiplier
`456
`Four-Quadrant Multipliers with Wide Dynamic Range
`Practical Analog Multiplier Circuits
`462
`Balanced Modulators
`469
`Applications of Balanced Modulators
`
`472
`
`CHAPTER 10. VOLTAGE REGULATORS
`
`Part!
`
`Series Regulators
`
`10.1.
`
`Fundamentals of Series Regulators
`
`482
`
`459
`
`481
`
`482
`
`Samsung, EX1012, p. 9
`
`Samsung, EX1012, p. 9
`
`

`

`
`
`CONTENTS
`
`xii
`
`10.2
`
`10.3.
`10.4
`10.5
`
`Protection Circuits
`
`489
`
`497
`Practical Series Regulator Circuits
`509
`Layout Considerations for Power Circuits
`Failure Mechanisms in Power Devices
`512
`
`Part II
`
`Switching Regulators
`
`514
`Fundamentals of Switching Regulators
`10.6
`10.7 Modes of Operation with Inductive Output Circuits
`10.8
`Efficiency Considerations
`527
`10.9
`Practical Switching Regulator Circuits
`
`528
`
`521
`
`CHAPTER 11.
`
`INTEGRATED-CIRCUIT OSCILLATORS AND TIMERS
`
`Part!
`
`Integrated-Circuit Oscillators
`
`541
`
`An Overview of Oscillator Types
`11.1
`Tuned Oscillator Circuits
`543
`11.2
`Relaxation Oscillators
`556
`11.3.
`571
`Emitter-Coupled Multivibrators
`11.4
`581
`CMOSRelaxation Oscillators
`11.5
`Limitations of Relaxation Oscillators
`11.6
`11.7. Monolithic Wave-Shaping Techniques
`
`586
`591
`
`Part II
`
`Integrated-Circuit Timers
`
`Fundamentals of Integrated-Circuit Timers
`11.8
`One-Shot Timers
`600
`11.9
`11.10 Timer/Counter Circuits
`
`609
`
`599
`
`Part II! Frequency-to-Voltage and Voltage-to-Frequency Converters
`
`11.11 Voltage-to-Frequency Converters
`11.12 Frequency-to-Voltage Converters
`
`615
`622
`
`CHAPTER 12.
`
`PHASE-LOCKED-LOOP CIRCUITS
`
`Part!
`
`Fundamentals of Phase-Locked Loops
`
`12.1
`12.2
`12.3.
`12.4
`
`Principle of Operation of a PLL System 628
`PLL in Locked Condition
`635
`Effects of Loop Filter and Loop Gain on PLL Performance
`Applications of Phase-Locked Loops
`647
`
`637
`
`Part II
`12.5
`
`Building Blocks of Monolithic Phase-Locked-Loop Circuits
`Phase Detectors
`657
`
`514
`
`541
`
`541
`
`599
`
`615
`
`627
`
`628
`
`657
`
`Samsung, EX1012, p. 10
`
`Samsung, EX1012, p. 10
`
`

`

`CONTENTS
`
`668
`Voltage-Controlled Oscillators
`12.6
`12.7 Monolithic PLL Design Example
`673
`
`CHAPTER 13.
`
`INTEGRATED-CIRCUIT FILTERS
`
`Part!
`
`A Review of Filter Characteristics
`
`xiti
`
`679
`
`681
`
`681
`Basic Filter Specifications
`13.1
`A Review of Basic Filter Types
`13.2.
`Biquadratic Filter Function
`687
`13.3.
`
`13.4—Sensitivity Considerations 696
`13.5
`Analog Sampled-Data Filters
`698
`
`684
`
`Part Il Switched-Capacitor Filters
`
`703
`
`Fundamentals of Switched-Capacitor Circuits
`13.6
`Characteristics of MOS Circuit Elements
`712
`13.7.
`Effects of Parasitic Capacitances
`719
`13.8
`Practical Design Constraints
`725
`13.9
`13.10 Second-Order Filter Configurations
`13.11 Higher Order Filters
`739
`13,12 Applications and Limitations of Switched-Capacitor Filters
`
`727
`
`703
`
`CHAPTER 14. DATA CONVERSION CIRCUITS:
`DIGITAL-TO-ANALOG CONVERTERS
`
`750
`
`753
`
`14.1
`754
`Principles of D/A Conversion
`14.2
`757
`Basic D/A Converter Circuits
`14.3.
`Definitions of D/A Converter Terms
`14.4
`D/A Converter Architecture
`770
`14.5
`Current Switches
`780
`14.6
`Resistor and Capacitor Networks
`14.7
`Voltage References
`790
`14.8
`791
` Biasing of Current Sources
`14.9
`Effects of Device Mismatches
`795
`14.10 Accuracy Considerations
`799
`802
`14.11 Monolithic Design Examples
`14.12 Ultraprecision D/A Converter Circuits
`
`785
`
`764
`
`817
`
`CHAPTER 15. DATA CONVERSION CIRCUITS:
`ANALOG-TO-DIGITAL CONVERTERS
`
`825
`
`15.1
`15.2
`
`Fundamentals of A/D Conversion
` Integrating-Type A/D Converters
`
`827
`835
`
`Samsung, EX1012, p. 11
`
`Samsung, EX1012, p. 11
`
`

`

`xiv
`
`CONTENTS
`
`15.3
`15.4
`15.5
`15.6
`15.7
`15.8
`15.9
`
`846
`Digital-Ramp-Type A/D Converters
`847
`Successive-Approximation A/D Converters
`Successive-Approximation Converters Using MOS Technology
`Parallel A/D Converters
`865
`871
`Other High-Speed A/D Conversion Techniques
`Nonlinear A/D Converters for Telecommunications
`An Overview of A/D Converter Techniques
`876
`
`873
`
`INDEX
`
`852
`
`881
`
`Samsung, EX1012, p. 12
`
`Samsung, EX1012, p. 12
`
`

`

`514
`
`VOLTAGE REGULATORS
`
`The thickness of a metal interconnection layer may be reduced at the points
`where the metal trace is forced to run over oxide steps on the chip surface, due
`to different mask layers. This results in a reduction of the conductor cross
`section, and correspondingly,
`is an increase of the current density within the
`metal trace at these localized spots. In order to avoid potential metal-migration
`problemsat such points within the circuit, the high-currentinterconnection paths
`should not be routed over such oxidesteps.
`
`PART Il: SWITCHING REGULATORS
`
`10.6.
`
`FUNDAMENTALS OF SWITCHING REGULATORS
`
`The switching regulators, which are also called switch-mode regulators, find a
`wide range of applications in power supply design where high power and high
`efficiency are important. The principle of operation of a switching regulator
`differs significantly from that of a conventionalseries regulator circuit discussed
`in the first part of this chapter. In the caseof series regulators, the pass transistor
`is operatedinits linear region to provide a controlled voltage drop acrossit with
`a steady de currentflow. In the case of switching regulators, the pass transistor
`is used as a “controlled switch” and is operated at either the cutoff or the
`saturated state. In this manner, the poweris transmitted across the pass device
`in discrete current pulses, rather than as a steady current flow.
`The most important advantage of switching regulators over the conventional
`series regulatorsis their greater efficiency, since the pass device is operated as
`a low-impedance switch. Whenthe pass device is at cutoff, there is no current
`throughit, thus it dissipates no power. When the pass deviceis in saturation, it
`is nearly a short circuit with negligible voltage drop acrossit, thus it dissipates
`only a small amount of average power, provided that it can handle the peak
`current loads. In either case, very little power is wasted in the regulator and pass
`devices, and almostall the power is transferred to the load. In this manner, a
`very high degree of regulator efficiency is achieved, typically in the range of
`70-90%, relatively independent of the input-output voltage differentials. The
`efficiency of switching regulators is particularly apparent when there is a large
`input-output voltage difference across the regulator. For example, if one consid-
`ers the case of a regulator operating with a 28-V input and delivering a 5-V
`outputat 1-A current, a conventionalseries regulator would require a drop of 23
`V across the series pass transistor. Thus, a total of 23 W of poweris wasted in
`the regulator, resulting in an overall regulator efficiency of approximately 18%
`[see Eq. (10.12)]. As will be described in later sections, a switching regulator
`can be readily designed to perform the same function with greater than 75%
`efficiency under similar operating conditions.
`
`Samsung, EX1012, p. 13
`
`Samsung, EX1012, p. 13
`
`

`

`10.6
`
`FUNDAMENTALS OF SWITCHING REGULATORS
`
`515
`
`Another important advantage of the switching regulator circuit is its ver-
`satility. It can provide output voltages which can be less than, greater than, or
`of opposite polarity to the input voltage, as determined by the mode of operation
`of the circuit. In this manner, one can step up, step down, orinvert the polarity
`of an input voltage to generate any arbitary set of dc voltages within a system.
`Switching regulators also have some drawbacks. They are more complex and
`require external components such as inductors or transformers. They generate
`more noise and output ripple than conventionalseries regulators, and are slower
`respondingto transient load changes. Onearea of caution, when using switching
`regulators, is the generation of electromagnetic and radio-frequency interference
`(RFI). This interference problem is usually solved by the use of feedthrough
`low-passfilters isolating the powerlines into the regulator, and by using ground
`shields around the regulator to suppress the interference. However, even with
`these precautions, switching regulator circuits are not recommended for power-
`ing very-low-level signal-processing circuitry, where noise characteristics are
`very critical.
`A switching-regulator power supply system is made upofthree basic blocks:
`(1) the switching element which is normally a power transistor, (2) control
`circuitry which sets the duty cycle (i.e., on-off time) of the switching element,
`and (3) outputcircuitry which converts the pulsed input powerto a steady output
`power flow. The switching-regulator power supplies are classified into three
`categories, depending on the type of output circuitry used. These classes are
`
`1. Single-ended inductorcircuits.
`2. Diode—capacitor circuits.
`3. Transformer-coupled circuits.
`
`Figure 10.22 shows the three basic configurations for the output circuitry of
`single-ended inductor-type switching regulators. These are among the most
`frequently used output circuit configurations, since they are by far the easiest to
`design and control for medium- and high-current applications.
`using
`Figure
`10.23
`shows
`the
`switching
`regulator
`configurations
`diode—capacitor circuits. These circuits are typically used for very-low-current
`applications, and primarily as voltage multipliers, to increase the voltage level
`available from a low-voltage battery. Examples of such applications are the
`generation of voltage drive for LCD displays in watch circuits from a 1.5- or 3-V
`battery and in low-voltage hearing aid amplifiers.
`Figure 10.24 shows the two basic configurations of transformer-coupled
`output circuits. The circuit of Figure 10.24a is the so-called push-pull circuit
`used in conventional dc-to-de converters, with each switch controlled for O-45%
`duty cycle modulation. The configuration of Figure 10.245 is the so-called
`single-ended flyback converter, whichis useful at low- to medium-currentloads.
`The design of power supply systems using discrete circuits, and the various
`types of output circuitry shownin Figures 10.22 through 10.24, are well covered
`
`Samsung, EX1012, p. 14
`
`Samsung, EX1012, p. 14
`
`

`

`516
`
`VOLTAGE REGULATORS
`
`{e)
`
`Inductive switching regulator configurations:
`FIGURE 10.22.
`(c) polarity-inverting.
`
`(4) Step-down;
`
`(b) step-up;
`
`in the literature.” In the following discussions, we will primarily focus on
`switching regulator circuits using the single-ended inductor-type output circuit
`shown in Figure 10.22, since they represent by far the most commonand general
`categories of application, and are especially suited to high-currentapplications.
`The control circuitry section of a switching regulator system, which controls
`the on-off duty cycle of the switch transistors, can be readily integrated in a
`monolithic IC form. In many cases, the switching transistors up to 1-A current
`rating can also be incorporated into the monolithic chip. If higher powerlevels
`are required, the switch transistor on the chip is used as a drive for an external
`high-current switch.
`Figure 10.25 shows a simplified block diagram of a typical switching regu-
`lator integrated circuit used in conjunction with the single-ended inductor
`configuration of Figure 10.22a. The circuit generates a stream of pulses which
`turn switch S; on and off. The output de level is sensed through the sampling
`resistors R; and R, and compared against an internal voltage reference Vis, and
`the on-off time on the duty cycle of the switch 5, is varied accordingly to keep
`the output voltage constant under changing load conditions.
`
`Samsung, EX1012, p. 15
`
`Samsung, EX1012, p. 15
`
`

`

`Vin ? Vo
`
`| Vin | >|Vo|
`
`+
`
`Vin < Vo C2 Vz
`51
`
`FIGURE 10.23 Diode—capacitor-type switching regulator configurations: (a) Step-down; (b) volt-
`age multiplier; (c) polarity inverting.
`
`fc)
`
`FIGURE 10.24. Transformer—coupled switching regulator
`(b) flyback.
`
`configurations:
`
`(a) Push-pull;
`
`517
`
`Samsung, EX1012, p. 16
`
`Samsung, EX1012, p. 16
`
`

`

`
`
`
`
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`Samsung, EX1012, p. 17
`
`Samsung, EX1012, p. 17
`
`
`
`
`

`

`10.6
`
`FUNDAMENTALS OF SWITCHING REGULATORS
`
`519
`
`Neglecting the current in the sampling resistors, the average or dc value of the
`output current 7, delivered to the load is proportional to the duty cycle of the
`power switch S,, as shown in Figure 10.26. If the sampled output voltage is
`lower than V,.;, the polarity of the comparator output signal causes the control
`logic to increase the duty cycle of S, and, thus, causes the output voltage level
`to increase until the equilibrium is reached such that the output voltage, scaled
`down by the sampling resistors,
`is equal to the internal reference voltage.
`Similarly, if the output load current J, is decreased, this would cause the output
`voltage to increase, which in turn would be sensed by the control circuitry and
`would reduce the duty cycle of the switch accordingly.
`
`Control of the Duty Cycle
`
`The switch duty cycle t,,/T can be controlled by pulse-width modulation methods
`at a fixed frequency, or by fixing the on or off time and controlling the fre-
`quency. The relative merits and disadvantages of these techniques are briefly
`examined below.
`
`In this type of a switching
`Fixed-Frequency, Variable-Duty Cycle Operation.
`regulator, the operating frequencyis fixed, and the duty cycle ofthe pulsetrain
`is varied to change the average power. This method is often referred to as
`
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`
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`0.8
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`a
`
`FIGURE 10.26. Output load current /, as a function of switch duty cycle.
`
`Samsung, EX1012, p. 18
`
`Samsung, EX1012, p. 18
`
`

`

`520
`
`VOLTAGE REGULATORS
`
`pulse-width modulation (PWM). The fixed-frequency concept is particularly
`advantageous for systems employing transformer-coupled output stages. The
`fixed-frequency aspect enables the efficient design of the associated magnetics,
`In addition, filtering or shielding the surroundings from the radio-frequency or
`electromagnetic interference generated by the regulator is somewhat simplified
`because of the fixed frequency of switching. Because of these features,
`the
`majority of the monolithic switching regulators utilize the fixed-frequency,
`variable-duty-cycle contro] method.
`
`In this method, the switch
`Fixed-On-Time, Variable-Frequency Operation.
`has a fixed or predeterminedon time; and the duty cycle is varied by varying the
`frequency orrepetition rate of the control pulses. This method provides ease of
`design in voltage conversion applications using the single-ended inductive out-
`put circuit configurations of Figure 10.22, and simplifies design calculations for
`the inductor value. The fixed-on-time methodis also advantageousfor inductive
`output circuitry since a consistent amountof charge is developedin the inductor
`during the fixed on time. This eases the design or the selection of the inductor
`by defining the operating area to which the inductor is subjected undertransient
`load conditions. Figure 10.27a shows the typical frequency versus load current
`characteristics of a fixed-on-time, variable-frequency regulator, where the fre-
`quencyincreases linearly with increasing load.
`
`In this type of a voltage reg-
`Fixed-Off-Time, Variable-Frequency Operation.
`ulator, the de voltage at the output is varied by changing the on timef,, of the
`switch while maintaining a fixed off time Ty. As shown in Figure 10.275, the
`fixed-off-time switching regulator behaves in an opposite manner to the
`fixed-on-time system: as the load current increases, the on time becomes longer,
`thus decreasing the frequency. This approach is advantageousfor the design of
`switching regulators that will operate at a well-defined minimum frequency and
`low-tipple current under full-load conditions. One basic drawback of the
`
`
`
`LoadcurrentJ,
`
`
`
`LoadcurrentI,
`
`Frequency ——>
`
`Frequency ——>
`
`l/toit
`
`load current versus frequency characteristics of variable-frequency
`FIGURE 10.27. Typical
`switching regulators: (a) Hixed on time; (b) fixed off time.
`
`Samsung, EX1012, p. 19
`
`Samsung, EX1012, p. 19
`
`

`

`10.7. MODES OF OPERATION WITH INDUCTIVE OUTPUT CIRCUITS
`
`521
`
`fixed-off-time system is that the maximum current in the inductor, under tran-
`sient load conditions, is not well-defined. Thus, additional care is required to
`ensure that the saturation characteristics of the inductor are not exceeded.
`
`10.7. MODES OF OPERATION WITH INDUCTIVE OUTPUT CIRCUITS
`
`Two of the most important advantages of switched regulators are their high
`efficiency and their ability to step up, step down, or change polarity of an input
`voltage. These basic features can be best understood by examing the voltage and
`current waveformsat the output of the regulator. In this section, some of the
`waveforms and key design equations associated with the inductive output cir-
`cuits of Figure 10.22 will be examined for various modes of operation under
`steady-state load conditions. The rigorous derivations of the circuit equations
`associated with the basic current and voltage waveformsis straightforward and
`is available in the literature.“*!Therefore, for the sake of brevity, rigorous
`derivations will be omitted and only their conclusions will be presented.
`
`Step-Down Operation
`
`In the step-down modeof operation, the switching regulator produces an output
`de voltage V, which is /owerthan the input voltage V;,. Figure 10.28 shows the
`basic voltage and current waveforms associated with the circuit under steady-
`state operation. The switch 5; is assumed to have a voltage drop of V,in its on
`condition, and the diode D, has a forward drop of Vp whenit is conducting.
`When S; is closed or on, D, is off and the current /, in the inductor rises
`linearly from zero to its peak value /,,, with the slope
`
`dl,—Vi Vin — Vea — Vo
`
`==SS 10.22
`ad
`ob
`Ls
`(
`)
`
`Atthe end of the switch on-time f,,, this current reachesits maximum valueJ,,.
`When 5S; is off, the inductor generates the necessary voltage to forward bias D,
`and keep /; from changing instantaneously. During this portion of the cycle, I;
`linearly decays to zero at the end of the off time Zor.
`At steady state, average, or dc current through Co is zero; therefore, the
`output de current J, is equal to the average value of J, or
`I
`I, — (av = =
`From the waveforms shownin Figure 10.28, the switch on and off times can
`be related to the voltage levels at the input and output of the circuit as
`
`(10.23)
`
`
`Cort
`
`ton _ __Vo + Vp= 10.24
`Vi, ~ Va ~ Vo
`‘
`
`)
`
`Samsung, EX1012, p. 20
`
`Samsung, EX1012, p. 20
`
`

`

`522
`
`VOLTAGE REGULATORS
`
`
`
`
`
`
`
`
`
` Vo —- Vok
`
`
`
`
`
`
`FIGURE 10.28, Voltage and current waveforms in step-down mode.
`
`or, V, can be related to the rest of the voltages as
`ton
`to
`(10.25)
`Vo = BW= Von) ~ FVD
`Assuming the ideal case where both the saturation and the diode voltages are
`zero or negligible, this reduces to
`(Voidea! = ad
`
`(10.26)
`
`fo
`
`Samsung, EX1012, p. 21
`
`Samsung, EX1012, p. 21
`
`

`

`10.7 MODES OF OPERATION WITH INDUCTIVE OUTPUT CIRCUITS
`
`523
`
`Equation (10.26) implies that, ideally, the switching regulator in its step-
`down mode provides a downscaling of the input voltage by a scale factor equal
`to the duty cycle of the switch transistor.
`Another important parameter of the step-down regulator is the peak-to-peak
`output ripple voltage (AV,),,. Assuming that Co is sufficiently large so that the
`ripple voltage is much lower than the average or dc valueof the output, (AV.)pp
`can be expressed as
`
`(10.27)
`(AVo)pp ~ ac, + tor) = Bo
`where f = 1/T is the frequency or the repetition rate at which the switch opens
`and closes.
`
`i
`
`I
`
`Step-Up Operation
`
`In the step-up mode, the switching regulator produces an output dc voltage V,
`which is higher than V,,. The circuit configuration for this mode of operation,
`along with the associated voltage and current waveforms, is shown in Figure
`10.29 under steady-state operation.
`With reference to Figure 10.29, the operation of the circuit can be sum-
`marized as follows. Assuming that 5, is open and closes at the moment J, = 0,
`the current in the inductorrises linearly from zero to a peak value J, during fo.
`At the end of the on time, S, is opened. Since /, cannot change instantaneously,
`the inductor generates the necessary voltage at node A to forward bias D, and
`keep the current continuous. Duringthe off time, J, decays linearly, and reaches
`zero at fo. Then S$, closes again, and the cycle repeats itself. While D,
`is
`conducting, it supplies current both to the load and to the holding capacitor Co;
`and when D, is nonconducting, the output current is drawn from Co. Note that
`at steady state, the averageor de current through D, is equal to the output or load
`current J, and the net charge supplied to Cy, per cycle of operation, is zero.
`The peak current /,, is related to the steady-state output current as
`
`Vp F Vo ~~ Via
`Vin ~~ Visat
`°
`fn
`and the on-off times of the switch, necessary for /, to ramp from zero to J, and
`back to zero, are related as
`
`(10.28)
`
`= 21,
`
`bon — Vo + Vo — Vin
`lore
`Vin ~ Vsat
`Solving Eq. (10.29) for V,, one obtains
`
`ton
`T
`V, = Via Vout — Vp
`lott
`Fog
`
`10.29
`
`)
`
`(
`
`(10.30)
`
`where T (= tor + fon) is the period of one full cycle of operation. In the idealized
`
`Samsung, EX1012, p. 22
`
`Samsung, EX1012, p. 22
`
`

`

`VOLTAGE REGULATORS
`
`
`
`
`
`
`
`
`
`
`
`
`
`524
`
`Va
`
`Is
`
`Ip
`
`In
`
`Tok —
`
`Vo
`
`
`
`FIGURE 10.29. Voltage and current waveforms in step-up mode.
`
`case, where the diode drop and V,of the switch are negligible, Eq. (10.30)
`reduces to
`
`(V, idea! = Vin —_r
`lotr
`
`(10.31)
`
`or, in other words, the step-up modeof operation results in up-scaling the input
`voltage by the ratio T/tos;.
`
`Samsung, EX1012, p. 23
`
`Samsung, EX1012, p. 23
`
`

`

`10.7 MODES OF OPERATION WITH INDUCTIVE OUTPUT CIRCUITS
`
`525
`
`The peak-to peak output ripple voltage can be expressed as‘
`(Ix _ i,)P for
`2px
`Co
`
`(AV,)op ~
`with the assumption that (AV,),) < Vo.
`
`(10.32)
`
`Polar