United States Patent
`Hulkko et al.
`
`[19]
`
`11111111111111111111111111111111111111111111
`US005734683A
`5,734,683
`[ill Patent Number:
`[45] Date of Patent: Mar. 31, 1998
`
`[54] DEMODULATION OF AN INTERMEDIATE
`FREQUENCY SIGNAL BY A SIGMA-DELTA
`CONVERTER
`
`[75]
`
`Inventors: Jaakko A. Hulkko; Veijo L. H.
`Kontas, both of Oulu, Finland; Lauri
`T. Siren, San Diego, Calif.
`
`[73] Assignee: Nokia Mobile Phones Limited. Salo,
`Finland
`
`7/1986 European Pat. Off. .
`0 186 151 A3
`0335 037 10/1989 European Pat. Off. .
`0 461 720 Al 12/1991 European Pat. Off. .
`0461 720 12/1991 European Pat Off. .
`8/1992 European Pat. Off. .
`0499 827
`1/1991 United Kingdom .
`2233 518
`WO 89/07368
`8/1989 WIPO .
`7/1991 WIPO
`WO 91/10283
`
`OTHER PUBLICATIONS
`
`[21] Appl. No.: 624,113
`
`[22] Filed:
`
`Mar. 29, 1996
`
`Related U.S. Application Data
`
`[63] Continuation of Ser. No. 303,613, Sep. 9, 1994, abandoned.
`
`[30]
`
`Foreign Application Priority Data
`
`Sep. 10, 1993
`
`[Fl]
`
`Finland
`
` 933989
`
`[51] Int. C1.6
`[52] U.S. CL
`
` H04B 14/06; HO4L 27/22
` 375/316; 375/247; 375/328;
`375/329
` 375/240. 244,
`[58] Field of Search
`375/247, 316, 328-329, 332; 341/143
`
`[56]
`
`References Cited
`
`U.S. PATENT DOCUMENTS
`
`5,155,743
`5,179,380
`5,191,332
`5,198,817
`5,305,004
`
`10/1992 Jacobs
`1/1993 White
`............... ........ ......
`3/1993 Shieu
`3/1993 Walden et al.
`4/1994 Fattaruso
`
` 375/247
` 341/143
`341/143
` 341/143
` 341/143
`
`FOREIGN PATENT DOCUMENTS
`
`IFF Journal of Solid—State Circuits., vol. 26, No. 12, Dec.
`1991. New York US pp. 1951-1957, K. Lakshmikumar et
`al., `A Baseband Codec For Digital Cellular Telephony'.
`Signal Processing.. vol. 22, No. 2, Feb. 1991, Amsterdam
`NL, pp. 139-151, H.J. Dressler. `Interpolative Banolpass—A/
`D—Conversion'.
`The Design of Sigma—Delta Modulation, Analog—To—Digi-
`tal Converters. B. Boser et al , IFF.F Journal of Solid—State
`Circuits. vol. 23. No. 6. Dec. 1988, pp. 1298-1308.
`Digital Communication. E. Lee et al., Kluwer Academic
`Publishers, Boston, 1990, pp. 173-176.
`
`Primary Examiner—Young T. Tse
`Attorney, Agent, or Finn—Ferman & Green, LLP
`
`[57]
`
`ABSTRACT
`
`A sigma-delta signal converter is implemented using
`switched capacitor switching elements in which a first
`switch (31) serves as a mixer (11). The output of the mixer
`is directed to the second input of an adder (16). and its
`second input is the feedback signal (fl) of the sigma-delta
`signal converter, which is also directed into a base-
`frequency output signal through a decimator (14) and low-
`pass filtering (15).
`
`0168 220 1/1986 European Pat. Off. .
`
`16 Claims, 3 Drawing Sheets
`
`SIGMA DELTA CONVERTER
`
`11
`
`12
`
`13
`
`{
`MODULATOR
`
`AGC
`
`( 14
`
`DECIMATOR
`
`12
`
`( 13
`
`( 14
`
`AGC
`
`MODULATOR
`
`- 1Erd DECIMATOR
`
`40
`I
`
`15
`
`(
`POST
`FILTER
`
`15
`
`(
`POST
`FILTER
`
`( 10
`
` IF
`
`9
`
`L
`
`PHI3
`
`PHI4
`
`LO1
`
`LG Ex. 1004
`LG Electronics Inc. v. ParkerVision, Inc.
`IPR2022-00246
`Page 00001
`
`

`

`juajed °S°11
`
`8661 `I£ IBIAI
`
`£ Jo 1 ;aayS
`
`PRIOR ART
`FIG. 1
`
`L02
`
`LO 1
`
`PHI4
`
`PH I3
`
`PHI 2 PHI 1
`
`(8
`
`FILTER
`POST
`
`FILTER
`POST
`
`r 8
`
`DECIMATOR
`
`MODULATOR
`
`AGC
`
`(7
`
`6
`
`5
`
`L
`
`r
`
`3
`
`2
`
`DECIMATOR
`
`MODULATOR
`
`AGC
`
`7
`
`1
`6
`
`5
`
`3
`
`r
`
`2
`
`IF
`
`IF
`
`IPR2022-00246 Page 00002
`
`

`

`FIG.2
`
`LO1
`
`PHI4
`
`PHI3
`
`lualed 'Sit
`
`8661 `I£ Jell
`
`£ JO Z pays
`
`FILTER
`POST
`
`FILTER
`POST
`
`-1-111•-DECIMATOR
`
`MODULATOR
`
`AGC
`
`(1 4
`
` 1
`13
`
`-J
`
`11 712
`
`DECIMATOR
`
`MODULATOR
`
`AGC
`
`(15
`
`7-14
`
`1
`13
`
`SIGMA DELTA CONVERTER
`
`(40
`
`
`
`9
`
`IF
`
`IPR2022-00246 Page 00003
`
`

`

`U.S. Patent
`
`Mar. 31, 1998
`
`Sheet 3 of 3
`
`5,734,683
`
`MODULATOR, 13
`
`/ (11
`
`16
`
`(17
`
`18
`
`(19
`
`(20
`
`IN
`
`MIX
`
`.11
`
`110111••=.1111.
`
`INT1
`
`+
`
`I NT2
`
`COMP
`
` •-- OUT
`TO
`DECIMATOR,
`14
`
`L01
`
`f1
`
`MIXING
`ELEMENT,
`31
`
`i
`
`LO1
`.J—U—L
`
`INPUT
`
`I
`
`f2
`
`F1G.3
`
`38
`
`34 39
`
`--
`--Id-
`•--•
`_ (36 35 37
`.--...
`
`.--112-.N(.--4. c 30
`
`FIG.4
`
`/ AGC,
`12
`
`• OUTPUT
`
`--‘
`
`11, MIX
`
`32
`
`IPR2022-00246 Page 00004
`
`

`

`5,734,683
`
`2
`where 0)0 is the angle frequency of the carrier wave and PHI
`is a momentary phase modulation (QAM. MSK, QPSK,
`GMSK . . . ).
`Ideally the term cos(b) represents a clean, mixing oscil-
`5 lator frequency (LO):
`
`b=rt * cut * r (n=1, 2, 3, . . . )
`
`(3)
`
`10
`
`where Owl is the angular frequency of the oscillator of the
`mixer.
`In the ideal case the frequency and the phase of the
`oscillator are locked to the frequency and the phase of the
`carrier wave of the input signal (in). in these conditions
`ei0=n*oil, and the term 1/2 *cos(a—b) is reduced to '*cos
`(PHI). This base-frequency phase difference signal conveys
`15 the data symbols. Term Vi*cos(a+b) represents the compo-
`nent of the frequency spectrum on frequency 2*(o0.
`When prior art receive arrangements such as those
`described above are implemented by discrete components
`they require a very large area on the printed circuit board. In
`20 addition, as the signal entering the sigma-delta converter is
`a baseband signal the ac-coupled branches need very low
`high-pass corner frequencies with high time constants in
`order to accomplish de blocking. This means that it is not
`efficient for the arrangement to be powered down as often as
`25 would be ideal, as powering up again is slow and as a result
`the circuit cannot be powered down for short periods, the
`circuit therefore, consumes a large amount of power.
`
`30
`
`SUMMARY OF THE INVENTION
`In accordance with the present invention there is provided
`a receiver for receiving a modulated carrier signal
`comprising, a sigma-delta signal converter having at least
`one adder included in a feedback loop, characterised in that
`the arrangement comprises a time discrete sampling means
`35 for down converting the modulated carrier signal prior to the
`feedback loop.
`By down convening the carrier frequency signal using a
`time discrete sampling means a number of advantages are
`40 provided. Firstly, an expensive sinusoidal oscillator is no
`longer required with space and cost benefits. Secondly,
`although use of time discrete sampling means, rather than a
`pure sinusoidal local oscillator for down converting the IF
`signal means that mixing occurs at the frequency of sam-
`45 piing and also at harmonics of the sampling frequency this
`perceived disadvantage can be used to the system's advan-
`tage enabling samples to be taken using a local oscillator
`sampling at a subharmonic frequency of the carrier signal.
`Thus can also give important power savings.
`so One way in which the invention can be implemented is by
`using the input stage of a sigma-delta signal Converter
`having switched capacitor switching elements to implement
`the time discrete sampling means that acts as a mixer. The
`sigma-delta converter with the desired switched capacitor
`55 switching elements provided at the input stage may be
`implemented as an ASIC. The output of this mixer can then
`be directed to the first input of an adder included in the
`closed feedback signal loop of the sigma-delta converter.
`This adder comprises, as the second input, the feedback
`60 signal of the sigma-delta signal converter, which is also
`directed through a decimator and low-pass filter to provide
`(1)
`an output signal which is provided to the second input of the
`Equation (1) holds only if both products are pure cosine
`adder.
`signals.
`In circuits of embodiments of the invention the incoming
`If cos(a) now represents the modulated int-carrier wave 6
`5 modulated signal may be mixed into a base-band frequency
`then:
`signal or a frequency approaching the base-band frequency
`prior to entering the closed feedback loop.
`
`1
`DEMODULATION OF AN INTERMEDIATE
`FREQUENCY SIGNAL BY A SIGMA-DELTA
`CONVERTER
`
`This is a continuation of application Ser. No. 08/303,613
`filed on Sep. 9, 1994. now abandoned.
`
`BACKGROUND OF THE INVENTION
`
`The invention relates to a receive arrangement for receiv-
`ing a modulated carrier signal.
`EP 0 461 720-Al describes a known receive arrangement
`for receiving a modulated carrier signal, comprising a mixer/
`demodulator driven with a sinusoidal oscillator of carrier
`frequency fc, at least one adder, a low-pass filter, a pulse
`shaper constituting a one-bit sigma-delta signal converter,
`all included in a closed signal loop, the pulse shaper being
`driven with sampling frequency fs, and further comprising a
`digital decimation filter. In this type of receive arrangement
`the modulated carrier signal is demodulated in the closed
`signal loop by the mixer/demodulator, the output signal of
`which is converted after passing through the low-pass filter
`into a digital signal by the sigma-delta converter.
`A typical prior art sigma-delta converter arrangement is
`described in greater detail with reference to FIG. 1 of the
`drawings.
`An incoming intermediate frequency IF carrier signal is
`provided to each branch of the receive arrangement. In each
`branch the incoming signal is filtered through bandpass filter
`1 and mixed to a baseband signal in a linear mixer 2 using
`a sinusoidal local oscillator signal LO1 at the IF frequency.
`A high time constant capacitor 3 is provided on each of the
`incoming signal branches to remove direct currents from the
`baseband signal. The gain of the circuit is controlled through
`Automatic Gain Controllers (AGC) 5 and the baseband
`signals are converted to digital signals in modulators 6. After
`modulation the signals pass through respective decimators 7
`and post filters 8 to remove spurious signals created by the
`decimators. The specific details of the local oscillator fre-
`quency and phase shifts of the particular arrangement illus-
`trated in FIG. 1 are as follows:
`PHI1=+45°
`PHI2=-45°
`PHL3=PH144°
`LO1=lF
`LO2=aversampling frequency
`The demodulation, i.e., the mixing of the intermediate
`frequency (bandpass-filtered) int-signal down to the base
`frequency is traditionally based on the use of a multiplier.
`Thus the modulated IF-signal is multiplied by the sinusoidal
`oscillator signal (LO1). In the synchronous demodulation
`the frequency and the phase of the oscillator are locked to
`the carrier wave with the aid of a phase-locked loop (PLL),
`for instance. The frequency spectrum of the mixed product
`consists of the desired base-frequency component and the
`component spectrums which are removed by low-pass fil-
`tering prior to entering the sigma-delta converter. Such a
`mixing process may be described by the following trigono-
`metric equation:
`
`cos(a) * cos(b)/2 * cos(a—b)+Y2 * cos(a+b)
`
`a=r.o0 • t +PHI
`
`(2)
`
`IPR2022-00246 Page 00005
`
`

`

`5,734,683
`
`3
`Sigma-delta converters are traditionally used in convert-
`ing base-band signals, However, in accordance with the
`invention, they can now be adapted for converting interme-
`diate frequency signals directly.
`Difficult ac-coupling problems, control and high-pass
`filtering problems are solved by circuit arrangements of
`embodiments of the invention. Similarly, power consump-
`tion can be decreased by shortening the time for switching
`the receiver on from stand-by to the active State. This
`enables the circuit to be powered down when not in use for
`shorter periods than would conventionally be possible as a
`capacitance with a lower time constant is adequate for dc
`blocking.
`An additional advantage can be gained by using switched
`capacitors to provide some of the automatic gain control of
`the circuit. This means that the number of AGC-circuits
`required by the receive arrangement as a whole can be
`decreased. With embodiments of the invention part of the
`necessary filtering can also be provided without the need for
`additional filters between the mixing and a—d converting
`stages by utilising a digital filter already present in the
`sigma-delta converter.
`Embodiments of the invention can be utilized advanta-
`geously in, for example, radio telephones.
`
`BRIEF DESCRIPTION OF THE DRAWINGS
`Embodiments of the invention will now be described in
`greater detail with reference to FIGS. 2 to 4 of the drawings
`of which:
`FIG. 1 is a block diagram of a conventional sigma-delta
`converter of the prior art;
`FIG. 2 is a block diagram of a sigma-delta converter
`included in a receive arrangement according to one embodi-
`ment of the invention;
`FIG. 3 is a schematic representation of an embodiment of
`a receive arrangement including the sigma-delta converter of
`FIG. 2 operating with a local oscillator at or near the carrier
`frequency of the incoming signal; and
`FIG. 4 is a schematic representation of a switched capaci-
`tor switching element suitable for implementing the mixing
`and automatic gain control functions of the embodiment of
`FIG. 2.
`
`DETAILED DESCRIPTION OF THE
`INVENTION
`The receive arrangement of an embodiment of the inven-
`tion is illustrated in FIG. 2 using a sigma-delta analog-digital
`converter with a large dynamic input range in which a mixer
`11 is implemented using switched capacitor switching ele-
`ments 30-39 illustrated in FIG. 4. The receive arrangement
`of this embodiment receives radio signals for a radio tele-
`phone 40. The switched capacitor switching elements pro-
`viding the mixing function of the mixer 11 are driven by a
`square wave local oscillator signal (LO1) at (or near) the
`frequency of the IF signal. Both the mixer and the local
`oscillator signal are digital. Switched capacitor switching
`elements are also provided to implement an automatic gain
`controller (AGC) 12 providing an automatic gain control
`function for the circuit. The receive arrangement includes a
`bandpass filter 10, and each branch further includes a
`modulator 13 that converts signals from analog signals to
`digital signals, a decimator 14 and a post filter 15 which
`perform the same functions as the correspondingly named
`portions of the prior art receive arrangement illustrated in
`FIG. 1. The prefiltering of the signal (after modulation) can
`
`5
`
`4
`be designed to freely correspond to the design demands of
`the respective circuit and the dc-deviation of the sigma-delta
`converter can be corrected using the internal, digital correc-
`tion of deviations.
`The phase and frequency details for the local oscillator
`signals provided to the respective branches are as follows:
`PHI3=i-45°
`PHI4=-45°
`LO1=IF
`10 A base-frequency output signal is obtained from the
`modulator after the decimator and the low-pass filter which
`can be processed to retrieve the modulating information.
`Because the signal entering the sigma-delta converter
`arrangement is an IF signal, only a short time-constant
`15 capacitor 9 is necessary for preventing do signals from
`transferring to the sigma-delta converter. This means that the
`device can be powered up and down more quickly and as
`less power is required to power up. short term power downs
`are practical making the arrangement more power efficient
`20 than conventional receive arrangements.
`The mixer 11 and modulator 13 are described in greater
`detail with reference to FIG. 3. The modulated reception
`signal (in), for instance a bandpass-filtered int-signal from
`the RF-part of the radio telephone 40, is directed to the mixer
`25 11 (mix) to which the local oscillator (LO1) signal is also
`applied. LO1 may be on or around the carrier frequency of
`the received signal (in) or a subharmonic of that frequency.
`The output of mixer 11 is directed to a first adder 16. the
`second input of which is a feedback signal fl. The output of
`30 the first adder 16 is directed to an integrator 17. The output
`of the integrator 17 is directed to a second adder 18, the
`second input of which is a feedback signal f2. The output of
`the second adder 18 is directed to a second integrator 19 and
`further to a comparator 20. The output signal (out) of the
`35 comparator 20 is further directed to the decimator 14 the
`(low-pass) post filter 15 for filtering out unwanted signals
`resulting from mixing of LO1 and the carrier signal.
`The output signal provides a base-frequency signal which
`can be processed using digital signal processing means, for
`ao instance. The output signal (out) is coupled to the first and
`second adders 16 (fl) and 18 (f2) in respective feedback
`branches.
`The second adder 18, the second integrator 19 (int2) and
`the comparator 20 (cmp) provide a second closed feedback
`45 loop in the circuit. Those skilled in the art know the basic
`idea of the sigma-delta converter, therefore it is not
`described in more detail in this connection. More detailed
`discussion can be found in the articles: The Design of
`Sigma-Delta Modulation Analog-to-Digital Converters.
`50 Bernhard E. Boser, Bruce A. Wooley, JFFF Journal of
`Solid-State Circuits, Vol. 23 No. 6 December 1988 and
`Oversampling Delta-Sigma Data Converters. Theory,
`Design and Simulation J. C. Candy and G. C. Temes IFEF
`press 1992 both incorporated herein by reference.
`Typically in the prior art an analog bandpass filter is
`provided prior to entering the modulation stage of the
`sigma-delta converter to remove unwanted signals resulting
`from mixing. In the present case, however, the digital
`filtering function of the sigma-delta converter itself can be
`60 used to remove the unwanted signals.
`FIG. 4 shows the input stage of the receive arrangement
`of the embodiment of FIG. 2 showing switched capacitor
`switching elements of the mixer 11 and the AGC 12 in
`greater detail. A first capacitor 30 is used to sample end hold
`65 the incoming signal, First switches 31, 32 are closed to
`provide a sample to the first capacitor 30. Once the input
`signal has been sampled, a third switch 33 is closed to
`
`55
`
`IPR2022-00246 Page 00006
`
`

`

`5.734,683
`
`6
`oscillator for driving the mixer 11 does not coincide exactly
`with the IF carrier signal. Another example is when it is
`desirable to provide four times oversampling. Under these
`circumstances the subsequent digital mixer can be imple-
`5 mented more easily when the signals are at a frequency M
`offset from the baseband. Typically a signal within 1 MHz
`of the carrier frequency is acceptable for down converting
`the incoming signal.
`In a conventional arrangement. if the mixing frequency of
`io a sinusoidal local oscillator differs from the IF carrier signal
`by M. the term cos(b) of equation (1) is solved by the
`following formula:
`
`b=n*oh*t+Acoi't
`
`(5)
`
`5
`transfer the charge on the first capacitor 30 to the output.
`Second and third (and possibly further) capacitors 34, 35 are
`provided in parallel with the first capacitor 30. These are
`each controllably connected to the input and output through
`a pair of switches 36, 37; 38, 39. By closing the appropriate
`switches and adding parallel capacitance from one or more
`of the second and third capacitors 34, 35 the signal transfer
`ratio can be changed. The switches are under the control of
`an external cpu and can be used to replace automatic gain
`control steps of the circuit as a whole. In this way amplifi-
`cation steps can be included in the sigma-delta modulator by
`altering the ratios of the input capacitances.
`The mixer 11 can be considered as a sample and a hold
`circuit that samples the input signal in synchronization with
`the oscillator and directs the samples to the output as a signal
`which remains constant for the period of the sampling
`interval. The oscillator signal (LO1) is therefore represented
`by a square wave with a base frequency of n*ail. Instead of
`the term cos(bcos(n*cal*t) of equation (1), the following
`series of odd harmonics is obtained:
`
`cos(n*col*t)+1/4* cos (3*n*col*04-1,6* cos (5*n *col st)+. . .
`(4)
`The cosine terms of a higher order are mixed in the mixer
`(11) with the input signal (in) producing sum and difference
`components of the frequencies to the spectrum of the output 25
`signal of the mixer (11). In the prior art, instead, all input
`signals on a higher frequency than the base frequency are
`filtered by a filter before entering the mixing stage of the
`sigma-delta converter.
`It is preferable to use the first switch 31 of the switched 30
`capacitor switching element as the mixing element. In this
`case, signal bands around the multiples of the frequency of
`the local oscillator signal LO1 are folded onto the base
`frequency. The local oscillator base frequency or its subhar-
`monics can therefore be used to down convert the carrier
`signal to the base-band or a frequency approaching the
`base-band. The unwanted signals resulting from mixing
`using the local oscillator are removed by filtering.
`Referring back to FIG. 2, the inventive idea is realized in
`the circuit arrangement of this embodiment of the invention
`in accordance with which switched capacitor switching
`elements present in the input stage of a sigma-delta con-
`verter are used to implement the mixer 11 which directly
`demodulates the IF-signal into a base-frequency signal; in
`other words, the IF-signal and its multiples are folded on the
`base frequency. The first switch 31 in the input of stage of
`the sigma-delta converter is utilized here to serve as the
`mixer 11.
`The embodiment of FIGS. 2 to 4 can be implemented
`using a local oscillator having a frequency LO1 that is the
`same as or approaching the frequency of the IF carrier
`signal. Although it would generally be desirable for LO1 to
`have the same frequency as the incoming signal it may in
`many instances be desirable or merely practical to use a local
`oscillator frequency LO1 offset slightly from the frequency
`of the IF carrier signal, the mixing frequency could, for
`instance, be LO+M (where LO is the frequency of the IF
`carrier). In this case the signal (in) applied to the input is
`folded to the frequency of which is almost to the baseband
`frequency. If the modulated intermediate frequency (in) is
`1010 kHz, the mixing signal LO+M may be 900 kHz, for
`instance, whereby the demodulated signal is on the fre-
`quency of —110 kHz with respect to the baseband frequency.
`One example of when it may be practical to use a local
`oscillator with a frequency slightly offset from the IF carrier
`frequency is when a driver conveniently located in the
`sigma-delta converter for providing a square wave local
`
`15
`
`20
`
`where Cao is the angular frequency corresponding to fre-
`quency M.
`Although equation (1) deals with the mathematics of the
`prior art solution using a sinusoidal local oscillator, once the
`down converted signal generated by mixing using the time
`discrete sampler of embodiments of the present invention is
`filtered prior to entering the a—d modulation stage, it is
`effectively a pure cosine signal and equation (1) holds.
`The receive arrangement of FIGS. 2 to 4 can also be used
`to down convert the incoming signal to the baseband (or a
`frequency approaching the baseband) using a subharmonic
`of the carrier frequency. In these circumstances the phase
`and frequency details for the local oscillator signals pro-
`vided to the respective branches are as follows:
`PHI3---+45/N°
`PHI4=-45/N°
`LO1=IF/N
`In other respects the arrangement operates in the same
`manner as previously described.
`35 When the input signal (in) is branched into two different
`branches, it is possible to arrange the receive circuit arrange-
`ments of embodiments of this invention in each of the
`branches. The demodulation of an I/Q-modulated signal
`(I=in the phase, Q=in the phase shift of 90 degrees) may be
`ao implemented simply in this way using principles known per
`se and described by way of example in Digital
`Communication, Edward A. Lee, David G. Messershmitt,
`Kluwer Academic Publishers, Boston, 1990 incorporated
`herein by reference. The clocks of the modulators of both
`45 branches are synchronized.
`Those skilled in the art will notice that the circuit arrange-
`ments of embodiments of the invention are simple to imple-
`ment using relatively few circuit elements. They result in
`decreased power consumption and accelerated operation of
`50 the circuit (fast shifting from the stand-by state to the active
`operating state and vice versa) which is especially signifi-
`cant for radio telephones.
`The present invention includes any novel feature or
`combination of features disclosed herein either explicitly or
`55 any generalisation thereof irrespective of whether or not it
`relates to the claimed invention or mitigates any or all of the
`problems addressed.
`In view of the foregoing description it will be evident to
`a person skilled in the art that various modifications may be
`60 made within the scope of the invention.
`What is claimed is:
`1. A receiver for receiving a modulated carrier signal
`comprising, a sigma-delta signal converter having at least
`one summing node included in a feedback loop, character-
`65 ized in that the sigma-delta signal converter comprises a
`time discrete sampling means located at an input stage of the
`sigma-delta signal converter controlled by a square wave
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`local oscillator signal for down converting the received
`modulated carrier signal to a base-band signal, wherein the
`frequency of the square wave local oscillator signal is equal
`to one of the carrier signal frequency of the received
`modulated carrier signal plus an offset frequency Af, where
`Al is equal to one of a value of zero or a value greater than
`zero, or a subharmonic of the modulated carrier signal
`frequency.
`2. A receiver according to claim 1 wherein the time
`discrete sampling means comprises a switching member
`having an input node coupled to said received modulated
`carrier signal, and wherein said square wave local oscillator
`signal is coupled to said switching member for controlling
`the opening and closing of said switching member.
`3. A receiver according to claim 1 wherein the time
`discrete sampling means comprises switched capacitor
`switching elements.
`4. A receiver as set forth in claim 1, wherein said
`modulated carrier signal is divided into two signals having
`a predetermined phase relationship one to another, and
`wherein said receiver is comprised of first and second
`sigma-delta signal converters each having an input stage
`comprised of said time discrete sampling means that is
`controlled by said square wave local oscillator signal, each
`of said first and second sigma-delta signal converters being
`coupled to one of said two signals.
`5. A receiver as set forth in claim 1, wherein said receiver
`is a component part of a radio telephone.
`6. A circuit for receiving a modulated radio frequency
`(RF) carrier signal for down converting the received carrier
`signal to a base-band signal, the circuit comprising a sigma-
`delta signal converter circuit comprised of a switched
`capacitor input stage that is switched with local oscillator
`signal provided in the form of a non-sinusoidal switching
`signal having a frequency equal to or approximately equal to
`one of a frequency of said received carrier signal, or a
`sub-harmonic of said received carrier signal; said switched
`capacitor input stage outputting the base-band signal; a first
`summing node having a first input coupled to an output of
`said input stage; a first integrator coupled to an output of said
`first summing node; a second summing node having a first
`input coupled to an output of said first integrator; a second
`integrator coupled to an output of said second summing
`node; and a comparator coupled to an output of said second
`integrator; wherein an output of said comparator is coupled
`back to a second input of each of said first summing node
`and said second summing node; said sigma-delta signal
`converter circuit further comprising a decimator coupled to
`an output of said comparator and a filter coupled to an output
`of said decimator.
`7. A circuit according to claim 6, characterized in that the
`local oscillator signal has a frequency given by LO-FM,
`whereby the modulated carrier signal is folded on an inter-
`mediate frequency Al.
`8. A circuit according to claim 6, characterized in that said
`input stage is further comprised of a switched capacitor
`circuit that is controlled by said local oscillator signal.
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`9. A circuit according to claim 6, characterized in that said
`input stage comprises a switched capacitor circuit having at
`least one capacitor that is switchably coupled in parallel with
`said switched capacitor circuit for implementing an auto-
`5 matic gain control (AGC) function.
`10. A circuit as set forth in claim 6, wherein said modu-
`lated carrier signal is divided into two signals having a
`predetermined phase relationship one to another, and
`wherein said circuit is comprised of first and second sigma-
`10 delta signal converter circuits each having said switched
`capacitor input stage that is switched with said local oscil-
`lator signal, each of said first and second sigma-delta signal
`converter circuits being coupled to one of said two signals.
`11. A circuit as set forth in claim 6, wherein said circuit
`15 is a component part of a radio telephone.
`12. In a radio telephone, a method for converting a
`received modulated intermediate frequency (IF) signal to a
`base-band signal. comprising the steps of:
`applying the IF signal to an input stage of a sigma-delta
`converter circuit, the input stage comprising a switched
`capacitor circuit;
`mixing the IF signal to the base-band signal in the
`switched capacitor circuit by applying a switch control
`local oscillator (LO) signal having a frequency equal to
`one of the frequency of the received modulated IF
`signal plus or minus an offset frequency Al, where Af
`is equal to one of a value of zero or a value greater than
`zero, or a subharmonic of the received modulated IF
`signal; and
`demodulating and converting an output of the switched
`capacitor circuit to a digital signal by the sigma-delta
`converter circuit.
`13. A method as set forth in claim 12. wherein the step of
`applying applies the IF signal to input stages of at least two
`35 sigma-delta converter circuits, the input stage of each com-
`prising the switched capacitor circuit, wherein the step of
`mixing occurs in each sigma-delta converter circuit by
`applying to each input stage the switch control local oscil-
`lator (LO) signal, and wherein the LO signals have a
`4o predetermined phase relationship one to another.
`14. A method as set forth in claim 12, wherein the step of
`applying includes a preliminary step of bandpass filtering
`the IF signal.
`15. A method as set forth in claim 12, wherein the step of
`45 mixing includes a step of varying a signal transfer ratio of
`the switched capacitor circuit by switchably connecting one
`or more capacitances in parallel with the switched capacitor
`circuit
`16. A method as set forth in claim 12, wherein the step of
`so applying the switch control LO signal applies a nominally
`square wave signal having a period that defines a first phase
`and a second phase, and wherein the step of mixing includes
`the steps of sampling the applied IF signal during the first
`phase, and outputting the sampled IF signal during the
`55 second phase.
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