`
`11111111111111111111111111f1I111111111p111111111111 11110111111
`
`(12) United States Patent
`Mason
`
`(10) Patent No.:
`(45) Date of Patent:
`
`US 8,436,643 B2
`May 7, 2013
`
`(54)
`
`HIGH FREQUENCY SOLID STATE
`SWITCHING FOR IMPEDANCE MATCHING
`
`(75)
`
`Inventor: Christopher C. Mason, Fort Collins,
`CO (US)
`
`(73)
`
`Assignee: Advanced Energy Industries, Inc., Fort
`Collins, CO (US)
`
`(*)
`
`Notice:
`
`Subject to any disclaimer, the term of this
`patent is extended or adjusted under 35
`U.S.C. 154(b) by 0 days.
`
`(21)
`
`Appl. No.: 13/288,712
`
`(22) Filed:
`
`Nov. 3, 2011
`
`(65)
`
`Prior Publication Data
`
`US 2012/0112815 Al May 10, 2012
`
`Related U.S. Application Data
`
`(60) Provisional application No. 61/410,330, filed on Nov.
`4, 2010.
`
`(51) Int. Cl.
`H03K 17/16
`H03K 19/003
`(52) U.S. Cl.
` 326/30; 326/33; 326/34
`USPC
`(58) Field of Classification Search
`None
`See application file for complete search history.
`
`(2006.01)
`(2006.01)
`
`(56)
`
`References Cited
`
`U.S. PATENT DOCUMENTS
`3,872,325 A
`3/1975 Adams et al.
`5,054,114 A
`10/1991 Erickson
`5,670,881 A *
`9/1997 Arakawa et al.
`6,677,828 B1
`1/2004 Harnett et al.
`6,887,339 B1
`5/2005 Goodman et al.
`6,927,647 B2
`8/2005 Starri et al.
`6,992,543 B2
`1/2006 Luetzelschwab et al.
`7,042,311 B1
`5/2006
`Hilliker et al.
`7,109,788 B2
`9/2006
`Jevtic et al.
`7,368,971 B2
`5/2008 Pengelly
`7,369,096 B2
`5/2008 Castaneda et al.
`7,764,140 B2
`7/2010 Nagarkatti et al.
`7,796,969 B2
`9/2010
`Kelly et al.
`2006/0119451 Al
`6/2006
`Chen
`2008/0238569 Al * 10/2008 Matsuo
`
`324/322
`
`333/32
`
`* cited by examiner
`Primary Examiner — Anh Tran
`(74) Attorney, Agent, or Firm — Neugeboren O'Dowd PC
`(57)
`ABSTRACT
`In accordance with this invention the above and other prob-
`lems are solved by a switching apparatus and method that
`uses a switching circuit having a pair of parallel solid-state
`diodes (e.g., PN diodes), one of which is connected to a
`transistor (e.g., power MOSFET or IGBT), to switch a capaci-
`tor in or out of a variable capacitance element of an imped-
`ance matching network. Charging a body capacitance of the
`transistor reverse biases one of the two diodes so as to isolate
`the transistor from the RF signal enabling a low-cost high
`capacitance transistor to be used. Multiple such switching
`circuits and capacitors are connected in parallel to provide
`variable impedance for the purpose of impedance matching.
`
`27 Claims, 11 Drawing Sheets
`
`TO GENERATOR 102
`
`202
`
`220
`
`03
`2081
`/
`1
`
`
`'-'V2-1 I T
`
`/00
`1 1
`I T
`
`TO PLASMA LOAD 106
`
`1
`1
`
`1
`1
`TT
`
`0
`
`/°2
`511
`
`50
`
`24
`
`08
`
`06
`
`—1 218
`210 j
`
`20f
`
`DC Bias
`
`Switch Control
`
`Switch Control
`
`50
`
`526
`
`532
`
`52
`
`530
`
`DC Bias
`
`Page 1 of 20
`
`ADVANCED ENERGY INDUSTRIES INC.
`Exhibit 1005
`
`

`

`lualud °Sil
`
`H JO 1 JamiS
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`Zll £179`9£17`8 Sf1
`
`FIGURE 1
`
`106
`
`Plasma Load
`
`104
`
`Match Network
`
`102
`
`Generator
`
`110
`
`108
`
`Page 2 of 20
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`ADVANCED ENERGY INDUSTRIES INC.
`Exhibit 1005
`
`

`

`lualud °Sil
`
`H JO Z WIN
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`FIGURE 2
`
`212
`
`''....0.
`_J..,
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`201
`
`210 j
`218
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`206
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`204
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`2221
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`el
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`r
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`202
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`220
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`203
`
`/
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`0
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`mi.
`
`TO PLASMA LOAD 106
`
`/00
`
`TO GENERATOR 102
`
`-41
`
`Page 3 of 20
`
`ADVANCED ENERGY INDUSTRIES INC.
`Exhibit 1005
`
`

`

`lualud °Sil
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`H JO £ JamiS
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`FIGURE 3
`
`Biased
`
`Diode (206) is Reverse
`
`•-,-,
`
`,-„,-,
`
`312 current nn
`
`308 current
`
`318 voltage
`
`314 voltage 0
`
`..-
`
`
`
`310 voltage
`
`-%
`
`1.
`
`)
`
`....-..„
`
`304 current
`
`306 voltage
`
`I
`
`Transistor On
`
`Transistor Off
`
`Transistor On
`
`206
`
`Across Diode
`
`rw)nn
`
`Node 218
`
`Node 222
`
`Node 220
`
`Page 4 of 20
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`ADVANCED ENERGY INDUSTRIES INC.
`Exhibit 1005
`
`

`

`lualud °Sil
`
`H Jo 17 PatiS
`
`Zll £179`9£17`8 Sf1
`
`FIGURE 4
`
`418
`
`410
`
`Switch Control
`
`401
`
`._
`
`DC Bias
`
`406
`
`A
`
`>
`
`404
`
`405
`
`424
`
`i
`
`0
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`x408
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`r
`
`40t
`
`402
`
`0
`
`Page 5 of 20
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`ADVANCED ENERGY INDUSTRIES INC.
`Exhibit 1005
`
`

`

`lualud °Sil
`
`H JO S WIN
`
`Zll £179`9£17`8 Sf1
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`FIGURE 5
`
`532
`
`DC Bias
`
`+
`
`530
`
`528
`
`518
`
`510
`
`506
`
`-1
`
`/08
`
`526
`
`501
`
`._
`
`524
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`505
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`r
`50t
`
`5()2
`
`o
`
`Switch Control
`
`Switch Control
`
`DC Bias
`
`Page 6 of 20
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`ADVANCED ENERGY INDUSTRIES INC.
`Exhibit 1005
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`

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`lualud °Sil
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`II JO 9 JamiS
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`Zll £179`9£17`8 Sf1
`
`End
`
`complete?
`
`Tuning
`
`FIGURE 6
`
`616
`
`608
`
`increasing impedance of the variable capacitance
`Increase current in the switched capacitor, thus
`
`element
`
`reducing impedance of the variable capacitance
`Reduce current in the switched capacitor, thus
`
`element
`
`614
`
`Forward bias the first diode
`
`606
`
`Reverse bias the first diode
`
`Discharge body capacitance of the transistor
`
`Charge body capacitance of the transistor
`
`612
`
`604
`
`610
`
`Turn transistor on
`
`602
`
`Turn transistor off
`
`or decrease impedance?
`
`601
`
`Increase
`
`match tuning
`impedance
`
`Begin
`
`Page 7 of 20
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`ADVANCED ENERGY INDUSTRIES INC.
`Exhibit 1005
`
`

`

`U.S. Patent
`
`May 7, 2013
`
`Sheet 7 of 11
`
`US 8,436,643 B2
`
`700
`
`701
`
`COMPUTER SYSTEM
`/\/
`740
`
`720
`
`<
`
`NETWORK
`INTERFACE
`
`721
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`
`PROCESSOR(S)
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`
`CACHE
`
`703
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`
`MEMORY
`
`704
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`705 1
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`706 1
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`RAM
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`ROM
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`BIOS
`
`STORAGE CONTROL <
`
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`
`A
`
`707
`
`708
`
`V
`STORAGE
`
`709 1 OPERATING
`
`SYSTEM
`
`710 1
`
`EXECs
`
`DATA
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`
`APPLICATIONS
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`VIDEO INTERFACE
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`724
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`OUTPUT
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`725
`
`735
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`> STORAGE DEVICE
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`INTERFACE
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`> STORAGE
`
`DEVICE(S)
`
`726
`
`STORAGE MEDIUM
`INTERFACE
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`736
`..-1
`
`> STORAGE
`
`MEDIUM
`
`FIGURE 7
`
`Page 8 of 20
`
`ADVANCED ENERGY INDUSTRIES INC.
`Exhibit 1005
`
`

`

`lualud °Sil
`
`H Jo 8 JamiS
`
`Zll £179`9£17`8 Sfl
`
`50ps
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`40ps
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`FIGURE 8
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`30ps
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`Page 9 of 20
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`ADVANCED ENERGY INDUSTRIES INC.
`Exhibit 1005
`
`

`

`lualud °Sil
`
`H JO 6 WIN
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`Zll £179`9£17`8 Sfl
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`50ps
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`40ps
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`FIGURE 9
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`1 .2KV
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`Page 10 of 20
`
`ADVANCED ENERGY INDUSTRIES INC.
`Exhibit 1005
`
`

`

`lualud °Sil
`
`H JO OT WIN
`
`Zll £179`9£17`8 Sfl
`
`50ps
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`40ps
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`FIGURE 10
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`Page 11 of 20
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`ADVANCED ENERGY INDUSTRIES INC.
`Exhibit 1005
`
`

`

`lualud °Sil
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`H jo H WIN
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`Zll £179`9£17`8 Sf1
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`FIGURE 11
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`Page 12 of 20
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`ADVANCED ENERGY INDUSTRIES INC.
`Exhibit 1005
`
`

`

`1
`HIGH FREQUENCY SOLID STATE
`SWITCHING FOR IMPEDANCE MATCHING
`
`FIELD OF THE DISCLOSURE
`
`The present invention relates generally to plasma process-
`ing. In particular, but not by way of limitation, the present
`invention relates to systems, methods and apparatuses for
`impedance-matching radio frequency power transmitted
`from a radio frequency generator to a plasma load in a semi-
`conductor processing chamber.
`
`BACKGROUND
`
`In the semiconductor manufacturing world, manufacturers
`produce plasma processing chambers that utilize radio fre-
`quency (RF) power to generate a plasma. In order to achieve
`efficient power transfer between the RF generator ("genera-
`tor") and the plasma load, an impedance-matching network
`("match") is often used to match the load impedance to a
`desired input impedance, typically 50 ohm. Plasma load
`impedance may vary depending on variables such as genera-
`tor frequency, power, chamber pressure, gas composition, and
`plasma ignition. The match accounts for these variations in
`load impedance by varying electrical elements, typically
`vacuum variable capacitors, internal to the match to maintain
`the desired input impedance.
`Match networks typically contain reactance elements,
`meaning elements that store energy in electrical and magnetic
`fields as opposed to resistive elements that dissipate electrical
`power. The most common reactance elements are capacitors,
`inductors and coupled inductors but others such as distributed
`circuits are also used. Match networks can also include loss-
`less elements including transmission lines and transformers.
`The only resistive elements in a match network are typically
`associated with losses in non-ideal reactive and los sles s com-
`ponents or components that do not take part in the impedance
`transformation such as components for sensing voltage, cur-
`rent, power or temperature.
`Match networks can comprise a number of variable reac-
`tance elements. For instance, vacuum variable capacitors can
`be used. However, these are bulky and expensive. In the
`alternative, banks of parallel capacitors having different
`capacitances, and being added or removed from the parallel
`circuit via electrical switches have also been considered.
`Often, such capacitor banks use high power PIN diodes (con-
`trolled by a transistor) to switch the capacitors in and out of
`the parallel system. However, such PIN diodes can be too
`slow for RF power applications, or can require excessive
`power to accomplish the switching at acceptable speeds. This
`in turn results in running the PIN diodes at high temperatures.
`PIN diodes are also expensive and only produced by a handful
`of manufacturers.
`
`SUMMARY OF THE DISCLOSURE
`
`Exemplary embodiments of the present invention that are
`shown in the drawings are summarized below. These and
`other embodiments are more fully described in the Detailed
`Description section. It is to be understood, however, that there
`is no intention to limit the invention to the forms described in
`this Summary of the Invention or in the Detailed Description.
`One skilled in the art can recognize that there are numerous
`modifications, equivalents and alternative constructions that
`fall within the spirit and scope of the invention as expressed in
`the claims.
`
`US 8,436,643 B2
`
`2
`Some embodiments of the disclosure may be characterized
`as a circuit of a variable capacitance element of an impedance
`matching network. The match comprises a capacitor, a first
`and second diode, and a transistor. The capacitor is coupled
`5 between a first voltage line and a first node. The first diode has
`an anode coupled to the first node and a cathode coupled to a
`second node. The second diode has an anode to couple to a
`second voltage line and a cathode to couple to the first node.
`The transistor has a first, second, and control terminals. The
`0 first terminal is coupled to the second node, the second ter-
`minal is coupled to the second voltage line, and the control
`terminal is coupled to a controller. The capacitor is switched
`into the variable capacitance element when the transistor is on
`and switched out after the transistor is off.
`5 Other embodiments of the disclosure may also be charac-
`terized as a circuit of a variable capacitance element of an
`impedance matching network. The match network includes a
`capacitor, a first diode, a low power DC bias, and a transistor.
`The capacitor is coupled between a first voltage line and a first
`20 node. The first diode has an anode coupled to the first node
`and a cathode coupled to a second node. The low power DC
`bias source provides a first DC bias to the first node via a
`second diode. The transistor has a first terminal, a second
`terminal, and a control terminal. The first terminal is coupled
`25 to the second node, the second terminal is coupled to a second
`voltage line, and the control terminal receives signals control-
`ling switching of the transistor. The capacitor is switched into
`the variable capacitance element when the transistor is on and
`switched out after the transistor is off
`30 Other embodiments of the disclosure can be characterized
`as a method of switching a capacitor in and out of a variable
`capacitance element of an impedance matching network. The
`method includes turning a transistor of the variable capaci-
`tance element off. Also, charging a body capacitance of the
`35 transistor via a first diode using current from an RF signal
`passing through the capacitor. Further, the method includes
`reverse biasing the first diode with a voltage supplied by the
`body capacitance of the transistor after charging. Also, reduc-
`ing the current from the RF signal through the capacitor to
`40 near zero amperes so as to reduce an impedance of the vari-
`able capacitance element. Additionally, the method include
`discharging the body capacitance of the transistor and for-
`ward biasing the first diode with current from the RF signal
`passing through the capacitor.
`
`45
`
`BRIEF DESCRIPTION OF THE DRAWINGS
`
`55
`
`Various objects and advantages and a more complete
`understanding of the present invention are apparent and more
`so readily appreciated by referring to the following detailed
`description and to the appended claims when taken in con-
`junction with the accompanying drawings:
`FIG. 1 is a circuit diagram of a plasma processing system
`according to one embodiment of this invention.
`FIG. 2 illustrates a variable capacitance element according
`to one embodiment herein disclosed.
`FIG. 3 illustrates voltage and current characteristics of the
`variable capacitance element illustrated in FIG. 2.
`FIG. 4 illustrates another embodiment of a switched
`60 capacitor and its respective switching circuit.
`FIG. 5 illustrates yet another embodiment of a switched
`capacitor and its respective switching circuit.
`FIG. 6 illustrates a method of tuning an impedance match
`network.
`FIG. 7 shows a diagrammatic representation of one
`embodiment of a machine in the exemplary form of a com-
`puter system.
`
`65
`
`Page 13 of 20
`
`ADVANCED ENERGY INDUSTRIES INC.
`Exhibit 1005
`
`

`

`US 8,436,643 B2
`
`3
`FIG. 8 illustrates an exemplary RF voltage plot.
`FIG. 9 illustrates another exemplary voltage plot.
`FIG. 10 illustrates yet another exemplary current plot.
`FIG. 11 illustrates still another exemplary voltage plot.
`
`DETAILED DESCRIPTION
`
`The present disclosure relates generally to plasma process-
`ing. More specifically, but without limitation, the present
`disclosure relates to match networks of a power supply for
`generating and sustaining a plasma in, or provided to, a
`plasma processing chamber.
`FIG. 1 is a circuit diagram of a plasma processing system
`according to one embodiment of this invention. A generator
`102 transmits RF power to a match network 104 ("match")
`via a transmission line 108 (e.g., coaxial cable) and then onto
`a plasma load 106 via an electrical connection 110. The match
`network 104 varies its internal electrical elements such that
`the input impedance of the match network 104 is close to the
`desired input impedance.
`The match 104 can include two or more variable capaci-
`tance elements coupled in parallel. Often such variable
`capacitance elements can be mechanically-varied capacitors,
`which, as described above, are bulky, slow, and expensive. In
`the alternative, variable capacitive elements can be made
`from banks of parallel electronically switched capacitors,
`which are smaller, faster, and cheaper than their mechanical
`counterparts. While the prior art uses PIN diodes to switch
`capacitors in and out of the variable capacitance element,
`FIG. 2 illustrates an embodiment in which common and inex-
`pensive transistors can be used to switch diodes (e.g., PN
`diodes) and thus switch the capacitors in and out of the vari-
`able capacitance element.
`The variable capacitance element 200 comprises various
`switched capacitors 208 coupled in parallel such that the
`switching in and out of each of the various switched capaci-
`tors 208 alters the impedance of the variable capacitance
`element 200. The switched capacitors 208 each have a switch-
`ing circuit 203 for switching the switched capacitors 208 in
`and out of the variable capacitance element 200. Each switch-
`ing circuit 208 can have a pair of solid-state diodes (e.g., PN
`or Schottky diodes) 204, 206, one of which can be connected
`to a transistor 210 (e.g., MOSFET, power MOSFET, IGBT, to
`name a few), configured to switch the switched capacitor 208
`into the variable capacitance element 200 when the transistor
`210 is on (closed). This switching can be achieved without the
`use of any external bias of the diode 206 (although small
`biases may be desirable under circumstances to be discussed
`later).
`When the transistor 210 is on, RF current passes between
`the first voltage line 202 and the second voltage line 201
`passing in a forward biased direction through each of the
`diodes 204, 206 alternately. As such, when the switched
`capacitor 208 is switched into the variable capacitance ele-
`ment 200, AC (e.g., RF) current can pass between the first
`voltage line 202 and the second voltage line 201 and the
`impedance of the match 104 increases.
`When the transistor is off, positive portions of the RF signal
`push current through the diode 206 and charge a body capaci-
`tance of the transistor 210 until a voltage across the transistor
`210, from node 218 to the second voltage line 201, is greater
`than a voltage between node 222 and the second voltage line
`201 (in other words the voltage across diode 204). When such
`a voltage exists, the diode 206 is reverse biased, thus prevent-
`ing substantially all current from passing through the
`
`5
`
`4
`switched capacitor 208. At this point the switching circuit 203
`can be considered off and the impedance of the match 104 is
`reduced.
`Capacitors 208 can be selected so that their capacitance
`increases as a power of 2 from left to right in the variable
`capacitance element 200. Accordingly, the capacitance can
`increases from left to right as squares of the capacitance of the
`leftmost capacitor (e.g., C*1, C*2, C*4, C*8, C*16, C*32,
`C*64, C*128). Therefore it is possible to vary the capacitance
`10 of the match in 256 steps by selectively driving the switching
`circuits 203.
`The variable capacitance element 200 can include any
`number of switched capacitors 208, although in the illustrated
`embodiment, there are eight (8) switched capacitors 208.
`15 Each switched capacitor 208 is wired in parallel to a generator
`102 and a plasma load 106, and between a first voltage line
`202 and a second voltage line 201 assuming a floating vari-
`able capacitance element 200. In some embodiments, the
`second voltage line 201 can be replaced by a grounded volt-
`2o age line or ground connections to each element illustrated as
`coupling to the second voltage line 201. The switched capaci-
`tor 208 is switched into the variable capacitance element 200
`(altering the reactance of the match network) when current
`passes through the switched capacitor 208 either to or from
`25 the first voltage line 202. This occurs when the transistor 210
`is closed (on).
`Inexpensive transistors have not been used to switch
`switched capacitors such as 208 because they typically dissi-
`pate large amounts of heat when operated at the RF frequen-
`30 cies and high powers associated with plasma processing.
`They also tend to have transient times far greater than that of
`the RF signal from the generator 102, in which case the
`transistor are typically unable to turn off. Transient time is the
`time required to turn a device on or off. In other words, the
`35 amount of time between the beginning of a switching action
`and a point at which a higher/lower steady state voltage or
`current has been achieved. SiC transistors, which have neg-
`ligible transient times, have been used with some success, but
`their expense along with distortion of the RF signal to the
`40 plasma load 106 makes them less than preferable.
`Typical match networks are designed to tune and then hold
`the tuned impedance until the load impedance changes. Dis-
`tortion occurs, when the impedance of the match continues to
`change after tuning is complete. Given a switched capacitor
`45 208 that is switched out during tuning, distortion means that
`some current still passes through the switched capacitor 208
`even though it is switched out. Given a switched capacitor 208
`that is switched in during tuning, distortion means that less
`than a full current passes through the switched capacitor 208
`so even though it is switched in. For instance, if a transistor were
`used as the sole switching component in switching circuit
`203, it would cause distortion since current would continue to
`pass through the switched capacitor 208 and charge and dis-
`charge a body capacitance of the transistor when it was off
`55 (the body capacitance is an inherent capacitance in transistor
`structures measured between the collector and ground termi-
`nal of a BJT or IGBT, or between the drain and ground of a
`FET). In other words, the switched capacitor 208 would never
`be completely switched out and hence would distort the RF
`60 signal after tuning was completed.
`Furthermore, the transistor body capacitance is in series
`with the switched capacitor 208 and therefore affects the
`current swing through the switched capacitor 208 occurring
`during switching. Typical body capacitance of a transistor is
`65 many times (e.g., two to four orders of magnitude) greater
`than that of the switched capacitor 208. Thus, far more volt-
`age drops across the switched capacitor 208 than across the
`
`Page 14 of 20
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`ADVANCED ENERGY INDUSTRIES INC.
`Exhibit 1005
`
`

`

`US 8,436,643 B2
`
`5
`body capacitance of a transistor when they are in series and
`the transistor is off. When the transistor is turned on, only a
`small increase in voltage drop across the switched capacitor
`208 occurs and thus only a small change in current. Thus, the
`switched capacitor 208 may not be effective at changing the
`impedance of the variable capacitance element 200 since the
`current through it will only nominally change when switched.
`In short, were the switching circuit 203 to merely comprise a
`transistor, the switched capacitor 208 would not have a very
`appreciable effect on the impedance when switched in and out
`of the variable capacitance element 200.
`This disclosure overcomes these deficiencies by shielding
`the transistor 210 (and thus its body capacitance) from the
`switched capacitor 208 when the transistor 210 is off and
`forcing RF current to only pass through the transistor 210 in
`one direction when it is on. To achieve these goals, an arrange-
`ment of two parallel diodes having opposite polarity are used.
`A first diode 206 is arranged between the switched capacitor
`208 and the transistor 210, with anode coupled to a first node
`222 and a cathode coupled to a second node 218, and the first
`diode 206 is biased such that current is largely precluded from
`passing from the first voltage line 202 through the first diode
`206 to the transistor 210 when the transistor 210 is off (open).
`In other words, when the transistor 210 is off, the first diode
`206 is reverse biased.
`When a voltage on the first voltage line 202 swings low, the
`body capacitance of the transistor 210 does not discharge
`through the switched capacitor 208 because the first diode
`206 is still reverse biased. In other words, when the transistor
`210 is off, the first diode 206 is reverse biased regardless of
`whether the voltage on the first voltage line 202 is positive or
`negative. As such, current from the first voltage line 202 or
`from the second voltage line 201 is largely unable to pass
`through the transistor 210 when it is off, and the large off-state
`body capacitance of the transistor 210 is unseen by the
`switched capacitor 208.
`It should be clear to one of skill in the art that the first diode
`206 is reverse biased when the voltage on the first voltage line
`202 swings negative. But the ability of the first diode 206 to
`remain reverse biased when the voltage on the first voltage
`line 202 swings positive is novel and unexpected. In this case,
`there is a short time wherein the first diode 206 can be forward
`biased, but during this time the body capacitance of the tran-
`sistor 210 charges and the voltage between the first node 218
`and second voltage line 201 rises. When this voltage ("tran-
`sistor body voltage") is larger than a voltage from the second
`node 222 to the second voltage line 201 (minus a diode
`voltage drop), the first diode 206 becomes reverse biased.
`This can be referred to as autobiasing, since the first diode 206
`is reverse biased by the voltages inherent in the switching
`circuit 203 rather than via an external bias supply.
`Accordingly, some embodiments of this disclosure com-
`prise a switching circuit 203 to switch a switched capacitor
`208 in and out of a variable capacitance element 200, using
`low-cost transistors that achieve large changes in voltage
`across the switched capacitor 208 when the capacitor 208 is
`switched in and out of the variable capacitance element 200.
`They do so with low distortion of the RF signal on the first
`voltage line 202, and do so with a minimum of devices (e.g.,
`two diodes in parallel) and no further biasing sources.
`To illustrate an example of the operation of a switching
`circuit 203, assume the RF signal has a peak-to-peak voltage
`of two hundred volts (200 V), and transistor 210 is turned off.
`The voltage at the node 220 goes to negative one hundred
`volts (-100 V) in the negative half cycle of the RF signal.
`Second diode 204 is forward bias, so the voltage at the first
`node 222 is just below ground (--0.7 V). The First diode 206
`
`5
`
`35
`
`10
`
`6
`is reverse biased. In the positive half cycle of RF signal, the
`second diode 204 is reverse biased. The voltage at node 220
`goes from negative hundred volts (-100y) to positive hun-
`dred volts (+100y), and the voltage at the first node 222 rises
`toward two hundred volts (+200 V). As the voltage at the first
`node 222 goes positive, the first diode 206 turns on and starts
`to charge the body capacitance of transistor 210 because the
`transistor is turned off. When the voltage on the body capaci-
`tance of transistor 210 goes more positive than the voltage
`across the second diode 204, the first diode 206 turns off.
`Depending on the frequency of the RF signal, the voltage
`across the body capacitance of the transistor 210 will rise in
`one or more cycles to two hundred volts (200 V). With both
`15 diodes 204,206 reverse biased (biased off), no current flows
`through switched capacitor 208, and the switched capacitor
`208 is electrically removed from the variable capacitance
`element 200 thus reducing the match impedance.
`When the transistor 210 is turned on (closed), the voltage
`20 on the body capacitance of the transistor 210 discharges, and
`the voltage at the first node 222 goes near to the voltage of the
`second voltage line 201. Now diodes 204 and 206 are holding
`the voltage at the first node 222 near the voltage of the second
`voltage line 201 throughout the entire cycle of the RF signal,
`25 and switched capacitor 208 is electrically added into the
`circuit of the variable capacitor element 200 thus increasing
`the match impedance.
`It should be noted that while the transistor 210 is illustrated
`as an N-channel MOSFET, various other transistors can also
`30 be implemented including, but not limited to, IGBTs.
`A controller 212 provides the control signal (e.g., gate
`signal in a FET or base signal in a BJT or IGBT) to a control
`terminal of the transistor 210 to control the on and off state of
`the transistor 210.
`FIG. 3 illustrates voltage and current characteristics of the
`variable capacitance element 200 illustrated in FIG. 2. For
`these plots, a sinusoidal RF signal is illustrated as seen by the
`current 304 and voltage 306 measured at the node 220. The
`voltage 306 is measured relative to the second voltage line
`40 201 (which is optionally grounded). While a more complex
`RF signal can be used in practice, these descriptions are made
`simpler by using a simple sinusoidal signal. When the tran-
`sistor 210 is turned off, the current 304 and voltage 306 drop
`slightly accounting for the voltage and current that are being
`45 applied to the switched capacitor 208 and the switching cir-
`cuit 203.
`At the first node 222, between the switched capacitor 208
`and the diodes 204, 206, current 308 at the first node 222
`when the transistor 210 is on is proportional to the current 304
`so at the node 220. When the transistor 210 turns off the
`switched capacitor 208 is switched out of the variable capaci-
`tance element 200 and current 308 ceases to pass through the
`switched capacitor 208.
`In contrast, while the transistor 210 is on, there is very little
`55 voltage drop across the diodes 204, 206, and the transistor
`210, and thus the voltage 310 at the first node 222 is near zero
`(the voltage 310 may fluctuate around 0 V when the transistor
`210 is on, but is not illustrated in FIG. 3 for simplicity). When
`the transistor 210 turns off, the voltage 310 at the first node
`60 222 begins to fluctuate in phase with the voltage 306 on the
`first voltage line 202, but with an amplitude roughly equal to
`the peak-to-peak voltage 306 on the first voltage line 202. In
`addition, while the transistor 210 is off, some current periodi-
`cally passes through the switched capacitor 208 and the first
`65 diode 206 and charges the body capacitance of the transistor
`210. This effect is responsible for the gradual rise in voltage
`310 seen while the transistor 210 is off When the transistor is
`
`Page 15 of 20
`
`ADVANCED ENERGY INDUSTRIES INC.
`Exhibit 1005
`
`

`

`US 8,436,643 B2
`
`7
`closed, the voltage 310 returns to near 0 V and the current 308
`again resembles the current 304 on the first voltage line 202.
`At the second node 218, between the first diode 206 and the
`transistor 210, current is the same as the current 308 at the first
`node 222, except that the first diode 206 rectifies the current
`312 such that only positive current 312 reaches the second
`node 218. The voltage 314, which is the voltage across the
`transistor 210, is small when the transistor is on, and thus is
`not illustrated for simplicity. When the transistor 210 turns
`off, the voltage 310 at the first node 222 begins to charge the
`body capacitance of the transistor 210 whenever the voltage
`310 is greater than the voltage 314 across the transistor 210.
`This leads to the illustrated step-like waveform, wherein the
`voltage 314 bumps up or increases every time that the voltage
`310 rises above the transistor voltage 314. This voltage 314
`falls to near zero when the transistor 210 turns back on, and
`similarly the rectified current 312 begins to flow through the
`first diode 206 and the transistor 210 again.
`Little to no voltage 318 is across the first diode 206 when
`the transistor 210 is on. However, when the transistor 210
`turns off, the first diode 206 is forward biased by the voltage
`310. The amount of forward bias quickly declines as the body
`capacitance of the transistor 210 charges and the voltage 314
`across the transistor 210 increases. Eventually, the voltage
`314 across the transistor 210 is large enough that it reverse
`biases the first diode 206 thus turning the first diode 206 off
`and isolating the transistor 210 from the RF voltages 306 and
`310. This process of reverse biasing the first diode 206 using
`just the RF voltage and body capacitance of the transistor 210
`is herein referred to as autobiasing the first diode 206.
`The plots in FIG. 3 are not illustrated to scale and do not
`represent preferred frequencies or amplitudes. These are
`merely exaggerated and simplified renditions of voltage and
`current characteristics illustrated for the purposes of aiding
`the reader in understanding the functioning of the circuit
`elements of FIG. 2 as well as further embodiments discussed
`later in this disclosure. More realistic plots are illustrated in
`FIGS. 8-11, although it should be recognized that even these
`plots are based on Spice models and thus do not perfectly
`represent actual current and voltage waveforms. Also varia-
`tions in the circuit elements used in different embodiments
`(e.g., larger or smaller capacitance) may alter the shape, phas-
`ing, and amplitude of the waveforms illustrated in FIGS.
`8-11.
`FIG. 8 illustrates an exemplary RF voltage signal at the
`node 220. The illustrated RF voltage signal is analogous to the
`voltage 306 in FIG. 3. This signal represents a simple sinu-
`soidal RF signal as provided by a generator such as generator
`102 and as measured on the first voltage line 202.
`FIG. 9 illustrates an