lQ
`
`QT110
`QTOUCH™ SENSOR IC
`
`Vss
`
`Sns2
`
`Sns1
`
`Gain
`
`5678
`
`QT110
`
`1 2 3 4
`
`Vdd
`
`Out
`
`Opt1
`
`Opt2
`
` Less expensive than many mechanical switches
` Projects a ‘touch button’ through any dielectric
` 100% autocal for life - no adjustments required
` No active external components
` Piezo sounder direct drive for ‘tactile’ click feedback
` LED drive for visual feedback
` 2.5 ~ 5V single supply operation
`
` 10µA at 2.5V - very low power drain
` Toggle mode for on/off control (via option pins)
` 10s or 60s auto-recalibration timeout (via option pins)
` Pulse output mode (via option pins)
` Gain settings in 3 discrete levels
` Simple 2-wire operation possible
` HeartBeat™ health indicator on output
` Pb-Free packages
`
`APPLICATIONS -
`
` Light switches
` Industrial panels
`
` Appliance control
` Security systems
`
` Access systems
` Pointing devices
`
` Elevator buttons
` Consumer electronics
`
`The QT110 charge-transfer (“QT’”) sensor IC is a self-contained digital IC used to implement near-proximity or touch sensors. It
`projects sense fields through almost any dielectric, like glass, plastic, stone, ceramic, and wood. It can also turn small metal-bearing
`objects into intrinsic sensors, making them respond to proximity or touch. This capability coupled with an ability to self-calibrate
`continuously leads to entirely new product concepts.
`
`The QT110 is designed specifically for human interfaces, like control panels, appliances, toys, lighting controls, or anywhere a
`mechanical switch or button may be found; they may also be used for some material sensing and control applications provided that
`the presence duration of objects does not exceed the recalibration timeout interval.
`
`A piezo element can also be connected to create a feedback click sound.
`
`This IC requires only a common inexpensive capacitor in order to function. Average power consumption is under 20µA in most
`applications, allowing battery operation.
`
`The QT110 employs digital signal processing techniques pioneered by Quantum, designed to make it survive real-world challenges,
`such as ‘stuck sensor’ conditions and signal drift. Sensitivity is digitally determined for the highest possible stability. No external active
`components are required for operation.
`
`The device includes several user-selectable built-in features. One, toggle mode, permits on/off touch control for example for light
`switch replacement. Another makes the sensor output a pulse instead of a DC level, which allows the device to 'talk' over the power
`rail, permitting a simple 2-wire twisted-pair interface. Quantum’s unique HeartBeat™ signal is also included, allowing a host controller
`to continuously monitor sensor health.
`
`By using the charge transfer principle, the QT110 delivers a level of performance clearly superior to older technologies in a highly
`cost-effective package.
`
`TA
`00C to +700C
`-400C to +850C
`
`AVAILABLE OPTIONS (Pb-FREE)
`SOIC
`-
`QT110-ISG
`
`8-PIN DIP
`QT110-DG
`-
`
`lq
`
`©1999-2004 Quantum Research Group
`QT110 R1.04/0405
`
`Petitioner STMICROELECTRONICS, INC.,
`Ex. 1015, IPR2021-01160, Page 1 of 12
`
`

`

`1 - OVERVIEW
`The QT110 is a digital burst mode charge-transfer (QT) sensor
`designed specifically for touch controls; it includes all hardware
`and signal processing functions necessary to provide stable
`sensing under a wide variety of changing conditions. Only a
`few low cost, non-critical discrete external parts are required for
`operation.
`
`Figure 1-1 shows the basic QT110 circuit using the device,
`with a conventional output drive and power supply
`connections. Figure 1-2 shows a second configuration using a
`common power/signal rail which can be a long twisted pair from
`a controller; this configuration uses the built-in pulse mode to
`transmit output state to the host controller (QT110 only).
`
`1.1 BASIC OPERATION
`The QT110 employs low duty cycle bursts of charge-transfer
`cycles to acquire its signal. Burst mode permits power
`consumption in the low microamp range, dramatically reduces
`EMC problems, and yet permits excellent response time.
`Internally the signals are digitally processed to reject impulse
`noise, using a 'consensus' filter which requires four
`consecutive confirmations of a detection before the output is
`activated.
`
`The QT switches and charge measurement hardware functions
`are all internal to the QT110 (Figure 1-3). A single-slope
`switched capacitor ADC includes both the required QT charge
`and transfer switches in a configuration that provides direct
`ADC conversion. Vdd is used as the charge reference voltage.
`
`Larger values of Cx cause the charge transferred into Cs to
`rise more rapidly, reducing available resolution; as a minimum
`resolution is required for proper operation, this can result in
`dramatically reduced apparent gain.
`
`Figure 1-1 Standard mode options
`
`+2.5 ~ +5
`
`1
`
`Vdd
`
`OUT
`
`SNS2
`
`OPT1
`
`GAIN
`
`OPT2
`
`SNS1
`
`7
`
`5
`
`6
`
`2
`
`3
`
`4
`
`RE
`
`SENSING
`ELECTRODE
`
`Rs
`
`Cs
`
`Cx
`
`OUTPUT = DC
`TIMEOUT = 10 Secs
`TOGGLE = OFF
`GAIN = HIGH
`
`Vss
`
`8
`
`2nF - 500nF
`
`1.2 ELECTRODE DRIVE
`The internal ADC treats Cs as a floating transfer capacitor; as a
`direct result, the sense electrode can in theory be connected to
`either SNS1 or SNS2 with no performance difference.
`However, the noise immunity of the device is improved by
`connecting the electrode to SNS2, preferably via a series
`resistor Re (Figure 1-1) to roll off higher harmonic frequencies,
`both outbound and inbound.
`
`In order to reduce power consumption and to assist in
`discharging Cs between acquisition bursts, a 470K series
`resistor Rs should be connected across Cs (Figure 1-1).
`
`The rule Cs >> Cx must be observed for proper operation.
`Normally Cx is on the order of 10pF or so, while Cs might be
`10nF (10,000pF), or a ratio of about 1:1000.
`
`The IC is highly tolerant of changes in Cs since it computes the
`signal threshold level ratiometrically. Cs is thus non-critical and
`can be an X7R type. As Cs changes with temperature, the
`internal drift compensation mechanism also adjusts for the drift
`automatically.
`
`It is important to minimize the amount of unnecessary stray
`capacitance Cx, for example by minimizing trace lengths and
`widths and backing off adjacent ground traces and planes so
`as keep gain high for a given value of Cs, and to allow for a
`larger sensing electrode size if so desired.
`
`Piezo sounder drive: The QT110 can drive a piezo sounder
`after a detection for feedback. The piezo sounder replaces or
`augments the Cs capacitor; this works since piezo sounders
`are also capacitors, albeit with a large thermal drift coefficient.
`If Cpiezo is in the proper range, no additional capacitor is
`required. If Cpiezo is too small, it can simply be ‘topped up’ with a
`ceramic capacitor in parallel. The QT110 drives a ~4kHz signal
`across SNS1 and SNS2 to make the piezo (if installed) sound a
`short tone for 75ms immediately after detection, to act as an
`audible confirmation.
`
`Option pins allow the selection or alteration of several other
`special features and sensitivity.
`
`The PCB traces, wiring, and any components associated with
`or in contact with SNS1 and SNS2 will become touch sensitive
`and should be treated with caution to limit the touch area to the
`desired location.
`
`1.3 ELECTRODE DESIGN
`
`1.3.1 ELECTRODE GEOMETRY AND SIZE
`There is no restriction on the shape of the electrode; in most
`cases common sense and a little experimentation can result in
`a good electrode design. The QT110 will operate equally well
`with long, thin electrodes as with round or square ones; even
`random shapes are acceptable. The electrode can also be a
`3-dimensional surface or object. Sensitivity is related to
`electrode surface area, orientation with respect to the object
`being sensed, object composition, and
`the ground coupling quality of both the
`sensor circuit and the sensed object.
`
`Figure 1-2 2-wire operation, self-powered
`
`3.5 - 5.5V
`
`CMOS
`LOGIC
`
`1K
`
`Twisted
` pair
`
`1N4148
`
`n-ch Mosfet
`
`2
`
`3
`
`4
`
`+
`
`1
`
`Vdd
`
`10µF
`
`OUT
`
`SNS2
`
`OPT1
`
`GAIN
`
`OPT2
`
`SNS1
`
`Vss
`
`8
`
`RE
`
`SENSING
`ELECTRODE
`
`Rs
`
`Cx
`
`Cs
`
`7
`
`5
`
`6
`
`1.3.2 KIRCHOFF’S CURRENT LAW
`Like all capacitance sensors, the QT110
`relies on Kirchoff’s Current Law (Figure
`1-5) to detect the change in capacitance
`of the electrode. This law as applied to
`capacitive sensing requires that the
`sensor’s field current must complete a
`loop, returning back to its source in
`order for capacitance to be sensed.
`Although most designers relate to
`Kirchoff’s law with regard to hardwired
`circuits, it applies equally to capacitive
`
`LQ 2
`
`QT110 R1.04/0405
`
`Petitioner STMICROELECTRONICS, INC.,
`Ex. 1015, IPR2021-01160, Page 2 of 12
`
`

`

`field flows. By implication it requires that
`the signal ground and the target object
`must both be coupled together in some
`manner for a capacitive sensor to
`operate properly. Note that there is no
`need to provide actual hardwired ground
`connections; capacitive coupling to
`ground (Cx1) is always sufficient, even if
`the coupling might seem very tenuous.
`For example, powering the sensor via an
`isolated transformer will provide ample
`ground coupling, since there is
`capacitance between the windings
`and/or the transformer core, and from
`the power wiring itself directly to 'local
`earth'. Even when battery powered, just
`the physical size of the PCB and the
`object into which the electronics is
`embedded will generally be enough to
`couple a few picofarads back to local
`earth.
`
`1.3.3 VIRTUAL CAPACITIVE GROUNDS
`When detecting human contact (e.g. a fingertip), grounding of
`the person is never required. The human body naturally has
`several hundred picofarads of ‘free space’ capacitance to the
`local environment (Cx3 in Figure 1-3), which is more than two
`orders of magnitude greater than that required to create a
`return path to the QT110 via earth. The QT110's PCB however
`can be physically quite small, so there may be little ‘free space’
`coupling (Cx1 in Figure 1-3) between it and the environment to
`complete the return path. If the QT110 circuit ground cannot be
`earth grounded by wire, for example via the supply
`connections, then a ‘virtual capacitive ground’ may be required
`to increase return coupling.
`
`A ‘virtual capacitive ground’ can be created by connecting the
`QT110’s own circuit ground to:
`
`- A nearby piece of metal or metallized housing;
`- A floating conductive ground plane;
`- Another electronic device (to which its might be connected
`already).
`
`Free-floating ground planes such as metal foils should
`maximize exposed surface area in a flat plane if possible. A
`square of metal foil will have little effect if it is rolled up or
`crumpled into a ball. Virtual ground planes are more effective
`and can be made smaller if they are physically bonded to other
`surfaces, for example a wall or floor.
`
`1.3.4 SENSITIVITY
`The QT110 can be set for one of 3 gain levels using option pin
`5 (Table 1-1). If left open, the gain setting is high. The
`sensitivity change is made by altering the numerical threshold
`level required for a detection. It is also a function of other
`things: electrode size, shape, and orientation, the composition
`and aspect of the object to be sensed, the thickness and
`composition of any overlaying panel material, and the degree
`of ground coupling of both sensor and object are all influences.
`
`Gain plots of the device are shown on page 9.
`
`The Gain input should never be tied to anything other than
`SNS1 or SNS2, or left unconnected (for high gain setting).
`
`Figure 1-3 Internal Switching & Timing
`
`E LE C T RO DE
`
`S NS 2
`
`C s
`
`C x
`
`S NS 1
`
`Switched Capacitor ADC
`
`Single-Slope 14-bit
`
`Do ne
`
`Burst Controller
`
`R esul t
`
`S tart
`
`C ha rg e
`A m p
`
`In some cases it may be desirable to increase sensitivity
`further, for example when using the sensor with very thick
`panels having a low dielectric constant.
`
`Sensitivity can often be increased by using a bigger electrode,
`reducing panel thickness, or altering panel composition to one
`having a higher dielectric constant. Increasing electrode size
`can have diminishing returns, as high values of Cx will reduce
`sensor gain.
`
`Increasing the electrode's surface area will not substantially
`increase touch sensitivity if its diameter is already much larger
`in surface area than the object being detected. Metal areas
`near the electrode will reduce the field strength and increase
`Cx loading and are to be avoided for maximal gain.
`
`Ground planes around and under the electrode and its SNS
`trace will cause high Cx loading and destroy gain. The possible
`signal-to-noise ratio benefits of ground area are more than
`negated by the decreased gain from the circuit, and so ground
`areas around electrodes are discouraged. Keep ground,
`power, and other signals traces away from the electrodes and
`SNS wiring.
`
`The value of Cs has a minimal effect on sensitivity with these
`devices, but if the Cs value is too low there can be a sharp
`drop-off in sensitivity.
`
`Figure 1-5 Kirchoff's Current Law
`
`CX 2
`
`S e n se E le ctro de
`
`S E NS O R
`
`Table 1-1 Gain Strap Options
`
`CX 1
`
`Gain
`
`High
`
`Medium
`
`Low
`
`Tie Pin 5 to:
`
`Leave open
`
`Pin 6
`
`Pin 7
`
`Su rro un d ing e n v iro n m e n t
`
`C
`
`X 3
`
`LQ 3
`
`QT110 R1.04/0405
`
`Petitioner STMICROELECTRONICS, INC.,
`Ex. 1015, IPR2021-01160, Page 3 of 12
`
`

`

`2 - QT110 SPECIFICS
`
`2.1 SIGNAL PROCESSING
`The QT110 processes all signals using a number of algorithms
`pioneered by Quantum. The algorithms are specifically
`designed to provide for high 'survivability' in the face of all kinds
`of adverse environmental changes.
`
`2.1.1 DRIFT COMPENSATION ALGORITHM
`Signal drift can occur because of changes in Cx and Cs over
`time. It is crucial that drift be compensated for, otherwise false
`detections, non-detections, and sensitivity shifts will follow. Cs
`drift has almost no effect on gain since the threshold method
`used is ratiometric. However Cs drift can still cause false
`detections if the drift occurs rapidly.
`
`Drift compensation (Figure 2-1) is performed by making the
`reference level track the raw signal at a slow rate, but only
`while there is no detection in effect. The rate of adjustment
`must be performed slowly, otherwise legitimate detections
`could be ignored. The QT110 drift compensates using a
`slew-rate limited change to the reference level; the threshold
`and hysteresis values are slaved to this reference.
`
`Once an object is sensed, the drift compensation mechanism
`ceases since the signal is legitimately high, and therefore
`should not cause the reference level to change.
`
`The QT110's drift compensation is 'asymmetric': the reference
`level drift-compensates in one direction faster than it does in
`the other. Specifically, it compensates faster for decreasing
`signals than for increasing signals. Increasing signals should
`not be compensated for quickly, since an approaching finger
`could be compensated for partially or entirely before even
`touching the sense pad. However, an obstruction over the
`sense pad, for which the sensor has already made full
`allowance for, could suddenly be removed leaving the sensor
`with an artificially elevated reference level and thus become
`insensitive to touch. In this latter case, the sensor will
`compensate for the object's removal very quickly, usually in
`only a few seconds.
`
`2.1.2 THRESHOLD CALCULATION
`Sensitivity is dependent on the threshold level as well as ADC
`gain; threshold in turn is based on the internal signal reference
`level plus a small differential value. The threshold value is
`established as a percentage of the absolute signal level. Thus,
`sensitivity remains constant even if Cs is altered dramatically,
`so long as electrode coupling to the user remains constant.
`Furthermore, as Cx and Cs drift, the threshold level is
`automatically recomputed in real time so that it is never in error.
`
`The QT110 employs a hysteresis dropout below the threshold
`level of 50% of the delta between the reference and threshold
`levels.
`
`The threshold setting is determined by option jumper; see
`Section 1.3.4.
`
`2.1.4 DETECTION INTEGRATOR
`It is desirable to suppress detections generated by electrical
`noise or from quick brushes with an object. To accomplish this,
`the QT110 incorporates a detect integration counter that
`increments with each detection until a limit is reached, after
`which the output is activated. If no detection is sensed prior to
`the final count, the counter is reset immediately to zero. In the
`QT110, the required count is 4.
`
`The Detection Integrator can also be viewed as a 'consensus'
`filter, that requires four detections in four successive bursts to
`create an output. As the basic burst spacing is 75ms, if this
`spacing was maintained throughout all 4 counts the sensor
`would react very slowly. In the QT110, after an initial detection
`is sensed, the remaining three bursts are spaced about 20ms
`apart, so that the slowest reaction time possible is
`75+20+20+20 or 135ms and the fastest possible is 60ms,
`depending on where in the initial burst interval the contact first
`occurred. The response time will thus average about 95ms.
`
`2.1.5 FORCED SENSOR RECALIBRATION
`The QT110 has no recalibration pin; a forced recalibration is
`accomplished only when the device is powered up. However,
`the supply drain is so low it is a simple matter to treat the entire
`IC as a controllable load; simply driving the QT110's Vdd pin
`directly from another logic gate or a microprocessor port
`(Figure 2-2) will serve as both power and 'forced recal'. The
`source resistance of most CMOS gates and microprocessors is
`low enough to provide direct power without any problems.
`Almost any CMOS logic gate can directly power the QT110.
`
`A 0.01uF minimum bypass capacitor close to the device is
`essential; without it the device can break into high frequency
`oscillation.
`
`Option strap configurations are read by the QT110 only on
`powerup. Configurations can only be changed by powering the
`QT110 down and back up again; again, a microcontroller can
`directly alter most of the configurations and cycle power to put
`them in effect.
`
`2.2 OUTPUT FEATURES
`The devices are designed for maximum flexibility and can
`accommodate most popular sensing requirements. These are
`selectable using strap options on pins OPT1 and OPT2. All
`options are shown in Table 2-1.
`
`OPT1 and OPT2 should never be left floating. If they are
`floated, the device will draw excess power and the options will
`not be properly read on powerup. Intentionally, there are no
`pullup resistors on these lines, since pullup resistors add to
`power drain if the pin(s) are tied low.
`
`2.2.1 DC MODE OUTPUT
`The output of the device can respond in a DC mode, where the
`output is active-low upon detection. The output will remain
`active for the duration of the detection, or until the Max
`
`2.1.3 MAX ON-DURATION
`If an object or material obstructs the sense pad the
`signal may rise enough to create a detection,
`preventing further operation. To prevent this, the
`sensor includes a timer which monitors detections.
`If a detection exceeds the timer setting, the timer
`causes the sensor to perform a full recalibration.
`This is known as the Max On-Duration feature.
`
`After the Max On-Duration interval, the sensor will
`once again function normally, even if partially or
`fully obstructed, to the best of its ability given
`electrode conditions. There are two nominal
`timeout durations available via strap option: 10 and
`60 seconds. The accuracy of these timeouts is
`approximate.
`
`Figure 2-1 Drift Compensation
`
`Signal
`
`H ysteresis
`
`Threshold
`
`R eference
`
`Output
`
`LQ 4
`
`QT110 R1.04/0405
`
`Petitioner STMICROELECTRONICS, INC.,
`Ex. 1015, IPR2021-01160, Page 4 of 12
`
`

`

`Figure 2-2 Powering From a CMOS Port Pin
`
`Figure 2-3 Damping Piezo Clicks with Rs
`
`RE
`
`SENSING
`ELECTRODE
`
`Rs
`
`Cx
`
`+2.5 ~ +5
`
`10-30nF
`
`Piezo Sounder
`
`7
`
`5
`
`6
`
`2
`
`3
`
`4
`
`1
`
`Vdd
`
`OUT
`
`SNS1
`
`OPT1
`
`GAIN
`
`OPT2
`
`SNS2
`
`Vss
`
`8
`
`P O RT X .m
`
` C MO S
`
`m icro controller
`
`P O RT X .n
`
`O UT
`
`0 .01µF
`
`V dd
`
`Q T11 0
`
`V ss
`
`On-Duration expires, whichever occurs first. If the latter occurs
`first, the sensor performs a full recalibration and the output
`becomes inactive until the next detection.
`
`In this mode, two Max On-Duration timeouts are available: 10
`and 60 seconds.
`
`2.2.2 TOGGLE MODE OUTPUT
`This makes the sensor respond in an on/off mode like a flip
`flop. It is most useful for controlling power loads, for example in
`kitchen appliances, power tools, light switches, etc.
`
`Max On-Duration in Toggle mode is fixed at 10 seconds. When
`a timeout occurs, the sensor recalibrates but leaves the output
`state unchanged.
`
`Table 2-1 Output Mode Strap Options
`
`Tie
`Pin 3 to:
`
`Tie
`Pin 4 to:
`
`Max On-
`Duration
`
`DC Out
`
`DC Out
`
`Toggle
`
`Pulse
`
`Vdd
`
`Vdd
`
`Gnd
`
`Gnd
`
`Vdd
`
`Gnd
`
`Gnd
`
`Vdd
`
`10s
`
`60s
`
`10s
`
`10s
`
`2.2.3 PULSE MODE OUTPUT
`This mode generates a negative pulse of 75ms duration with
`every new detection. It is most useful for 2-wire operation, but
`can also be used when bussing together several devices onto
`a common output line with the help of steering diodes or logic
`gates, in order to control a common load from several places.
`
`Max On-Duration is fixed at 10 seconds if in Pulse output
`mode.
`
`Note that the beeper drive does not operate in Pulse mode.
`
`2.2.4 PIEZO ACOUSTIC DRIVE
`A piezo drive signal is generated for use with a piezo sounder
`immediately after a detection is made; the tone lasts for a
`nominal 95ms to create a ‘tactile feedback’ sound.
`
`The sensor drives the piezo using an H-bridge configuration for
`the highest possible sound level. The piezo is connected
`across pins SNS1 and SNS2 in place of Cs or in addition to a
`parallel Cs capacitor. The piezo sounder should be selected to
`have a peak acoustic output in the 3.5kHz to 4.5kHz region.
`
`Since piezo sounders are merely high-K ceramic capacitors,
`the sounder will double as the Cs capacitor, and the piezo's
`metal disc can even act as the sensing electrode. Piezo
`transducer capacitances typically range from 6nF to 30nF in
`value; at the lower end of this range an additional capacitor
`should be added to bring the total Cs across SNS1 and SNS2
`to at least 10nF, or possibly more if Cx is above 5pF
`
`Piezo sounders have very high, uncharacterized thermal
`coefficients and should not be used if fast temperature swings
`are anticipated, especially at high gains. They are also
`generally unstable at high gains; even if the total value of Cs is
`largely from an added capacitor the piezo can cause periodic
`false detections.
`
`The burst acquisition process induces a small but audible
`voltage step across the piezo resonator, which occurs when
`SNS1 and SNS2 rapidly discharge residual voltage stored on
`the resonator. The resulting slight clicking sound can be greatly
`reduced by placing a 470K resistor Rs in parallel with the
`resonator; this acts to slowly discharge the resonator,
`attenuating of the harmonic-rich audible step (Figure 2-3).
`
`Note that the piezo drive does not operate in Pulse mode.
`
`2.2.5 HEARTBEAT™ OUTPUT
`The output has a full-time HeartBeat™ ‘health’ indicator
`superimposed on it. This operates by taking 'Out' into a 3-state
`mode for 350µs once before every QT burst. This output state
`can be used to determine that the sensor is operating properly,
`or, it can be ignored using one of several simple methods.
`
`The HeartBeat indicator can be sampled by using a pulldown
`resistor on Out, and feeding the resulting negative-going pulse
`into a counter, flip flop, one-shot, or other circuit. Since Out is
`normally high, a pulldown resistor will create negative
`HeartBeat pulses (Figure 2-4) when the sensor is not detecting
`an object; when detecting an object, the output will remain
`active for the duration of the detection, and no HeartBeat pulse
`will be evident.
`
`If the sensor is wired to a microcontroller as shown in Figure
`2-5, the controller can reconfigure the load resistor to either
`ground or Vcc depending on the output state of the device, so
`that the pulses are evident in either state.
`
`Electromechanical devices will usually ignore this short pulse.
`The pulse also has too low a duty cycle to visibly activate
`LED’s. It can be filtered completely if desired, by adding an RC
`timeconstant to filter the output, or if interfacing directly and
`only to a high-impedance CMOS input, by doing nothing or at
`most adding a small non-critical capacitor from Out to ground
`(Figure 2-6).
`
`2.2.6 OUTPUT DRIVE
`The QT110’s output is active low ; it can source 1mA or sink
`5mA of non-inductive current.
`
`Care should be taken when the IC and the load are both
`powered from the same supply, and the supply is minimally
`regulated. The device derives its internal references from the
`power supply, and sensitivity shifts can occur with changes in
`Vdd, as happens when loads are switched on. This can induce
`detection ‘cycling’, whereby an object is detected, the load is
`turned on, the supply sags, the detection is no longer sensed,
`
`LQ 5
`
`QT110 R1.04/0405
`
`Petitioner STMICROELECTRONICS, INC.,
`Ex. 1015, IPR2021-01160, Page 5 of 12
`
`

`

`Figure 2-4
`Getting HB pulses with a pull-down resistor
`
`
`Figure 2-5
`Using a micro to obtain HB pulses in either output state
`
`H eart Be at™ P ulses
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`R o
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`2
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`3
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`4
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`+ 2 .5 to 5
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`1
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`V d d
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`O UT
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`S NS 2
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`O PT 1
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`G A IN
`
`O PT 2
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`S NS 1
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`V ss
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`8
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`7
`
`5
`
`6
`
`M icro pro ce sso r
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`P O RT _ M .x
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`R o
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`P O RT _ M .y
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`2
`
`3
`
`4
`
`O U T
`
`SN S 2
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`O P T 1
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`GA IN
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`O P T 2
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`SN S 1
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`7
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`5
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`6
`
`the load is turned off, the supply rises and the object is
`reacquired, ad infinitum. To prevent this occurrence, the output
`should only be lightly loaded if the device is operated from an
`unregulated supply, e.g. batteries. Detection ‘stiction’, the
`opposite effect, can occur if a load is shed when Out is active.
`
`The output of the QT110 can directly drive a resistively limited
`LED. The LED should be connected with its cathode to the
`output and its anode towards Vcc, so that it lights when the
`sensor is active-low. If desired the LED can be connected from
`Out to ground, and driven on when the sensor is inactive, but
`only with less drive current (1mA).
`
`to reduce stray loading (which will dramatically reduce
`sensitivity).
`
`2. Keep Cs, Rs, and Re very close to the IC.
`
`3. Make Re as large as possible. As a test, check to be sure
`that an increase of Re by 50% does not appreciably
`decrease sensitivity; if it does, reduce Re until the 50%
`test increase has a negligible effect on sensitivity.
`
`4. Do not route the sense wire near other ‘live’ traces
`containing repetitive switching signals; the sense trace will
`pick up noise from them.
`
`3 - CIRCUIT GUIDELINES
`
`3.1 SAMPLE CAPACITOR
`When used for most applications, the charge sampler Cs can
`be virtually any plastic film or good quality ceramic capacitor.
`The type should be relatively stable in the anticipated
`temperature range. If fast temperature swings are expected,
`especially at higher sensitivity, a more stable capacitor might
`be required for example PPS film.
`
`In most moderate applications a low-cost X7R type will work
`fine.
`
`3.2 ELECTRODE WIRING
`See also Section 3.4.
`
`The wiring of the electrode and its connecting trace is important
`to achieving high signal levels and low noise. Certain design
`rules should be adhered to for best results:
`
`1. Use a ground plane under the IC itself and Cs and Rs but
`NOT under Re, or under or closely around the electrode or
`its connecting trace. Keep ground away from these things
`
`Figure 2-6 Eliminating HB Pulses
`G AT E OR
`M IC RO INP U T
`
`C M O S
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`Co
`
`100p F
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`2
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`3
`
`4
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`O UT
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`SN S 2
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`O PT1
`
`GA IN
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`O PT2
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`SN S 1
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`7
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`5
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`6
`
`3.3 POWER SUPPLY, PCB LAYOUT
`See also Section 3.4.
`
`The power supply can range from 2.5 to 5.0 volts. At 2.5 volts
`current drain averages less than 10µA with Cs = 10nF,
`provided a 470K Rs resistor is used (Figure 2-6). Idd curves
`are shown in Figure 4-4.
`
`Higher values of Cs will raise current drain. Higher Cx values
`can actually decrease power drain. Operation can be from
`batteries, but be cautious about loads causing supply droop
`(see Output Drive, Section 2.2.6) if the batteries are
`unregulated.
`
`As battery voltage sags with use or fluctuates slowly with
`temperature, the IC will track and compensate for these
`changes automatically with only minor changes in sensitivity.
`
`If the power supply is shared with another electronic system,
`care should be taken to assure that the supply is free of digital
`spikes, sags, and surges which can adversely affect the
`device. The IC will track slow changes in Vdd, but it can be
`affected by rapid voltage steps.
`
`if desired, the supply can be regulated using a conventional
`low current regulator, for example CMOS LDO regulators that
`have nanoamp quiescent currents. Care should be taken that
`the regulator does not have a minimum load specification,
`which almost certainly will be violated by the QT110's low
`current requirement. Furthermore, some LDO regulators are
`unable to provide adequate transient regulation between the
`quiescent and acquire states, creating Vdd disturbances that
`will interfere with the acquisition process. This can usually be
`solved by adding a small extra load from Vdd to ground, such
`as 10K ohms, to provide a minimum load on the regulator.
`
`Conventional non-LDO type regulators are usually more stable
`than slow, low power CMOS LDO types. Consult the regulator
`manufacturer for recommendations.
`
`For proper operation a 100nF (0.1uF) ceramic bypass
`capacitor must be used between Vdd and Vss; the bypass cap
`
`LQ 6
`
`QT110 R1.04/0405
`
`Petitioner STMICROELECTRONICS, INC.,
`Ex. 1015, IPR2021-01160, Page 6 of 12
`
`

`

`should be placed very close to the device’s power pins.
`Without this capacitor the part can break into high frequency
`oscillation, get physically hot, stop working, or become
`damaged.
`
`extremely large amounts of nonlinear parasitic capacitance
`which will swamp the capacitance of the electrode and cause
`false detections and other forms of instability. Diodes also act
`as RF detectors and will cause serious RF immunity problems.
`
`PCB Cleanliness: All capacitive sensors should be treated as
`highly sensitive circuits which can be influenced by stray
`conductive leakage paths. QT devices have a basic resolution
`in the femtofarad range; in this region, there is no such thing as
`‘no clean flux’. Flux absorbs moisture and becomes conductive
`between solder joints, causing signal drift and resultant false
`detections or temporary loss of sensitivity. Conformal coatings
`will trap in existing amounts of moisture which will then become
`highly temperature sensitive.
`
`The designer should strongly consider ultrasonic cleaning as
`part of the manufacturing process, and in more extreme cases,
`the use of conformal coatings after cleaning and baking.
`
`3.3.1 SUPPLY CURRENT
`Measuring average power consumption is a challenging task
`due to the burst nature of the device’s operation. Even a good
`quality RMS DMM will have difficulty tracking the relatively slow
`burst rate, and will show erratic readings.
`
`The easiest way to measure Idd is to put a very large capacitor,
`such as 2,700µF across the power pins, and put a 220 ohm
`resistor from there back to the power source. Measure the
`voltage across the 220 resistor with a DMM and compute the
`current based on Ohm’s law. This circuit will average out
`current to provide a much smoother reading.
`
`To reduce the current consumption the most, use high or low
`gain pin settings only, the smallest value of Cs possible that
`works, and a 470K resistor (Rs) across Cs (Figure 1-1). Rs
`acts to help discharge capacitor Cs between bursts, and its
`presence substantially reduces power consumption.
`
`3.3.2 ESD PROTECTION
`In cases where the electrode is placed behind a dielectric
`panel, the IC will be protected from direct static discharge.
`However even with a panel transients can still flow into the
`electrode via induction, or in extreme cases via dielectric
`breakdown. Porous materials may allow a spark to tunnel right
`through the material. Testing is required to reveal any
`problems. The device has diode protection on its terminals
`which will absorb and protect the device from most ESD
`events; the usefulness of the internal clamping will depending
`on the dielectric properties, panel thickness, and rise time of
`the ESD transients.
`
`The best method available to suppress ESD and RFI is to
`insert a series resistor Re in series with the electrode as shown
`in Figure 1-1. The value should be the largest that does not
`affect sensing performance. If Re is too high, the gain of the
`sensor will decrease.
`
`Because the charge and transfer times of the QT110 are
`relatively long (~2µs), the circuit can tolerate a large value of
`Re, often more than 10k ohms in most cases.
`
`Diodes or semiconductor transient protection devices or MOV's
`on the electrode trace are not advised; these devices have
`
`3.4 EMC AND RELATED NOISE ISSUES
`External AC fields (EMI) due to RF transmitters or electrical
`noise sources can cause false detections or unexplained shifts
`in sensitivity.
`
`The influence of external fields on the sensor is reduced by
`me