Challenges
`in Portable
`RF
`Transceiver
`Design
`
`Behzad Razavi
`
`As wireless products such as cellular phones become an
`
`everyday part of people's lives, the need for higher
`performance at lower costs becomes even more impor(cid:173)
`tant. Overcoming the challenges involved in the design
`of radio-frequency (RF) transceivers can help meet this need. This
`article provides an overview of RF electronics in portable transceivers
`and describes design issues as well as current work toward achieving
`both high performance and low cost. To understand the implications
`in the design of .{{F integrated circuits (I Cs) we look at the properties
`of the mobile communications environment. We then study receiver
`and transmitter architectures and their viability in present IC technolo(cid:173)
`gies. An example of an RF transceiver is given and the design of
`transceiver building blocks is discussed. We conclude by looking at
`future directions in RF design.
`
`Wireless Communication Development
`Wireless technology came _to existence in 1901 when Guglielmo
`Marconi successfully transmitted radio signals across the Atlantic
`Ocean. The consequences and prospects of this demonstration were
`simply overwhelming; the possibility of replacing telegraph and tele- ~
`phone communications with wave transmission through the "ether" ~
`• portrayed an exciting future. However, while two-way wireless com-
`g>
`§
`munication did soon materialize in the military, wireless transmission
`.,
`in daily life remained limited to one-way radio and television broad- ~
`casting by large, expensive stations. Ordinary, two-way.phone con-
`versations would still go over wires for many decades. The invention j
`.!!!
`·ffi
`of the transistor, the development of Shannon's information theory,
`all at Bell Laboratories ~
`and the conception of the cellular system -
`
`8755-3996/96/$5.00©1996IEEE
`
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`

`paved the way for affordable mobile
`-
`communications, as originally implemented
`in car phones and eventually realized in
`portable cellular phones (cell phones).
`But, why the sudden surge in wireless
`electronics? Market surveys show that in the
`United States more than 20,000 people join
`the cellular phone system every day, moti(cid:173)
`vating competitive manufacturers to provide
`phone sets with increasingly higher perform(cid:173)
`ance and lower cost. In fact, the present goal
`is to reduce both the power consumption and
`price of cell phones by 30% every year -
`although it is not clear for how long this rate
`can be sustained. A more glorious prospect,
`however, lies in the power of two-way wire(cid:173)
`less communication when it is introduced in
`other facets of our lives: home phones, com(cid:173)
`puters, facsimile, and television.
`While an immediate objective of the
`wireless industry is to combine cordless and
`cellular phones to allow seamless commu-
`nications virtually everywhere, the long(cid:173)
`term plan is to produce an "omnipotent"
`wireless terminal that can handle voice, data,
`and video as well as provide computing
`power. Other luxury items such as the global
`positioning system ( GPS) are also likely to
`become available through this terminal
`sometime in the future. Personal communi(cid:173)
`cation services (PCS) are almost here.
`Today's pocket phones contain more
`than one million transistors, with only a very
`small fraction operating in the RF range and
`the rest performing low-frequency baseband
`signal processing. However, the RF section
`is still the design bottleneck of the entire
`system. This is primarily for three reasons.
`First, while digital circuits directly benefit
`from advances in integrated-circuit (IC)
`technologies, RF (analog) circuits do not
`benefit as much because they suffer from
`many more trade-offs and often require ex(cid:173)
`ternal components (such as inductors) that
`are difficult to bring onto the chip even in
`modem fabrication processes. Second, in
`contra,! to other types of analog circuits,
`proper RF design demands a solid under(cid:173)
`standing of many areas that are not directly
`related to integrated cirrnits, e.g., micro(cid:173)
`wave theory, communication theory. analog
`and digital modulation, transceiver architec(cid:173)
`tures. etc. Each of these disciplines h«s heen
`t111Ller developruenL for many decades, mak(cid:173)
`ing it difficult for an lC designer to acquire
`the necessary knowledge in a short time.
`Third, computer-aided analy,is and ,ynthe-
`sis toob for Rr an: still in their infancy,
`
`September 1996
`
`1. Simple RF front end.
`
`2. Effect of third-order nonlinearity in LNA.
`
`3. Definition of third-order intercept point.
`
`13
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`

`

`4. Simple homodyne receiver.
`
`forcing the designer to rely on .:xpcriencc
`and intuition t<J predict the performance. For
`these reasons, RF IC designers ha Ye bern a
`rare species.
`
`Wireless Enyironment
`The wireless communications envim111nent,
`especially in urban areas, is often called
`'·hostile'' because it imposes severe con(cid:173)
`straints 11po11 the tra11sceiYerde.sig11. Perhaps
`the most important constTaint is the limited
`spectrum allocated by regulatory orgcmiza
`tions to ,vircless users. From Shannon's
`theorem, this transJatcs to a limited rate of
`in/'onnati"n. 111anclating tlic, w,e of' sopliisti(cid:173)
`cated tedmiqttes such as coding, compres(cid:173)
`sion. and bandwidm-efficient modulation.
`even for voice signals.
`The narrow bandwiclth available- to each
`user also impacts the design of the RF front
`end. As depicted in Fig. l, the transmitter
`mnst employ narrowh111d arnplific:alion and
`filtering to avoid ·'leakage" Lo adjacent
`bands. :md the receiver must be able to proc(cid:173)
`ess the desired channel while sufficiently
`rejecring ,lrnng 11eighhoring channel,s. Tu
`gain a better feeling aho111 the lalter i,,ue,
`we note that if the front-end bandpass filter
`(BI'O in a 900-MHz receiver is to provide
`60 dB ofrc:jection at4'i kHz from the ,cme,·
`of the channel, then the equivalent Q of the
`filter is on the order of 10 7 , a value difficult
`to achic,·e even in surface acoustic WJYc
`(SAW) filters. Since typical filters exhibit a
`trnde-,Jff between the loss one! tl1e Q aml
`since in receiYing very small signals the loss
`must be minimized, the out-of-channel re(cid:173)
`jection of the from-end filters is usually
`insufficient, requiring further filtering in the
`following stages (typically at lower center
`frequencies). This will be clarified later in
`this article.
`The existence of large unwanted signals
`in the vicinity of the band of interest even
`after filtering creates difficulties in the de(cid:173)
`sign of the following circuits, in particular
`the front-end low-noise amplifier (LNA).
`As shown in Fig. 2, if the LNA exhibits
`nonlinearity, then the "intermodulation
`
`14
`
`products" of t,vo strong nnwanted signals
`may appear in the desired band, thereby
`corrupting the reception. As a simple exam(cid:173)
`ple, we note that if the input/output static
`charaderi,tic of (be LNA is approximate,]
`as y(I) = ap(f) +
`+ u3}(l) aucl x(t)
`= A1cos(l)11 + A2cosrn:1, then the cubic term
`yielcl., components at 2ro1 - rn2 and 2co2 -
`m1. either of which may fall in the haml. The
`standard approach to guantifyinp; this effect
`is to choose A 1 =A2 and, using ex.Lrapnlalion,
`calculate the input power that results in
`equal mngnitudcs for the fundan,ental com(cid:173)
`ponents and the imem10dulation products
`(rig. 1). Such \·aluL, of inpllt power is called
`the "'third-order intercept point'' (/P1). lt is
`interesting Lo note that this type ofuonlinear(cid:173)
`ity is important ewn if the signal carries
`
`5. Homodyne receiver with quadrature dovvn(cid:173)
`conversion.
`
`information in its phase or frequency rather
`than in its amplitude.
`Another impmiant issue in the design of
`wireless receivers is the dy11amic range of
`the input signal. Typically around 100 dB ( a
`factor of J 00,000 for voltage quantities), the
`clym1mic: rage i, lin,ikcl by a lower bound
`due to noise and an upper bound due to
`nonlinearities and saturation. The minimum
`detectable .signal in today's h:mdscts is in the
`vicinity of -110 dBm (=0.71 ,uVrms in a
`50-Q system), thus derna11ding very low
`noise in the receive path. For the upper
`bound, the ncceivc,· must achieve a high
`
`6. LO leakapo to input.
`
`7. Effect of second-order distortion.
`
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`

`linearity so as to minimize intermodulation
`products. Also, saturation effects at high
`input levels often mandate the use of gain
`control in various parts of receivers.
`
`Receiver Architectures
`Complexity, cost, power dissipation, and the
`number of external components have been
`the primary criteria in selecting receiver ar- ·
`chitectures. As IC technologies evolve, ar(cid:173)
`chitectures that once seemed impractical
`may return because, when they are imple(cid:173)
`mented in today's advanced processes, their
`advantages outweigh their drawbacks.
`
`Homodyne Architecture
`Also called "direct conversion" architec(cid:173)
`ture, the homodyne receiver is the natural
`topology for downconverting a signal from
`RF to baseband. The idea is simply to mix
`the RF signal with a local oscillator (LO)
`output and low-pass filter the result such that
`the center of the band of interest is translated
`directly to zero frequency (Fig. 4 ). Because
`of its typically high noise, the mixer is usu(cid:173)
`ally preceded by an LNA. Also, in phase and
`frequency modulation schemes, the RF sig(cid:173)
`nal is mixed with both the LO output and its
`quadrature so as to provide phase informa(cid:173)
`tion (Pig. 5).
`The simplicity of the homodyne archi(cid:173)
`tecture makes it attractive for compact, effi(cid:173)
`cient implementation of RF receivers [1, 2].
`However, several issues have impeded its
`widespread use. We briefly describe these
`issues and their impact on the design of
`related ICs.
`DC Offsets. Since in a homodyne re(cid:173)
`ceiver the downconverted hand extends to
`the vicinity of the zero frequency, extrane(cid:173)
`ous offset voltage, can corrupt the signal
`and, more importantly, saturate the follow(cid:173)
`ing stages. To understand the origin and
`impact of offsets, consider the more realistic
`circuit shown in Fig. 6. Here, the mixer is
`followed by a low-pass filter, a post-ampli(cid:173)
`fier, and an ·analog-to-digital converter
`(ADC). We make two observations: (1) The
`isolation bet ween the LO and RF ports of the
`mixer is not perfect; due to capacitive cou(cid:173)
`pling and, if the LO signal is supplied exter(cid:173)
`nally, bond wire coupling, a finite amount of
`feedthrough exists from the LO port to
`points A and B. This effect is called "LO
`leakage." The leakage signal appearing at
`the input of the LNA is amplified and mixed
`with the LO signal, thus producing a DC
`component at point C. This phenomenon is
`
`September 1996
`
`8. Heterodyne architecture.
`
`9. (a) Problem of image, (b) image rejection by filtering.
`
`15
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`

`c;allcd "sel r-mixi11g." (2) The total gain Crom
`the antenna to point X is typically a.round
`100 dB so tbal the microvolt i11pnl signal
`reaches a level that can be digitized by a
`low-cost ADC. Of this gain, approximately
`
`25 lo 3() dB is contsibuted by the Ll\A/rnixer
`comhinatinn.
`\Vith the above observations and noting
`that the LO power is typically around O dBm
`(approximately O.b Yrr), and the LO leak-
`
`+}
`
`·i-J o,a
`
`0
`
`•o)
`
`Desired
`Image Channel
`
`\
`I
`JliJL .~JI--~
`
`0
`
`0)
`
`►t--
`
`SlllWLol
`
`COSC)Lof
`
`I
`
`14,16.
`
`0
`
`(()
`
`LPFl---'-B __ _,
`
`··-0 ,M,.
`_jl_Q___
`
`O
`
`co
`
`1 o. Image re1ection using single-sideband mixing.
`
`Q
`
`U.1
`
`+j
`+j
`_Q_,ll_-
`0
`
`6)
`
`1. Weaver architecture.
`
`----- - - - - - -
`
`- - - - - - - - - - - - - - - - - - - -~
`
`PA
`
`BPF
`
`12. Direct conversion transmitter.
`
`16
`
`age to point A on the order of -60 dB, we
`infer that the DC component at the output or
`the mixer due to self-mixing iii roughly
`eqt1al to 0 dBm -60 dB+ 30 dB= -30 dBm,
`corresponding: to a level of 10 m V. We also
`note that the ,ignal !Gvel at this point can be
`as low as 25µ Vrrns- Thus, if directly ampli(cid:173)
`fied hy the remaining gain of70 dB, the DC
`component saturates the following circuits,
`prohibiting the amplification of the desired
`signal.
`While high-pass filtering (i.e., AC cou(cid:173)
`pling) mav seem the solution here. in most
`cir lo,lay', rnudulation schemes the speclrum
`contains information at frequencies as lO\v
`as a few tens of hertz, mandating a very low
`corner frequency in the filter. In addition t0
`difficulties in implementing such a filter in
`TC form, a more fumlamemal problem is ils
`slow response, an important issue if the off(cid:173)
`set varies quickly. This occurs, for example.
`,vhe11 a cm mo1-cs ata liiglup0ed and the LO
`leahtge reflectjons rrom the surrnum]ing ob(cid:173)
`jects change the offset rapidly.
`For tlie,e reasons, hornodyne receivers
`require sophisticaled offscl-cancellation
`techniqncs. ln [3 J. for example, the offset in
`the analog signal path is red ucecl by feeding
`infonnation from the baseband digital signal
`processc,r (DSP). Alternatively, rnodu lati<Jn
`schemes can be sought that contain negli(cid:173)
`gible energy below a few kilohe1iz l4J.
`Even-Order Distortion. While lhinl(cid:173)
`order mixing was considered as a source of
`interference in Fig. 2, en:n-ordcr disto1iion
`also become, prnblernalic· ili h(,modyne
`downconYersion. As depicted in Fig. 7, if
`two strong interferers close to the channel of
`intere.\l experience a nonlinearity web as
`_v(r) = a, x(1) + U? x\t). then they are trans(cid:173)
`lated to a luw frequenq before lhe mixing
`operation and the result passes th.rough the
`1nixer with finite attenuati,)n. This i, be(cid:173)
`cause, in the presence of nlismatches Lhat
`degrade the symmetry of the llllXer. the mix(cid:173)
`ing operation can be viewed as x(t)(a + A
`cos Wt), indicating that a fraction of x(t)
`appears at the output without frequency
`translation. A similar effect occurs if the LO
`output duty cycle deviates from 50%. An(cid:173)
`other issue is that the second harmonic of the
`input signal ( due to the square term in the
`above equation) is mixed with the second
`harmonic of the LO output, thereby appear(cid:173)
`ing in the baseband and interfering with the
`actual signal [5]. For these reasons, even-or-
`
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`

`der intermodulation corrupts the baseband
`signal.
`1-Q Mismatch. As mentioned above,
`in most phase and frequency modulation
`schemes the downconvcrsion path mu,L
`use quadrature mixing. The required I and Q
`phases of the LO raise an issue related to the
`mismatches between these two signab. If
`the amplitudes of the I and Q ontputs are
`not equal or their phase difference deviates
`from 90°, the error rate in detecting the
`baseband signal rises. The task of generating
`I and Q phases with precise matching is
`discussed later.
`A second-order effect arises from the
`mismatches between the two mixers them(cid:173)
`selves. Since the mixers process high-fre(cid:173)
`quency signals here, their phase and gain are
`sensitive to parasitics and hence susceptible
`to mismatches.
`LO Leakage. In addition to introducing
`DC offsets, leakage of the LO signal to the
`antenna and radiation tl1erefrom creates in(cid:173)
`terference in the band of other receivers. The
`design of the wireless infrastructure and the
`regulatiom of the Federal Communications
`Commission (FCC) impose upper bounds
`on the amount of LO radiation, typically
`between -60 to -80 dBm.
`Flicker Noise. Owing to the limited gain
`provided hy the LNA and the mi>-.er, the
`downconvcrtcd signal is relatively small and
`quite sensitive to noise. Since device flicker
`noise becomes significant at low frequen-
`
`13. Disturbance of VCO by PA in direct conversion transmitter.
`
`14. Alternative transmitter architectures. (a) Two-step conversion, (b) Offset VCO.
`
`(b)
`
`WL01
`
`Frequency
`Synthesizer
`
`Duplexer
`Filter
`
`Matching
`Network
`
`LPF
`
`DAC
`
`Oversampled
`I
`
`LPF
`
`DAC
`
`Oversampled
`Q
`
`15. Representative RF transceiver.
`
`September 1996
`
`17
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`

`cies, amplification of the baseband signal
`with low noise is an important issue.
`
`Heterodoyrw An·/,ifpc/urr.1
`The design issues mentioned above for the
`homodyne receiver have motivated the in­
`vention of other architectures. Most com­
`monly used is the heterodyne topology. (In
`this article, we do not make a distinction
`between "heterodyne" and "superhetero­
`dyne.") lllustrated in Fig. 8 in a simple form,
`a heterodyne receiver first downeonverts the
`input to an "intermediate frequency" (II7).
`The resulting signal is subsequently band­
`pass filtered, ;m1plified, and downconverted
`again. In the case of digital modulation, the
`last downconversion generates both I and Q
`phases of the ,ignal.
`The heterodyne architecture alleviates
`two of the homodyne reception issues by
`avoiding them at high frequencies or low
`signal levels. The effect of DC offsets of the
`first few stages is removed by bandpass
`filtering, and that of the lasl stage is sup­
`pressed by the total gain in the preceding
`stages. Abo, I and Q mismatches occur at
`much lower frequencies and are therefore
`easier to control and correct. As for the LO
`leakage, since WL0I is ont of the band of
`interest, it is suppressed by the front-end
`BPF and its radiation from the antenna is less
`objectionable.
`Perhaps the most important feature of the
`heterodyne receiver is its selectivity, i.e., the
`capability to process and select small signals
`in the presence of strong interferers. While
`selecting a 30-kHz channel at a center fre­
`quency of 900 MH, ncquircs prohibitively
`large Qs, in Fig. 8 bandpass filtering is per­
`formed at progre,sively lower center fre­
`quencies. For example. the third BPF may
`operate at a center freguency of 400 kHz,
`thereby providing high selectivity for a 30-
`kHz channel. In other words, the filters have
`much more relaxed requirements.
`Despite the above merits, heterodyning
`entails a number of drawbacks. The most
`significant problem is the "image fre­
`quency." Since a simple mixer does not pre­
`serve the polarity of the difference between
`its input frequencies, it translates the bands
`both above and below the carrier to the s=e
`freqnency [Fig. 9(a)]. Thus, the mixing op­
`eration must be preceded by an "image re­
`ject" filter [Fig. 9(b)], usually a passive one.
`The issue of image rejection leads to an
`interesting trade-off among three parame­
`ters: the aniount of image noise, the spacing
`
`18
`
`between the band and the image (= 2 IF),
`and the loss of the filter. To minimize the
`image noise. we can either increase the TF
`(so thaL the filter provides more attenuation
`at the image frequency) or tolerate greater
`loss in the filter while increasing its Q. Since
`the LNA gain is typically less than 15 dB,
`the filter loss should not exceed a few dB,
`and the trade-off reduces to one between the
`image noise and the value of IF.
`How high can the IF be? Recall from Fig.
`8 that the filter following the first mixer must
`select the band. As the IF and, hence, the
`center frequency of this filter increase, so
`does the required Q, thereby imposing a
`fundamental trade-off between.image rejec­
`tion and ch::mnel selection. For the 900-MHz
`;md l .8-GHz hands, typical IFs range from
`70 :\IHz lo 200 MHz.
`Another drawback of the heterodyne ar­
`chitecture is that the LNA must drive a 50-Q
`impedance because the image-reject filter
`cannot be integrated and is therefore placed
`off-chip. This adds another dimension to the
`trade-offs among noise, linearity, gain, and
`power dissipation of the amplifier, further
`complicating the design. The image-reject
`and channel-selecl filters are typically ex­
`pensive ::md bulky, making the heterodyne
`approach less attractive for small, low-cost
`wireless tennina\s. Nevertheless, hetero­
`dyning has been the dominant choice for
`many decades [6, 7].
`
`Image-Reject Architectures
`The issues related to the image-reject filter
`have motivated RF designers to seek other
`techniques of rejecting the image in a l1et­
`erodyne receirnr. One such technique origi­
`nates from a single-sideband modulator
`intwdllced by Ralph Hartley in 1928 [8}.
`Illustrated in Fig. 10, Hartley's circuitrcixes
`the RF input with the quadrature outputs of
`the local oscillator. low-pass filters the re­
`sulting signals, and shifts one by 90 ° before
`adding them together. The reader can easily
`verify that if the input is equal to
`,4RFCOSWRFt+A,coso)]t, where WI is the im­
`age frequency, then the output is propor­
`tional to ARFCos(rnu.-c,lRF)I. As a more
`general case, we consider the input spectrum
`shown in Fig. 10 and note that mixing with
`sincoLot and cosCOL0I yields the ,pectra of
`Fig. 10 at nodes A and B, respectively (the
`factor ±j in these spectra is to indicate con­
`volution with ±jo(CO±CULo)/2 (spectrum of
`sinwwt) ). Since a phase shift of +90 ° in the
`
`signal at A corresponds to multiplication by
`+j and inverting the positive frequencies, we
`obtain the four spectra at nude, B ancl C as
`the inputs to the adder. The output is there­
`fore free from the image.
`The principal drawback of image-reject
`mixers is their sensitivity to mismatches. Frn
`ex=ple, if the phase difference between the
`LO quadrature phases deviates from 90 °, the
`cancellations shown in Fig. 10 are imperfect
`and some image noise corrupts the down-
`
`Noise ----Power
`
`Linearity
`
`Frequency
`
`/
`\
`I
`\ Supply - Gain
`
`Voltage
`
`16. RF design hexagon.
`
`. �n o---j
`
`;
`
`;;
`
`17. Low-noise amplifier.
`
`Vee
`
`18. Gilbert mixer.
`
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`

`-r---T- Vee
`
`19. Single-balanced mixer.
`
`converted signal [5]. For typical matching in
`IC technologies, the image is rejected by
`about 30 to 40 dB [9]. Another important
`issue is the higher power dissipation and/or
`noise due to the use of two high-frequency
`mixers. Also, circuits that shift the down(cid:173)
`converted signal by 90° generally suffer
`from trade-offs among linearity, noise, and
`power dissipation.
`
`Weaver Architecture
`In our discussion of images, we noted that
`any frequency translation leads to corrup(cid:173)
`tion of the signal by the image, except when
`a symmetric band is brought down to zero
`frequency (homodyne). The Weaver tech(cid:173)
`nique allows an arbitrary translation of the
`signal band without image interference [1 OJ.
`Illustrated in Fig. 11, this approach down(cid:173)
`converts the signal in two steps. In the first step,
`the input is mixed with the quadrature phases
`of the first local oscillator and the result is
`low-pass filtered, yielding the spectra at nodes
`A and B. In the second step, these signals are
`translated to zero frequency and added to(cid:173)
`gether, thereby effecting image cancellation.
`The important advantage of the Weaver
`architecture is that it does not require high-Q
`bandpass filters. Even though the LPFs shown
`in Fig. l l(a) must be preceded by capacitive
`coupling to eliminate DC offsets (similar to
`homodyne) and. as such, the combination is a
`bandpass filter, the out-of-band rejection of
`these filters is quite relaxed. Note that some
`amplification is necessary before the second
`set of mixers to reduce the effect of their noise.
`The Weaver method suffers from the same
`drawback as the image-reject mixer: incom(cid:173)
`plete cancellation of the image in the pres(cid:173)
`ence of mismatches.
`
`September 1996
`
`Transmitter Architectures
`In contrast to the variety of approaches in(cid:173)
`vented for RF reception, transmitter archi(cid:173)
`tect\ires are found in only a few forms. This
`is because issues such as image rejection and
`band selectivity are more relaxed in trans(cid:173)
`mitters, leaving the output power amplifier
`(PA) design as the primary challenge.
`A simple direct conversion transmitter is
`shown in Fig. 12. Here, the baseband signal
`is mixed with the LO output and the result
`is bandpass filtered and applied to the PA. A
`matching network is usually interposed be(cid:173)
`tween the PA and the antenna to allow maxi(cid:173)
`mum power transfer and filter out-of-band
`components that result from nonlinearities
`in the amplifier. Note that since the base(cid:173)
`band signal is produced in the transmitter
`and is therefore sufficiently strong, tlie noise
`of the mixers is much less critical here than
`in receivers.
`Direct conversion architectures suffer
`from an important drawback: disturbance of
`the transmit local oscillator by the output
`PA. Illustrated in Fig. 13, this issue arises
`because the PA output is a modulated wave(cid:173)
`form with high power and a spectrum cen(cid:173)
`tered around
`the voltage-controlled
`oscillator VCO frequency. Thus, despite
`various shielding techniques that attempt to
`isolate the VCO, the "noisy" output of the
`PA still corrupts the oscillator spectrum.
`(The actual mechanism of this corruption is
`
`called "injection pulling" or "injection lock(cid:173)
`ing." When disturbed by a close interferer at
`frequency ro;, an oscillator operating at roo
`tends to shift to co;.) This problem worsens
`if the PA is turned on and off periodically to
`save power.
`The above difficulty is alleviated if the
`PA output spectrum is sufficiently higher or
`lower than the VCO frequency. For exam(cid:173)
`ple, as shown in Fig. l 4(a), the upconversion
`can be performed in two steps, generating a
`final spectrum that differs from CO2 by co1
`[11]. Alternatively, the VCO frequency can
`be "offset" by adding or subtracting the out(cid:173)
`put frequency of another oscillator (Fig.
`14(b)) [7]. Note that in both cases, some
`filtering is required to reject unwanted parts
`of the spectrum.
`The most difficult part of transmitters to
`design is the PA, mainly because of severe
`trade-offs among its efficiency, linearity,
`and supply voltage. In typical PA topolo(cid:173)
`gies, the efficiency drops as the circuit is
`designed for higher linearity or lower supply
`voltage. For a typical peak output power of
`1 W, an efficiency of 50% means that an
`additional I Wis wasted, which is a substan(cid:173)
`tial amount with respect to the power dissi(cid:173)
`pation of the rest of a portable phone.
`The reader may wonder why the linearity
`of the PA is important if only the phase of
`the carrier is modulated. Indeed in analog
`
`Ideal Oscillator
`
`Actual Oscillator
`
`20. Phase noise and sidebands in the output of oscillators.
`
`vco r----.~ four
`
`21. Pulse swallow synthesizer.
`
`19
`
`ParkerVision Ex. 2011
`Intel Corp. v. ParkerVision, Inc.
`IPR2020-01265
`
`

`

`J-iJ\1 systems, the linearity is not critical and
`the efficiency trades only with the supply
`voltage, usually approaching 60% at the
`peak output power. On the othe>r hand, in
`digital modulation schemes such as quadra(cid:173)
`ture phase shift keying ( QPSK) the situation
`is more> rnmplirntecl. Since a QPSK signal
`has a relatively wide spectrum, il usually
`undergoes bamlpass filtering to limit its
`bandwidth lo that of one channel. Tbe result(cid:173)
`ing signaL however, does not have a con(cid:173)
`stant envefope, i.e .. it exhibits s(1111e
`amplitude modulation. Now, if this signal
`experiences nonlinear amplification, its
`spec:lrutn widens, spilling into adjacent
`channels and defeating the purpose of band(cid:173)
`pass filtering.
`
`In order to resoh·e this issue, RF system
`designers ha\·e ernployeLI l\Ao cliffereu!
`strategics. First. they have found digital
`modulation schemes in which the envelope
`of the signal remains constant after filtering
`and, hence, the spectrum does not widen in
`the presenc:c of PA nonlinearities. These
`schemes arc !mown as ·'continuous phase
`modulation," where 1he phase ,>f the canier
`varie.s smPothly from one bit to tl1e uexl.
`Second. they han) devi,cd feedback and
`feedforward circuit techniques to improve
`the linearity of P As with negligible degrada(cid:173)
`tion in efficiency [13, 14, 21].
`
`Overall System
`\Vith the above discussion of !rnnsceiver
`architectures, we can now consider a more
`
`fREF -- EJ- vco
`
`fcut
`
`Pulse
`Rernove1
`
`y
`
`X
`
`IH.i--
`
`Channel
`Select
`
`(a)
`
`~
`
`T
`
`..
`
`(0
`
`i 1 i
`..
`..
`
`2
`T
`(b)
`
`Vx
`
`Vy
`
`Vy
`
`Reference
`Input
`
`LP F Output !-
`0
`
`vco
`Output
`
`22. (a) Fractional-N synthesizer, (b) problem of reference sidebands.
`
`20
`
`complete system. Shown in Fig. 15 is a
`transceiver with lteterndyning in rhe recei\e
`path ~nd direct conwrs.ion in the transmit
`path. The transmit VCO may employ the
`offset technique of Fig. 14(b) to avoid injec(cid:173)
`tion pulling.
`In most mobile phone systems, the trans(cid:173)
`mit and receive bands are different, with the
`translation pe1formed at the base station. Tn
`a full-duplex syslem (where reception and
`transmission occur simultaneously 1hrough
`a single antenna), this is necessary because
`the two paths must be somehow separated.
`Witl1 lwo differem bawls, lhis is al'.com(cid:173)
`plishcd by a narrowband front-end filter,
`called the "duplexer." This filter also sup(cid:173)
`presses Plll-ol'-haml noise and interference
`in the recci vc path.
`ln Fig. 15, the receive and transmit
`LOs are embedded in a frequency synthe(cid:173)
`sizer. When iniliating a call, a lfalbile unil
`is assigned t.v. o cornmunicatioll channels
`(for receive and transmit) by the base sta(cid:173)
`tion. The synthesizer selects the proper
`carrier frequency for each channel accord(cid:173)
`ing lo a digit,11 inpul. The imporlanl issues
`here are how "pure" the synthesizer output
`is, and how fast can it can switch the LO
`frequency from ,me channel lo another.
`We relllrn to th<c:se issues in the section on
`frequency synthesizers.
`ln the recei,·e path, the downconverted
`signal is applied to an ADC. The ADC i~
`necessary e,en if the infonnalion lies in t11e
`phase (or frequency), because baseband op(cid:173)
`erations such as equalization, matched filter(cid:173)
`ing. and despreading are performed wirh
`higher precision in the digital domain than
`in the analog domain. Digital sigml proces(cid:173)
`sor, have thu, become an integral part <>f
`wireless transceivers.
`In the transmitter, the digitized voice un(cid:173)
`dergoes compression and coding. The re(cid:173)
`sulting stream of O1\Es and 7:EROs is
`subsequently oversampled aud subdivided
`into multi-bit words, which are then applied
`to two digital-to-analog converters (DACs)
`(Fig. 15). This operation takes place for an
`interesting reason. In digital modulation
`schemes, the ideal pulse shape for each bit
`produced in the baseband is quite different
`from a rectangular function. For example, as
`mentioned earlier, the modulated carrier
`may need to be such that its envelope re(cid:173)
`mains constant after filtering. Thus, it is
`usually necessary to convert the rectangular
`pulses to another shape. Furthermore, the
`bandpass filtering required after modulation
`
`Circuits & Devices
`
`ParkerVision Ex. 2011
`Intel Corp. v. ParkerVision, Inc.
`IPR2020-01265
`
`

`

`The very low noise required of the LNA
`usually mandate:, the use of only one active
`device at the input without any (high--fre(cid:173)
`quency) resistive feedback. In order to pro(cid:173)
`vide sufficient gain while driving 50 Q,
`LNAs typically employ more than one stage.
`An interesting example is shown in Fig. 17
`[15 J, where the first stage utilizes a bond(cid:173)
`wire inductance of 1.5 nH to degenerate the
`common-emitter amplifier without intro(cid:173)
`ducing additional noise. This technique both
`linearizes the LNA and makes it possible to
`achieve a 50-Q input impedance. Bias volt(cid:173)
`ages ht and Vb2 and the low-frequency
`feedback amplifier A 1 arc chosen so as to
`stabilize the gain against temperature and
`supply variations. The circuit exhibits a
`noise figure of 2.2 dB, an IP3 of -10 dBm,
`and a gain of 16 dB at 900 MHz.
`The issue of linearity becomes more sig(cid:173)
`nificant in mixers because they must handle
`signals that are amplified by the LNA. While
`ii may seem that the issue of noise is relaxed
`by the same factor. in practice, (active) mix(cid:173)
`ers exhibit much higher noise simply be(cid:173)
`cause they employ more devices in the
`signal path than do LNAs and suffer f