United States Patent (19)
`Tsujimoto
`
`54) TIME DIVERSITY TRANSMISSION
`RECEPTION SYSTEM
`
`75 Inventor: Ichiro Tsujimoto, Tokyo, Japan
`73 Assignee: NEC Corporation, Tokyo, Japan
`
`21 Appl. No.:719,983
`22 Filed:
`Sep. 24, 1996
`30
`Foreign Application Priority Data
`Oct. 23, 1995
`JP
`Japan .................................... 7-274330
`(51) Int. Cl. ................................................. H04B 7/02
`52 U.S. Cl. .......................... 375/200; 375/267; 370/441;
`455/103; 455/137
`58 Field of Search ..................................... 375/200, 202,
`375/203, 206, 207, 208, 260, 267, 299,
`347, 367; 370/441, 515; 455/102, 103,
`137, 139, 273, 276.1
`
`56)
`
`References Cited
`
`U.S. PATENT DOCUMENTS
`5,335,359 8/1994 Tsujimoto et al. ...................... 455/506
`5,349,609 9/1994 Tsujimoto .........
`... 375/347
`5,467,367 11/1995 Izumi et al. ..
`... 375/206
`5,524,023 6/1996 Tsujimoto ............................... 375/232
`5,550,811 8/1996 Kaku et al. ............................. 370/342
`5,568,523 10/1996 Tsujimoto ...
`... 375/347
`5,594,754
`1/1997 Dohi et al. ...
`... 375/200
`5,596,601
`1/1997 Bar-David .
`... 375/207
`5,598,428
`1/1997 Sato ......
`... 375/206
`5,621,769 4/1997 Wan et al. ........
`... 375/347
`4/1997 Bar-Dadid et al.
`5,623,511
`... 375/207
`... 375/200
`6/1997 Tsujimoto ............
`5,636.242
`5,646,964
`7/1997 Ushirokawa et al.
`... 375/347
`5,652,765
`7/1997 Adachi et al. .......................... 375/211
`
`USOO585987OA
`Patent Number:
`11
`(45) Date of Patent:
`
`5,859,870
`Jan. 12, 1999
`
`5,675,608 10/1997 Kim et al. ............................... 375/208
`5,687,162 11/1997 Yoshida et al.
`370/203
`5,692,006 11/1997 Ross ............
`... 375/200
`5,692,018 11/1997 Okamoto ................................. 375/347
`5,757,853 5/1998 Tsujimoto ............................... 375/200
`FOREIGN PATENT DOCUMENTS
`63-286027 11/1988 Japan.
`3284.011 12/1991 Japan.
`697914 4/1994 Japan.
`Primary Examiner Young T. Tse
`Attorney, Agent, or Firm Sughrue, Mion, Zinn, Macpeak
`& Seas, PLLC
`ABSTRACT
`57
`A time diversity transmission-reception System has, on
`transmit Side, at least one delay of a predetermined delay for
`providing at least a pair of delayed and undelayed data
`Symbol Sequences, at least a pair of modulators for modu
`lating the data Symbol Sequences into modulated interme
`diate frequency Signals, and at least a pair of Spread Spec
`trum Signal generators employing mutually different
`pseudo-random code Sequences for code division multiplex
`ing. On receive Side, the received signals are Subjected to
`code division demultiplexing, demodulation, and delaying
`to restore at least a pair of timed demodulated Signals. These
`Signals are then Subjected to adaptive matched filtering,
`whose outputs are combined and applied to adaptive equal
`ization to provide diversity reception Signal. The present
`System provides a plurality of diversity branches without
`employing a plurality of frequencies, planes of polarization
`of a carrier wave or receiving antennas. Also, principal part
`of the System of the present invention can be readily
`integrated into LSIs, contributing to the reduction of manu
`facturing cost.
`
`4 Claims, 7 Drawing Sheets
`
`Data Symbol
`Sequence
`an
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`309
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`ADAPTIVE
`ECUA.
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`ERICSSON v. UNILOC
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`U.S. Patent
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`Jan. 12, 1999
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`Sheet 1 of 7
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`5,859,870
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`U.S. Patent
`U.S. Patent
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`Jan. 12, 1999
`Jan. 12, 1999
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`Sheet 2 of 7
`Sheet 2 of 7
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`5,859,870
`5,859,870
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`U.S. Patent
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`Jan. 12, 1999
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`Sheet 3 of 7
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`U.S. Patent
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`Jan. 12, 1999
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`Sheet 4 of 7
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`5,859,870
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`FIG.4
`
`Received Signal
`Level
`
`401
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`Diversity
`Branch #1;
`402
`N
`S
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`Adaptive Filter
`input
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`Diversity
`Branch #2
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`ERICSSON v. UNILOC
`Ex. 1016 / Page 5 of 15
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`U.S. Patent
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`Jan. 12, 1999
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`Sheet 5 of 7
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`5,859,870
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`FIG. 5
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`Received Signal
`Level
`
`701
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`Diversity
`Branch #1
`502
`N
`S1
`
`Adaptive Filter
`Input
`
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`Diversity
`Branch #2
`503 A
`N
`S2
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`S2
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`
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`Adaptive Filter
`Output
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`Selector/Combiner
`Output
`
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`U.S. Patent
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`Jan. 12, 1999
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`U.S. Patent
`U.S. Patent
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`Jan. 12, 1999
`Jan. 12, 1999
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`Sheet 7 of 7
`Sheet 7 of 7
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`5,859,870
`5,859,870
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`ERICSSONv. UNILOC
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`1
`TIME DIVERSITY TRANSMISSION
`RECEPTION SYSTEM
`
`5,859,870
`
`2
`error rates and/or loss of frame Synchronization.
`Alternatively, Selector/combiner 111 may be composed of a
`diversity signal combiner designed to combine the two
`incoming demodulation output Signals after phase
`controlling them into timed State. The Signal combiner may
`be made of a maximal ratio combiner designed not only to
`control the phases of the incoming Signals into timed State
`but also to control their amplitudes to provide Squared and
`unsquared values thereof. The output of Selector/combiner
`111 is Supplied to adaptive equalizer 112 for eliminating
`waveform distortions caused by propagation through fading
`multipath channels, to provide reception data Signal A.
`The conventional system shown in FIGS. 1(a) and 1(b)
`may be called a frequency diversity System because two
`carrier waves of radio frequencies f and f are employed.
`However, this conventional System cannot achieve the fre
`quency diversity effect if the f-f correlation is high enough
`to make the Separation therebetween inadequate. On the
`other hand, delay t given to the Second branch transmit
`Signal provides the time diversity effect, if delay t is Set at
`a value longer than the fading period.
`In general, the f-f correlation cannot be reduced without
`expanding the frequency Separation therebetween, which is
`undesirable from the viewpoint of the more efficient use of
`frequencies. Therefore, the conventional System shown in
`FIGS. 1(a) and 1(b) depends more heavily on time diversity,
`with the frequency diversity used only for the purpose of
`providing time diversity branches. However, the use of two
`radio frequency carrier waves at frequencies f and f
`requires the use of two transmitters 103 and two receivers
`108, which are generally of large Scale and costly to manu
`facture. Furthermore, the problem of dimensions, Scale and
`manufacturing cost of the transmitter/receiver for the con
`ventional System becomes more Serious when the number of
`diversity branches is increased to more than two.
`Another prior art diversity transmission-reception System,
`which is outlined in Japanese Patent Application Kokai
`Publication No. Sho 63-286027 entitled “Transmission-path
`diversity-transmission system” and published Nov. 22,
`1988, has, as schematically shown in FIG. 2, modulator 201
`for data Signal, whose output is Supplied to a first transmit
`ting antenna 203 directly and also to a Second transmitting
`antenna 204 through a delay circuit 202 having delay t. On
`the receive side, the reception System has one receiving
`antenna 205, whose output is supplied to receiver 206,
`where the received RF signal is converted into an IF signal
`for detection by detector 207. The output of detector 207 is
`applied to waveform equalizer 208 for code decision by
`decision circuit 209. The delay t introduced by delay circuit
`202 is set at a value longer than one time slot assigned to
`each of the modulating data Symbols. The transmission
`carrier wave transmitted from the first and Second transmit
`ting antennas is propagated through mutually independent
`propagation paths, and Subsequently received by a single
`receiving antenna 205. Therefore, the propagated Signal
`consists of a plurality of multipath waves, whose undelayed
`and delayed components have respectively undergone Ray
`leigh fading mutually independently. Waveform equalizer
`208 is designed to select either the undelayed or delayed
`wave, while eliminating the unselected wave, thereby to
`realize two-branch Selective diversity reception. It is also
`described in the above-mentioned Kokai Publication that the
`undelayed and delayed waves are time-adjusted and com
`bined by maximal ratio combining.
`The prior art system of FIG. 2 is based on the concurrent
`use of time diversity and Space diversity reception Systems
`in that it utilizes the absence of Spatial correlation among
`
`BACKGROUND OF THE INVENTION
`(1) Field of the Invention
`The present invention relates to a diversity transmission
`reception System, and more particularly to a time diversity
`transmission-reception System utilizing code division mul
`tiplexing.
`(2) Description of the Related Art
`Diversity reception is generally necessary for radio com
`munication involving fading channels, and particularly for
`digital communication through fading multipath channels.
`Signal fading is classified into two, i.e., flat fading and
`frequency-Selective fading. Flat fading occurs as the ampli
`tude and phase variation in the propagated Signal, which has
`been received directly rather than through fading multipath
`channels. In contrast, frequency-Selective fading occurs as a
`result of propagation through fading multipath channels,
`each of which causes mutually independent amplitude and
`phase variations to the Signal propagated therethrough. In
`frequency-Selective fading, where the reception Signal is
`obtained through combining a plurality of Signals propa
`gated through fading multipath channels, the received Sig
`nals at Some frequencies may have mutually opposite phases
`resulting in Zero amplitude. This causes frequency-Selective
`fades or notches in the frequency spectrum of the reception
`signal. While the effect of flat fading is limited to the level
`of the reception signal with the waveform thereof
`unaffected, the frequency-Selective fading resulting from
`propagation through fading multipath channels causes the
`variation not only in reception Signal level but also in its
`waveform.
`To obviate the adverse effect of fading multipath
`channels, diversity techniques and adaptive equalization
`techniques have been conventionally utilized. Among a
`variety of possible combinations of these techniques, the
`present invention is directed to the use of time diversity
`combined with adaptive equalization.
`Referring to FIGS. 1(a) and 1(b) showing respectively in
`blocks the transmit and receive Sides of a conventional time
`diversity transmission-reception System, a data Signal (data
`Symbol Sequence) “a” to be transmitted is split into two and
`Supplied, one through delay means 101 (having delay t) and
`the other directly, to a pair of modulators 102 for modulation
`into modulated intermediate frequency (IF) signals, and then
`to a pair of transmitters 103 of carrier wave radio frequen
`cies (RF) f and f, whose outputs are combined at a
`combiner 104 and transmitted through a transmitting
`antenna 105. As a result, the two-branch transmit Signals are
`transmitted at radio frequencies f and f with a time spacing
`of t. On the receive side shown in FIG. 1(b), the RF signal
`received by receiving antenna 106 is applied to a branching
`filter 107, which splits the received RF signal into the f and
`f components. These frequency components are respec
`tively supplied to a pair of receivers 108 for amplification
`and conversion into intermediate frequency (IF) and then to
`a pair of demodulators 109 for synchronous detection. The
`output of demodulator 109 derived from the f component is
`then supplied through delay means 110 of delay t to opti
`mum Signal Selector/signal combiner 111, while the output
`of demodulator 109 derived from the f. component is
`supplied directly to selector/combiner 111. The selector/
`combiner 111 may consist of a Signal Selector for Selecting
`the demodulation output Signal of better quality out of the
`two incoming demodulation output Signals in response to bit
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`3
`propagation paths. While a Space diversity reception System
`ordinarily requires a plurality of receiving antennas, the
`above-cited prior art System has two transmitting antennas
`and a Single receiving antenna, reducing the equipment size
`on the receive Side and curtailing the manufacturing cost.
`The reduction of the number of receiving antenna to one is
`Significant for a microwave communication System, which
`requires large-aperture antennas and associated equipment
`in contrast to terrestrial mobile communication Systems
`requiring only Small-sized antennas.
`In the conventional diversity reception Systems outlined
`above, either frequency or space (propagation path) must be
`relied on as transmission media. Particularly, in the time
`diversity reception concurrently employing frequency diver
`sity for Separation and derivation of diversity branches, the
`expansion of required frequency bandwidth and equipment
`Scale is unavoidable, resulting in increased manufacturing
`cost. Similarly, the time diversity reception accompanied by
`Space (propagation path) diversity requires a plurality of
`antennas, increasing the manufacturing cost particularly
`when the antennas are of large aperture type. Furthermore,
`the increase in the number of diversity branches in the
`above-mentioned time diversity reception System accompa
`nied by either frequency or Space diversity makes the
`abovementioned problems more Serious.
`SUMMARY OF THE INVENTION
`An object of the invention, therefore, is to overcome the
`problems associated with the prior art and to provide a time
`diversity transmission-reception System adapted to realize
`time diversity transmission-reception utilizing spread
`Spectrum-based code division multiplexing.
`According to the present invention, there is provided a
`time diversity transmission-reception System, in which the
`time delay differences existing among the received signals
`Split through the code division demultiplexing and then
`demodulated are eliminated So that the received Signals may
`then be Supplied to adaptive matched filters, whose outputs
`are combined and then Supplied to adaptive equalizer.
`In the present invention, the transmission signal on the
`transmit Side may be split into more than two signals, which
`may be Subjected to mutually different delays to provide a
`plurality of time diversity transmission signals.
`BRIEF DESCRIPTION OF THE DRAWINGS
`The above and other objects, features and advantages of
`the present invention will be apparent from the following
`description of preferred embodiments of the invention taken
`with reference to the accompanying drawings, in which:
`FIGS. 1(a) to 1(b) show in blocks an example of the prior
`art time diversity transmission-reception System utilizing
`frequency diversity concurrently;
`FIG. 2 shows in blocks another example of the prior art
`time diversity transmission-reception System utilizing Space
`diversity concurrently;
`FIGS. 3(a) and 3(b) show in blocks a time diversity
`transmission-reception System embodying one feature of the
`present invention, in which the number of diversity branches
`is two;
`FIG. 4 illustrates the operation of the system of FIGS.3(a)
`and 3(b), when no multipath propagation is involved;
`FIG. 5 illustrates the operation of the system of FIGS.3(a)
`and 3(b) when multipath propagation is involved;
`FIGS. 6(a) and 6(b) show in blocks a time diversity
`transmission-reception System embodying another feature of
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`the present invention, in which the number of diversity
`branches is four, and
`FIG. 7 illustrates the operation of the system shown in
`FIGS. 6(a) and 6(b).
`PREFERRED EMBODIMENTS OF THE
`INVENTION
`Now, preferred embodiments of the invention will be
`described with reference to the drawings, in which like
`Structural elements as in the prior art described above are
`denoted by like reference numerals with their description
`unrepeated.
`Referring to FIG.3(a) showing in blocks the transmit side
`of a first embodiment of the present invention, the pair of
`outputs of modulators 102, which are supplied with a data
`Signal (Symbol Sequence) a, one directly and the other
`through delay means 101 of delay t (greater than signal
`fading period), are Supplied to a pair of spread spectrum
`Signal generators 303 employing mutually different pseudo
`noise code Sequences for code division multiplexing, whose
`outputs are combined at combiner 104 for transmission as an
`RF signal through transmitter 305 and transmitting antenna
`306. As shown, two code division multiplexed channels are
`assigned respectively to two diversity branches, which are
`defined by delay t and formed by an RF signal of a single
`frequency.
`On the receive side of the first embodiment of the present
`invention shown in FIG. 3(b), the RF signal propagated
`through the fading multipath channels is received at receiv
`ing antenna 307 and applied through a receiver 308, which
`is for low-noise amplification and conversion into interme
`diate frequency (IF), to a pair of spectrum-unspread signal
`restoring (code division demultiplexing) circuits 309, whose
`outputs are demodulated at demodulators 109, with an IF
`local oscillation Signal Supplied thereto from a local oscil
`lator 315 for quasi-synchronous detection. A pair of
`demodulated outputs from demodulators 109 are supplied,
`one through delay means 110 of delay t and the other
`directly, to a pair of adaptive matched filters 312, whose
`outputs are Summed at combiner 313 and then applied to
`adaptive equalizer 112 to provide reception data Signal An.
`Each of the adaptive matched filters 312 has: first and second
`delay elements 312a1 and 312a2 of delay T/2 (where T
`denotes symbol period of the output from the demodulator
`109) connected in series with respect to the output of
`demodulators 109; first, second and third complex multipli
`ers 312b1, 312b2 and 312b3 receiving inputs from taps P, Q
`and R defined by the serially connected first and second
`delay elements 312a1 and 312a2; a Summer 312c, and first,
`second and third complex correlation circuits 312d1, 312d2
`and 312.d3, which are adapted to take correlation between
`the output of the adaptive equalizer 112 and the above
`mentioned outputs from taps P, Q and R, respectively, and to
`Supply their correlation outputs to complex multipliers
`312b1,312b2 and 312b3, respectively. The delay tgiven by
`delay means 110 to the first diversity branch brings both the
`first and Second diversity branches into timed State due to the
`Same amount of delay t given to the Second diversity branch
`on the transmit side shown in FIG. 3(a).
`Each of the adaptive matched filters 312 supplied with the
`demodulation output is adapted to predict time-variant
`impulse response characteristics of transmission paths and to
`convolute into received signal the time-reversed complex
`conjugate values of the prediction results. It generally
`follows, therefore, that an adaptive matched filter has the
`structure of a transversal filter. In the embodiment shown in
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`FIGS. 3(a) and 3(b), the adaptive matched filter 312 has
`three taps P, Q and R having a tap interval of T/2 (where T
`is the modulating symbol period). According to communi
`cation theory, matched filtering of a received signal maxi
`mizes SNR (signal to noise power ratio). This is due to the
`fact that the Signal power, which has been time-dispersed by
`multipath channel propagation, is spread over the taps of the
`adaptive matched filter to allow the time-dispersed signal
`power to be time-concentrated or matched by transversal
`filtering for maximal ratio combining of Signals on those
`tapS.
`To describe the function of adaptive matched filter 312,
`the transmission path model is assumed to result in a
`principal Signal wave accompanied by two T/2-delayed
`waves propagated through multipath channels. ASSuming
`that the principal signal wave component h(O)a, is distrib
`uted to middle tap Q, first tap P is Supplied with a signal
`component h(T/2) corresponding to the T/2-delayed one. It
`is noted here that h() denotes complex amplitude value h(t)
`of an impulse response. It is also noted that the Signal
`components at taps P, Q and R are applied to the three
`complex correlators 312d1, 312d2 and 312.d3, which take
`correlation of the respective signal components with the
`output signal A of adaptive equalizer 112 to provide
`weighted coefficients W1, W2 and W, respectively.
`These coefficients W. (i=1,2,3; J=1,2,3; where istands for
`the i-th diversity branch and j for j-th tap of the adaptive
`matched filter), which are multiplied respectively at multi
`pliers 312b1, 312b2 and 312b3 with the above-mentioned
`Signal components Supplied from taps P, Q and R, can be
`calculated as follows:
`
`W=E h*(T2)axA.
`
`(1)
`
`(2)
`W=E h(0)axA,
`where E means integrating operation for time-averaging
`complex called expectation value operation, and * means
`conjugate complex Values.
`In this correlation operation, when the bit error rate is in
`the order of 10°, the following approximation holds:
`A.
`(3)
`On the other hand, Since the data Signal a, on the transmit
`side can be approximated to M series of pseudo-noise (PN)
`Signals, the following autocorrelation coefficients hold:
`
`passif
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`35
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`40
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`As a result, equations (1) and (2) are transformed respec
`tively to:
`
`50
`
`It follows, therefore, that the Signal components derived
`respectively from taps P and Q and multiplied respectively
`at multipliers 312b1 and 312b2 by coefficients W. and W.
`and Summed at Summing circuit 312c to constitute the
`matched filter output is given by:
`
`The first term of the above expression gives, as coefficients
`for modulating data Symbols a, , Squared components of
`response to delayed wave, while the Second term gives
`Squared components of response to principal wave, with
`both the first and Second terms giving real numbers. It is seen
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`from this expression that time-dispersed data Symbols a
`resulting from propagation through multipath medium are
`time-matched for maximal ratio combining. This is equiva
`lent to the So-called time diversity signal combining, whose
`gain is called implicit diversity gain. While the quasi
`Synchronous detection performed at the above-mentioned
`demodulators 109 is not a full synchronous detection, the
`difference between the quasi- and full-synchronous detec
`tion appears as the rotation of phase angle of impulse
`response. This phase angle rotation is canceled, as shown in
`expression (7), by the product of the impulse response and
`its conjugate complex values. In this way, the adaptive
`matched filter exhibits the function of carrier synchroniza
`tion.
`The Similar matched filtering is performed at the Second
`adaptive matched filter 312, so that the outputs from the two
`adaptive matched filters 312 may be combined by combiner
`313 to maximize their SNR.
`The Spread spectrum Signals employed in the present
`invention for the code division multiplexing do not easily
`Suffer from waveform distortion, which is otherwise caused
`by fading multipath channels. This is particularly true when
`multipath-propagated Signal waves have no correlation with
`principal Signal wave, wherein multipath-propagated Signal
`waves are Suppressed by Spectrum spreading. In other
`words, in those cases where the effect of fading multipath
`channels is light enough to make the delay of the principal
`wave Small, correlation between the principal and the
`delayed signal waves becomes greater then negligible, mak
`ing it impossible to Suppress signal waves (propagated
`through multipath channels) by the use of spread spectrum
`Signals. In Such a case, it is desirable to positively utilize as
`valid Signals those signal waves propagated through multi
`path channels. The adaptive matched filters 312 are
`employed in the present invention to utilize those multipath
`propagated Signals.
`The output of the combiner 313 is applied to adaptive
`equalizer 112 to eliminate code-to-code interference, which
`has been caused by multipath medium but has already been
`reduced considerably by the above-mentioned matched fil
`tering. Adaptive equalizer 112 may be of the adaptive
`filter-based type or of the MLSE (Maximum Likelihood
`Sequence Estimation)-based type. While the former can be
`linear filter-based one or nonlinear filter-based one, a non
`linear filter-based decision feedback-type equalizer (DFE)
`provides more powerful equalization when combined with a
`matched filter.
`While the relationship between adaptive matched filter
`and multipath channels has been described above, the effect
`of diversity reception achievable by the present invention in
`the absence of propagation through multipath channels will
`now be described.
`Referring to FIG. 4 illustrating the operation of the system
`of FIGS. 3(a) and 3(b), curve 401 shows time variation of
`the received signal level; reference numeral 402 shows
`Signal vector S of the Signal received at the input to the
`adaptive matched filter 312 of the first diversity branch; 403,
`Signal vector S of the Signal received at the input to the
`adaptive matched filter 312 of the second diversity branch;
`404, Signal vector of the Signal at the output of the adaptive
`matched filter 312 of the first diversity branch; 405, signal
`vector of the Signal at the output of the adaptive matched
`filter 312 of the second diversity branch; and 406, signal
`vector at the output of the combiner 313.
`In the absence of multipath-propagated waves, the adap
`tive matched filter 312 performs the ordinary maximal ratio
`combining for the principal waves exhibiting Signal vectors
`S and S.
`
`ERICSSON v. UNILOC
`Ex. 1016 / Page 11 of 15
`
`

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`5,859,870
`
`7
`It is assumed here, as shown in FIG. 4 by curve 401, that
`the received signal level Suffers at time t=t (first diversity
`branch) a deep fade caused by Signal fading, while it exhibits
`no fading effect at time t=to-t (second diversity branch). In
`this situation, the Signal transmitted through the first branch
`at t=to is given by the product of data Signal a, and complex
`transfer coefficients h(t). On the other hand, data Signal a
`is transmitted again after the lapse of t through the Second
`diversity branch, whose received signal is given by the
`product of a, and h (to-t).
`Since the time matching is achieved on the receive side by
`delaying the received signal on the first diversity branch by
`t, the received signals S and S. at the input to the middle
`tap Q of adaptive matched filter 312 are given by:
`
`(9)
`With correlation values W. and W. provided by complex
`correlators 312d2 of each of the first and second diversity
`branches, the output y from combiner 113 is given by:
`(10)
`y=W S+W. S.
`Substitution of the terms in equation (10) with equations (8)
`and (9) gives:
`y={Wixh(t)+W,xh(to-t)}xa,
`(11)
`Tap coefficients W. and W are given, similarly to equa
`tions (5) and (6) above, by:
`
`(12)
`
`W12 = E{h(to)an.* X an
`= E h(to)x Ea, a
`= h(to)*
`W22 = E{h(to + t)an.* x an
`= E h(to + t)x Ea, x a.
`= h(to + t)*
`Therefore, the output of combiner 313 expressed by equa
`tion (11) above is given by:
`
`(13)
`
`8
`variation in Signal vectorS S and S without any correlation
`therebetween. Therefore, the maximal ratio combining of
`Signal vectorS S and S. realizes time diversity reception.
`The diversity effect achieved by the present invention in
`the presence of multipath propagation will now be described
`with reference to FIG. 5. As in the case of FIG. 4, curve 501
`shows time variation of received signal level. Similarly,
`reference numerals 502 to 506 show vector diagrams cor
`responding to those denoted by reference numerals 402 to
`406, respectively. The illustration in FIG. 5 differs from that
`of FIG. 4 only in that the signal vectors shown therein are
`affected by multipath propagation, resulting in principal
`wave and delayed wave. In vector diagram 502 of FIG. 5,
`Signal vector S of principal wave is shown accompanied by
`Signal vector S' of delayed wave. These principal and
`delayed wave components resulting from multipath propa
`gation are Subjected to maximal ratio combining by adaptive
`matched filters 312. More specifically, as shown in vector
`diagram 504, signal vectors S and S on the first diversity
`branch are multiplied by tap coefficients W. and W,
`respectively, to bring the phases of the Signal vectorSS and
`S' into in-phase State on the real number axis and to bring
`their amplitudes into unsquared-Squared value relationship.
`Similarly, Signal vectorS S and S" on the Second diversity
`branch shown in vector diagram 503 are multiplied by tap
`coefficients W. and W, respectively, to provide resultant
`signal vector as shown in vector diagram 505. To take a
`close look at the first diversity branch, the received signal is
`never interrupted even if the level of the signal vector S of
`the received principal wave is lowered, So far as the level of
`the received delayed wave is not lowered. It follows,
`therefore, that the effect of dual diversity reception can be
`achieved even if only one diversity branch is utilized. Thus,
`each of the diversity branches provides the implicit diversity
`gain to enhance received signal level for Signal combining of
`improved quality. This leads to Signal combining of Virtual
`quadruple diversity reception. Comparison of the Vector
`diagram 505 of FIG. 5 with the corresponding diagram 405
`of FIG. 4 shows that the presence of multipath propagation
`provides enhanced diversity reception effect.
`While the embodiment described above with reference to
`FIGS. 3(a), 3(b), 4 and 5, which has two diversity branches,
`is effective enough to achieve practically adequate diversity
`reception effect in most cases, this arrangement is not
`capable of preventing reception Signal interruption if Signal
`fade occurs at both t=to and t=to-t. To overcome the problem
`of Such reception signal interruption and thereby to enhance
`the quality of communication channels, the number of
`diversity branches must be increased. FIGS. 6(a) and 6(b)
`show a Second embodiment of the present invention
`designed to meet such demand. Referring to FIG. 6(a)
`showing the transmit Side of a time diversity transmission
`reception System having four diversity branches, reference
`numerals 601, 602 and 603 denote delay means of delay t,
`2t, and 3t, respectively; 604, four modulators; 605, four
`Spread Spectrum Signal generators, 606, Signal combiner;
`607, RF transmitter; and 608, transmitting antenna. On the
`receive side shown in FIG. 6(b), the reference numeral 609
`denotes receiving antenna; 610, RF receiver; 611, four
`Spectrum-unspread signal generators (code division
`demultiplexers); 612, four demodulators; 613, 614 and 615,
`delay means having delay 3t, 2t, and t, respectively, 616, a
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`It should be noted here that the transfer coefficients for
`multiplication with data Symbols a, has a dimension of
`power and is in real number. To describe this with reference
`to the vector diagrams of FIG. 4, the multiplication of Signal
`vector S in vector diagram 402 with W lead