United States Patent (19)
`Tsujimoto
`
`USOO6075808A
`Patent Number:
`11
`(45) Date of Patent:
`
`6,075,808
`Jun. 13, 2000
`
`54 SPREAD SPECTRUM TIME DIVERSITY
`COMMUNICATIONS SYSTEM
`
`75 Inventor: Ichiro Tsujimoto, Tokyo, Japan
`73 Assignee: NEC Corporation, Tokyo, Japan
`
`21 Appl. No.: 09/018,893
`22 Filed:
`Feb. 5, 1998
`Related U.S. Application Data
`62 Division of application No. 08/620,246, Mar. 22, 1996, Pat.
`No. 5,757,853.
`Foreign Application Priority Data
`30
`Mar. 22, 1995
`JP
`Japan .................................... 7-062821
`(51) Int. Cl. ............................................... H04B 1/707
`52 U.S. Cl. ............................................. 375/143; 37.5/152
`58 Field of Search ..................................... 375/140, 141,
`375/142,143, 147, 150, 152; 370/320,
`335, 342, 441
`
`56)
`
`References Cited
`
`U.S. PATENT DOCUMENTS
`5,461,610 10/1995 Weerackody ............................ 375/146
`5,471,497 11/1995 Zehavi .........
`... 375/142
`5,677,929 10/1997 Asano et al. ............................ 375/141
`
`Primary Examiner Don N. Vo
`Attorney, Agent, or Firm Sughrue, Mion, Zinn, Macpeak
`& Seas, PLLC
`57
`ABSTRACT
`In a time diversity radio communications System,
`quadrature-modulated Spread spectrum signals of different
`mutual time delay are combined into a code division mul
`tiplex Signal and up-converted to a radio-frequency Signal
`and transmitted. At a receive Site, the Signal is received by
`an antenna and down-converted to recover the code division
`multiplex Signal. Quadrature-modulated component signals
`contained in the recovered code division multiplex Signal are
`converted to baseband Signals which are then time-aligned
`with each other. The time-aligned baseband Signals are
`multiplied respectively by complex weighting factors and
`are combined together to produce an input Signal for an
`adaptive equalizer. This input signal is also applied to an
`AGC amplifier where it is amplified and applied to corre
`lation detectors where correlations between the amplified
`Signal and the baseband Signals are detected and the com
`pleX weighting factors are derived respectively from the
`correlations. In a modified embodiment, the quadrature
`modulated component Signals are fed into an adaptive
`RAKE matched filter where the output of the AGC amplifier
`is used for detecting the correlations.
`
`7 Claims, 5 Drawing Sheets
`
`
`
`DESPREADING
`CODE'A'
`
`DESPREADING
`CODE "B"
`
`31 OB
`
`
`
`EQUALIZER
`
`ERICSSON v. UNILOC
`Ex. 1014 / Page 1 of 11
`
`

`

`U.S. Patent
`
`Jun. 13, 2000
`
`Sheet 1 of 5
`
`6,075,808
`
`FG. A
`
`SPREADING
`CODE "A"
`103A
`
`O4
`
`102A
`
`MODULATOR
`
`SCRAMBER
`
`
`
`
`
`
`
`MODULATOR
`
`SCRAMBLER
`
`SPREADING
`CODE "B"
`
`FG, 1B
`
`DESPREADING
`CODE "A"
`
`RA
`
`10A
`
`ADAPTIVE
`EQUALIZER
`
`DESPREADING
`CODE'8"
`
`ERICSSON v. UNILOC
`Ex. 1014 / Page 2 of 11
`
`

`

`U.S. Patent
`
`Jun. 13, 2000
`
`Sheet 2 of 5
`
`6,075,808
`
`FG, 2
`
`20
`
`OUTPUT OF DIVERSITY
`BRANCHRA
`
`OUTPUT OF DIVERSITY
`BRANCH RB
`
`
`
`204
`
`205
`
`OUTPUTOF
`MULTIPLER 2A
`
`206
`
`
`
`OUTPUT OF
`COMBINER 114
`
`ERICSSON v. UNILOC
`Ex. 1014 / Page 3 of 11
`
`

`

`U.S. Patent
`
`Jun. 13, 2000
`
`Sheet 3 of 5
`
`6,075,808
`
`FG, 3A
`102-0
`
`104
`
`103-0
`
`
`
`
`
`
`
`
`
`
`
`
`
`109.
`
`DESCRAMBLER
`
`14
`
`
`
`ADAPTIVE
`EQUALIZER
`
`
`
`ERICSSON v. UNILOC
`Ex. 1014 / Page 4 of 11
`
`

`

`U.S. Patent
`U.S. Patent
`
`Jun. 13, 2000
`Jun. 13, 2000
`
`Sheet 4 of 5
`Sheet 4 of 5
`
`6,075,808
`6,075,808
`
`
`
`ONIGVIEdSIC
`
`
`
`s¥,I00)
`
`IALAVOY
`
`¥3Z11¥03
`
`gOle
`
`ONIGVIUdSIG
`
`«9,140)
`
`bls
`
`ERICSSONv. UNILOC
`Ex. 1014/ Page 5 of 11
`
`ERICSSON v. UNILOC
`Ex. 1014 / Page 5 of 11
`
`
`
`

`

`U.S. Patent
`
`Jun. 13, 2000
`
`Sheet 5 of 5
`
`6,075,808
`
`FG, 5A
`
`O
`
`
`
`
`
`
`
`to
`
`to +
`FG, 5B
`
`to +2t
`
`time
`
`FG, 5C
`
`time
`
`time
`
`PRECURSOR
`
`MAN RESPONSE
`
`POSTCURSOR
`
`ERICSSON v. UNILOC
`Ex. 1014 / Page 6 of 11
`
`

`

`6,075,808
`
`1
`SPREAD SPECTRUM TIME DIVERSITY
`COMMUNICATIONS SYSTEM
`
`This is a divisional of application Ser. No. 08/620.246
`filed Mar. 22, 1996 which is now U.S. Pat. No. 5,757,853.
`
`BACKGROUND OF THE INVENTION
`
`2
`predetermined inter-signal time delay between the
`quadrature-modulated spread spectrum information-bearing
`Signals. The code division multiplex Signal is up-converted
`to a radio-frequency signal and transmitted. At a receive site,
`the transmitted Signal is received by an antenna and down
`converted to recover the code division multiplex Signal. A
`demultiplexer circuit is provided for converting a plurality
`of quadrature-modulated component Signals contained in the
`recovered multiplex Signal to a plurality of information
`bearing Signals using quadrature carriers of the Single fre
`quency and a plurality of despreading codes respectively
`identical to the spreading codes So that the plurality of
`information-bearing Signals are time coincident with each
`other. The information-bearing Signals from the multiplexer
`circuit are multiplied respectively by a plurality of complex
`weighting factors and combined together to produce a
`combined information-bearing Signal. An automatic gain
`controlled (AGC) amplifier is provided for amplifying the
`combined information-bearing Signal. The complex weight
`ing factors are derived from correlations between the output
`of the AGC amplifier and the information-bearing Signals
`from the demultiplexer circuit.
`According to a Second aspect of this invention, the receive
`Site includes a demultiplexercircuit for converting a plural
`ity of quadrature-modulated component Signals contained in
`the recovered multiplex Signal to a plurality of information
`bearing Signals using quadrature carriers of the Single fre
`quency and a plurality of despreading codes respectively
`identical to the Spreading codes, and a first combiner for
`combining the plurality of information-bearing Signals into
`a combined information-bearing Signal. An adaptive RAKE
`matched filter is provided, which comprises a delay line
`having a plurality of Successive taps for receiving the
`combined information-bearing Signal, the delay time
`between the Successive taps corresponding to the predeter
`mined inter-signal time delay, a plurality of complex mul
`tipliers for respectively weighting tap Signals from the taps
`of the delay line by complex weighting factors, a Second
`combiner for combining the weighted tap Signals to produce
`a combined weighted Signal for coupling to an adaptive
`equalizer, and a plurality of correlation detectors for respec
`tively deriving the complex weighting factors from corre
`lations between a decision output from the adaptive equal
`izer and a plurality of delayed versions of tap Signals at the
`taps of the delay line, one of the delayed versions being time
`coincident with the decision output.
`BRIEF DESCRIPTION OF THE DRAWINGS
`The present invention will be described in further detail
`with reference to the accompanying drawings, in which:
`FIGS. 1A and 1B are block diagrams of a transmitting
`apparatus and a receiving apparatus of a diversity radio
`communications System, respectively, according to a first
`embodiment of the present invention;
`FIG. 2 shows signal vectors appearing in the receiving
`apparatus of FIG. 1B when a deep fade occurs in a received
`Signal;
`FIGS. 3A and 3B are block diagrams of a generalized
`form of a transmit and a receive Site of the System;
`FIG. 4 is a block diagram of a modified receiving appa
`ratus of the present invention; and
`FIGS. 5A-5C show impulse response characteristics of
`the adaptive RAKE matched filter used in the modified
`receiving apparatus of FIG. 4.
`DETAILED DESCRIPTION
`Referring now to FIG. 1A, there is shown a transmitting
`apparatus of a time-diversity radio communications System
`
`1. Field of the Invention
`The present invention relates to a diversity radio commu
`nications Systems for multipath fading channels, using a
`Spread spectrum multiplexing technique.
`2. Description of the Related Art
`In radio communications Systems where frequency
`Selective, multipath fading occurs, the waveform of trans
`mitted symbols is affected by the time dispersal effect of the
`fading channel. Diversity reception and adaptive equaliza
`tion techniques are normally employed to combat this prob
`lem. One of the known techniques is the frequency-and-time
`diversity reception wherein an intermediate-frequency
`modulated information-bearing Signal and a delayed replica
`of this signal are up-converted to two radio signals of
`different frequencies and transmitted from a single antenna.
`At a receive site, the transmitted Signals are demultiplexed
`onto Separate diversity branches where they are low-noise
`amplified and down-converted to IF signals from which
`baseband Signals are recovered. One of the baseband Signals
`is delayed So that they are time coincident with each other
`and then compared with each other. One of the Signals that
`is leSS distorted is Selected as an input to an adaptive
`equalizer where multipath-fading related interSymbol inter
`ference is canceled. Alternatively, the time-aligned signals
`are modified both in phase and amplitude So that they are
`maximal-ratio combined to produce an input to the adaptive
`equalizer. However, due to the use of two radio frequencies,
`the cost of high-power transmitters and low-noise receivers
`is Substantial if the number of diversity branches increases.
`Another prior art approach is concerned with the Space
`time diversity reception technique. AS described in Japanese
`Provisional Patent Publication Sho-63-286027, a radio
`frequency modulated Signal and a delayed replica of this
`Signal are transmitted from respective antennas which are
`Spaced So that the transmitted Signals propagate through
`Separate multipath Rayleigh fading channels and received by
`a single antenna at a receive site, where the received signals
`are down-converted to baseband Signals with a differential
`delay between them. These signals are diversity combined in
`a RAKE equalizer. While the cost of the receive site is
`reduced by the use of a Single antenna, the use of two
`antennas at the transmit Site would add an extra cost if the
`System is to be used in a microwave communications where
`large aperture antennas are required.
`
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`SUMMARY OF THE INVENTION
`It is therefore an object of the present invention to use
`Spread spectrum multiplexing and time diversity reception
`techniques for radio communication to minimize cost and
`equipment size.
`According to a first aspect of the present invention, there
`is provided a radio communications System which comprises
`a multiplexercircuit for producing a plurality of quadrature
`modulated Spread Spectrum information-bearing Signals
`using quadrature carriers of Single frequency and a plurality
`of Spreading codes, and combining the quadrature
`modulated Spread spectrum information-bearing Signals to
`produce a code division multiplex Signal, there being a
`
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`ERICSSON v. UNILOC
`Ex. 1014 / Page 7 of 11
`
`

`

`3
`according to a first embodiment of the present invention.
`The transmiting apparatus of the System essentially com
`prises a multiplexercircuit connected to a signal Source 100
`which generates a Series of transmit information-bearing
`Symbols a, The multiplexer includes first and Second diver
`sity branches TA and TB. The first branch TA includes a
`quadrature modulator 102A such as PSK (phase shift
`keying) or QAM (quadrature amplitude modulation) modu
`lator connected to the signal source 100. In the modulator
`102A, the input Signal is mapped onto a set of complex
`Signals (sin and cosine values) which are respectively modu
`lated onto quadrature carriers at intermediate frequency by
`means of an in-phase mixer and a quadrature mixer and
`combined together by a Summer to produce a quadrature
`modulated (PSK or QAM) information-bearing symbol data.
`The output of modulator 102A is fed into a scrambler 103A
`where it is with a spreading code A of pseudorandom
`Sequence to produce a first code division component Signal.
`The second branch TB is similar to branch TA with the
`exception that it includes a delay element 101 for providing
`a predetermined amount of delay t to the symbol data. The
`delay time t is greater than the intervals at which the
`information-bearing Symbols occur. The delayed Symbol is
`modulated onto the intermediate frequency quadrature car
`riers by a modulator 102B, and Spread with a spreading code
`B in a scrambler 103B to produce a second code division
`component Signal.
`It is to be noted that, in each diversity branch, the
`positions of the modulator 102 and scrambler 103 can be
`interchanged Substantially without any effect on System
`performance. The outputs of scramblers 103A and 103B are
`combined together in a combiner 104 to produce a code
`division multiplex Signal. This signal is up-converted to a
`radio frequency Signal and high-power amplified in a trans
`mitter 105 and transmitted from an antenna 106.
`At a receive site of the diversity communications System,
`shown in FIG. 1B, the transmitted signal is received by an
`antenna 107, and then low-noise amplified and down
`converted by a receiver 108 using the Same quadrature
`radio-frequency carriers as used at the transmit Site. The
`output of receiver 108 is applied to first and second diversity
`branches RA and RB. The diversity branch RA includes a
`descrambler 109 for despreading the output of receiver 108
`with a first despreading code Aidentical to the first spreading
`code A to produce a replica of the output Signal of the
`modulator 102A. This signal is applied to a quadrature (PSK
`or QAM) demodulator 110A where it is demodulated with a
`carrier of the same intermediate frequency as that used in the
`transmit site. The output of demodulator 110A is delayed by
`a delay element 111 by the same amount t as that introduced
`to the transmitter's diversity branch TB at the transmit site,
`producing an output Signal of the first diversity branch RA.
`The receiver's diversity branch RB also includes a descram
`bler109B for despreading the output of receiver 108 with a
`Second despreading code Bidentical to the Second spreading
`code B to produce a replica of the output Signal of the
`modulator 102B. This signal is applied to a quadrature
`demodulator 110 B where it is quadrature demodulated with
`carriers of the same intermediate frequency as that used in
`the transmit Site, producing an output Signal of the receiver's
`diversity branch RB.
`The outputs of receiver's diversity branches RA and RB
`are Supplied respectively as signals S and S to complex
`multipliers 112A and 112B where they are multiplied with
`first and Second tap-weight signals W and W of complex
`value Supplied respectively from correlation detectors 113A
`and 113B. The outputs of complex multipliers 112A and
`
`4
`112B are maximal-ratio combined in a diversity combiner
`114 to produce a replica of the original Symbol data though
`it may have been corrupted during transmission. The output
`of diversity combiner 114 is fed to an adaptive equalizer 116
`Such as decision feedback equalizer to cancel multipath
`fading related interSymbol interference (ISI) and a decision
`output symbol A, is produced. The diversity combiner 114
`output is further used as a feedback signal by an automatic
`gain controlled amplifier 115 where the amplitude
`(envelope) of the Signal is detected.
`The detected amplitude is applied as a reference Signal R
`to the correlation detectors 113A and 113B which derive
`complex tap-weight coefficients W. and W from correla
`tions between the reference Signal and the output Signals S
`and S of the diversity branches RA and RB.
`ASSume that the receive level of the incoming Signal
`varies with time as indicated by a curve 201 in FIG. 2 where
`the signal drops significantly at time to due to a deep fade
`and the diversity branch output Signals S and S are
`respectively indicated by vectors 202 and 203 having dif
`ferent phase angles. The signal received at time to through
`the diversity branch RA is a product of the transmitted
`symbol “a” by a complex transfer coefficient h(t) of the
`communication channel. The same Symbol “a” is received
`again at time to--T through the diversity branch RB as a
`product of a, by a complex transfer coefficient h(tot) of the
`communication channel. Due to the delay element 111, the
`input Signals S and S of the complex multipliers 112A and
`112B are time coincident with each other and consequently
`given by:
`
`S=h(tot)'a,
`
`(2)
`
`The output signal “y” of the diversity combiner 114 is given
`in the form:
`
`y={Wh(t)+W, h(tot) 'a,
`
`(3)
`
`Since the reference signal R from the AGC amplifier 115 can
`be considered as having a normalized amplitude, it can be
`represented as:
`
`The tap-weight value W detected by correlator 113A is the
`result of multiplying complex conjugates of the input Signal
`S by the reference Signal R and time-averaging the product
`using an RC (resistance-capacitance) network lowpass filter
`in the case of analog circuits or Successive updating pro
`ceSSes in the case of digital circuits. Likewise, the tap
`weight value W detected by correlator 113B is the result of
`multiplying complex conjugates of the input signal S by
`reference Signal R and time-averaging the product.
`Therefore, tap-weight values W. and W are represented by:
`
`W ES a
`
`a
`= Eh (to)a,
`= Eh (to). Ea, a
`
`(5)
`
`6,075,808
`
`5
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`ERICSSON v. UNILOC
`Ex. 1014 / Page 8 of 11
`
`

`

`6,075,808
`
`W2 = EIS. a
`
`= Eh (to + i)a,
`a
`= Eh (to + i). Ea, a
`
`(6)
`
`where E represents the time-averaging process, and where
`the autocorrelation Ea, all of data symbols can be repre
`sented by Kronecker delta 8-i(if i=j) or 0(if izi).
`The time interval in which the time-averaging process is
`performed is sufficiently longer than the symbol interval but
`sufficiently shorter than the interval at which fading is likely
`to occur. Thus, the fading variations are not averaged out and
`consequently they do not contribute to the correlation.
`Equations (5) and (6) can therefore be rewritten as:
`W=h(t)
`
`(7)
`
`AS a result, the output Signals of the complex multipliers
`112A and 112B are given by the following relations:
`
`(10)
`W.S.-h (to-t) h(to-t)'a,
`Equations (9) and (10) indicate that the transfer coefficients
`are transformed into real numbers in the dimension of
`power.
`In FIG. 2, the outputs of the complex multipliers 112A and
`112B are aligned in phase to the real axis and amplified by
`a Squared value of transfer coefficient as indicated by vectors
`204 and 205. Therefore, the output signals of the complex
`multipliers 112A and 112B are maximal ratio combined by
`the diversity combiner 114 to produce an output signal “y”
`which is indicated by a vector 206 in the form:
`
`Since the terms h(t) and h(to+t) vary with independent
`Rayleigh fading, the variations in the diversity branch out
`puts S and S are uncorrelated with each other. By the
`maximal ratio combining process of the branch output
`Signals, the communications System of this invention oper
`ates in a time diversity mode.
`If a deep fade should occur at time tot as well as at time
`to, Signal interruption inevitably occurs. To ensure against
`Such interruptions, it is preferable to provide as many
`diversity branches as possible.
`A generalized form of the time diversity radio communi
`cations system is shown in FIGS. 3A and 3B. In FIG. 3A, the
`transmit site includes (n+1) diversity branches with a delay
`time difference t between adjacent branches. Delay ele
`ments 101-1 to 101-n of delay times t, 2, .
`.
`. , nt are
`provided in the Second to (n+1)th diversity branches, respec
`tively. Quadrature-modulators 102-0-102n are provided
`respectively in the first to the (n+1)th diversity branches and
`scramblers 103-0-103-n are connected to the corresponding
`quadrature modulators. Using (n+1) different spreading
`codes, the scramblers 103-0-103n supply their spread spec
`trum component Signals to the combiner 104 to produce a
`code division multiplex signal. In FIG. 3B, the receive site
`includes (n+1) diversity branches corresponding in number
`to the diversity branches of the transmit site. Similar to the
`transmit site, a delay time difference t is established between
`adjacent branches by delay elements 111-1, 111-2, ... 111-n
`having delay times T, 2t, . . . , m, respectively. The delay
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`6
`elements 111-1 to 111-n are respectively connected to the
`output of quadrature demodulators 110-1 to 101-n of the
`second to (n+1)th diversity branches. Descramblers 109
`0-109-n, connected to the receiver 108, use (n+1) despread
`ing codes identical to the (n+1) spreading codes to despread
`the output of the receiver and Supply their outputs to the
`corresponding demodulators. Correlators 113-0-113-in are
`connected to the output of AGC amplifier 115 to detect
`correlations between the reference Signal R and the output
`Signals So-S, from the corresponding diversity branches.
`With the use of an increasing number of diversity branches
`of different delay times, there is a less likelihood of encoun
`tering deep fades at Successive instants of time.
`According to a Second embodiment of the present
`invention, the receive site of FIG. 1B is modified as shown
`in FIG. 4. This embodiment is characterized by the use of an
`adaptive RAKE matched filter (where the term “RAKE”
`implies that it collects the Signal energy from all the received
`Signal paths that fall within the Span of its delay line and
`carry the same information and So its action is analogous to
`an ordinary garden rake). Similar to FIG. 1B, the receive site
`includes a receiver 308 to receive signals detected by
`antenna 307. Diversity branches RA and RB of the same
`configuration are connected to the output of receiver 308 for
`despreading the recovered code division multiplex Signal in
`descramblers 309A, 309B, using the despreading codes “A”
`and “B” respectively to produce replicas of the non-delayed
`and t-delayed modulated Signals which appear at the outputs
`of quadrature modulators 102A, 102B (FIG. 1A). These
`Signals are quadrature-demodulated in demodulators 310A,
`310B to produce replicas of the non-delayed and t-delayed
`baseband Signals which appear at the inputs of quadrature
`modulators 102A, 102B. Unlike the first embodiment, tim
`ing alignment is not provided between the diversity
`branches. The outputs of the diversity branches are summed
`together in a combiner 311.
`The matched filter 312 includes a first tapped delay line
`formed by a delay element 313 for receiving the output of
`combiner 311 to produce tap Signals at the input and output
`of the delay element. These taps of the first delay line are
`connected to tap-weight complex multipliers 314 and 315,
`respectively, for weighting the tap Signals by tap-weight
`coefficients W. and W Supplied respectively from correla
`tors 319 and 320. The outputs of the multipliers 314 and 315
`are combined together in a Summer 316 and Supplied to an
`adaptive equalizer 321 to produce a decision output Symbol
`A.
`A Second tapped-delay line is formed by delay elements
`317 and 318 to introduce delay times m and t, successively
`to the output of combiner 311. The first inputs of correlators
`319 and 320 are connected respectively to the input and
`output of delay element 318 and their second inputs are
`connected together to the output of the adaptive equalizer
`321 for ISI cancellation. The delay time m introduced by
`delay element 317 corresponds to the amount of time for a
`data symbol to travel from the point of entry to the matched
`filter 312 to the point of delivery from the adaptive equalizer
`321, whereby each data Symbol appearing at the respective
`tap of the Second delay line is made to coincide in time with
`a decision Symbol A. With this timing alignment, correla
`tors 319 and 320 derive first and second complex correlation
`signals W and W respectively from a first correlation
`between a decision symbol A and a m-delayed data
`Symbol r,
`and a Second correlation between an earlier
`decision Symbol A, and the (m+1)-delayed earlier data
`Symbol r, AS in the previous embodiment, the correlation
`detectors 319 and 320 provide the time-averaging of the
`
`ERICSSON v. UNILOC
`Ex. 1014 / Page 9 of 11
`
`

`

`6,075,808
`
`8
`-continued
`= h(to + i)Ea. . a = h(to + i)
`
`5
`
`The impulse response of the matched filter 312 is therefore
`given by:
`
`Equation (19) is an estimate of the channel impulse
`response, which is a time-reversed complex conjugate of the
`channel impulse response. By using the tap-weight values
`W and W. given by Equations (17) and (18), the tap signals
`on the first tapped-delay line are weighted, producing a
`convolutional filter response Y at the output of the matched
`filter 312 as follows:
`
`7
`products of their input and output signal for a duration that
`is sufficiently longer than the symbol interval but sufficiently
`smaller than the interval at which a deep fade is likely to
`OCC.
`The outputs of the demodulators 310A and 310B are
`maximal-ratio diversity combined in the matched filter 312
`as will be understood by the following description.
`The impulse response at the output of combiner 311 is
`given in the form:
`
`where 8(S)=1 if S=to or to--T, and 0 if Szto or to--T. The data
`symbol r, at the input of the matched filter 312 is represented
`by the convolutional integration of the impulse response
`H(t) by the transmitted symbol an as follows:
`
`15
`
`(13)
`
`Y = n
`
`: W + r. W2
`
`(20)
`
`where H represents the n-th Sample of impulse response H
`when the latter is sampled at the symbol rate. Since the
`impulse response H is only valid at times to and to-t, the
`Symbol r, produced in response to the transmitted Symbola,
`at the output of the delay element 313 is given in the form:
`
`25
`
`where M is the number of symbols present in the delay time
`T.
`The first term of Equation (13) represents the main
`component caused by the transmitted Symbol a, and the
`second term is the ISI component caused by an earlier
`transmitted Symbola. Therefore, the later-arriving Sym
`35
`bol r
`at the input of delay element 313 is expressed as:
`
`(15)
`rf-h(to) art-h(tott)'a,
`From the Second tapped delay line, m-delayed versions of
`these Symbols r,
`and r, are respectively Supplied to
`correlators 319 and 320. In time coincidence, a decision
`output symbol A, is supplied to the correlators 319 and 320
`to produce tap-weight Signals W and W, respectively. The
`tap-weight signal W is obtained by Solving the following
`Equation:
`
`W2 = Er. A
`
`(16)
`
`= Eh (to) a + h(to + i) at . A
`= h(to) Ea. A + ih (to + i)Ea.
`
`A.
`
`Since the decision Symbol A, is Substantially equal to the
`transmitted Symbol a, the Symbol's autocorrelation
`Ea, all can be represented by Kronecker delta 6-i(if i=j)
`or 0(if iz), and Equation (16) can be rewritten as:
`(17)
`W=h(t)Ea'ah (to)
`In a similar manner, the tap-weight Signal W is obtained by
`Solving the following Equation:
`
`W = Er. A
`
`(18)
`
`. A
`= Eh.(to) at +h(to + i), a
`= h(to)Ea. A + ih (to + i)Ea. A
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`The Second term of Equation (20) represents the maximal
`ratio combined output of the Signals of the time diversity
`branches RA and RB and the coefficient h(t)h(t)+h(t+
`t)h(to+t) is the main convolutional impulse response of the
`matched filter 312 (see FIG. 5). The first and third terms of
`the equation are the interSymbol interferences from Symbols
`a,
`and a
`respectively, and their coefficients h(to+t)
`h(t) and h(t)h(t+T) are the precursor and postcursor
`convolutional impulse responses of the matched filter having
`a Rayleigh fading related, random phase and amplitude.
`Therefore, the power level of the first and third terms is
`Significantly lower than the maximal-ratio combined com
`ponent as illustrated in FIG. 5.
`What is claimed is:
`1. A radio communications System comprising:
`a multiplexer circuit for producing a plurality of
`quadrature-modulated Spread spectrum information
`bearing Signals using quadrature carriers of Single
`frequency and a plurality of spreading codes, and
`combining the quadrature-modulated Spread spectrum
`information-bearing Signals to produce a code division
`multiplex Signal, there being a predetermined inter
`Signal time delay between said quadrature-modulated
`Spread spectrum information-bearing Signals,
`a transmitter for up-converting the code division multi
`plex Signal to a radio-frequency Signal;
`a transmit antenna for transmitting the radio-frequency
`Signal;
`a receive antenna for receiving the transmitted Signal;
`a receiver for down-converting the received Signal to
`recover Said code division multiplex Signal;
`a demultiplexer circuit for converting a plurality of
`quadrature-modulated component Signals contained in
`the recovered multiplex Signal to a plurality of
`information-bearing Signals using quadrature carriers
`of Said Single frequency and a plurality of despreading
`codes respectively identical to the spreading codes,
`a first combiner for combining Said plurality of
`information-bearing Signals into a combined
`information-bearing Signal;
`a delay line having a plurality of Successive taps for
`receiving the combined information-bearing Signal, the
`
`ERICSSON v. UNILOC
`Ex. 1014 / Page 10 of 11
`
`

`

`9
`delay time between the Successive taps corresponding
`to Said predetermined inter-signal time delay;
`a plurality of complex multipliers for respectively weight
`ing tap Signals from the taps of the delay line by
`complex weighting factors,
`a Second combiner for combining the weighted tap Signals
`to produce a combined weighted Signal;
`an adaptive equalizer for producing a decision output
`from the combined weighted Signal; and
`a plurality of correlation detectors for respectively deriv
`ing Said plurality of complex weighting factors from
`correlations between the decision output of the adaptive
`equalizer and a plurality of delayed versions of tap
`Signals at the taps of the delay line, one of the delayed
`versions being time coincident with Said decision out
`put.
`2. A radio communications System as claimed in claim 1,
`wherein Said predetermined inter-signal time delay is greater
`than Symbol interval of the information-bearing Signals.
`3. A radio communications System as claimed in claim 1,
`wherein each of Said correlation detectors multiplies the
`decision output by one of the delayed versions of tap Signals
`to produce complex products and averages the complex
`products during a time interval longer than Symbol interval
`of the tap Signals but shorter than intervals at which fading
`is likely to occur.
`4. A receiving apparatus for a radio communications
`System wherein a plurality of quadrature-modulated spread
`Spectrum information-bearing Signals are produced using
`quadrature carriers of Single frequency and a plurality of
`Spreading codes, and the quadrature-modulated spread Spec
`trum information-bearing Signals are combined into a code
`division multiplex Signal which is up-converted to a radio
`frequency Signal and transmitted, there being a predeter
`mined inter-Signal time delay between Said quadrature
`modulated spread spectrum information-bearing Signals, the
`apparatus comprising:
`an antenna for receiving the transmitted radio-frequency
`Signal;
`a receiver for down-converting the received radio
`frequency Signal to recover Said code division multi
`plex Signal;
`a demultiplexer circuit for converting a plurality of
`quadrature-modulated component Signals contained in
`the recovered multiplex Signal to a plurality of
`information-bearing Signals using quadrature carriers
`of Said Single frequency and a plurality of despreading
`codes respectively identical to the spreading codes,
`a first combiner for combining Said plurality of
`information-bearing Signals into a combined
`information-bearing Signal;
`a delay line having a plurality of Successive taps for
`receiving the combined information-bearing Signal, the
`delay time between the Successive taps corresponding
`to Said predetermined inter-signal time delay;
`a plurality of complex multipliers for respectively weight
`ing tap Signals from the taps of the delay line by
`complex weighting factors,
`a Second combiner for combining the weighted tap Signals
`to produce a combined weighted Signal;
`an adaptive equalizer for producing a decision output
`from the combined weighted Signal; and
`a plurality of correlation detectors for respectively deriv
`ing Said plurality of complex weighting factors from
`correlations between the decision output of the adaptive
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