United States Patent (19)
`Jasper et al.
`
`(54) MAXIMALRATIO DIVERSITY COMBINING
`TECHNIQUE
`
`75) Inventors: Steven C. Jasper, Hoffman Estates;
`Mark A. Birchler, Roselle, both of
`Il.
`
`73 Assignee: Motorola, Inc., Schaumburg, Ill.
`
`21 Appl. No.: 663,777
`
`22 Filed:
`
`Mar. 4, 1991
`
`Related U.S. Application Data
`Continuation of Ser. No. 536,839, Jun. 12, 1990, aban
`doned.
`
`63)
`
`51) Int. Cli ............................................... H04B 1/18
`52 U.S. C. .................................... 375/100; 37.5/102;
`455/37.
`58) Field of Search ....................... 375/14, 39, 40, 98,
`375/100, 101, 102; 455/52, 133, 134, 136, 137,
`139
`
`HHHHHHHHHHHHHIII
`USOO540615A
`5,140,615
`11
`Patent Number:
`Aug. 18, 1992
`45) Date of Patent:
`
`56
`
`References Cited
`U.S. PATENT DOCUMENTS
`3,633,107 1/1972 Brady .................................... 375/40
`4,577,332 3/1986 Brenig ................................. 375/100
`4,675,880 6/1987 Davarian ............................. 375/01
`4,733,402 3/1988 Monsen ............................... 375/100
`4,953,183 8/1990 Bergmans et al. .................. 375/10
`Primary Examiner-Benedict V. Safourek
`Assistant Examiner-Young Tse
`Attorney, Agent, or Firm-John W. Hayes .
`57
`ABSTRACT
`A method for. implementing diversity reception to
`counteract effects of channel fading on a transmitted
`information signal. In diversity receive paths, estimates
`of complex channel gain are computed based upon pilot
`symbols inserted from time to time in the transmitted
`information symbol stream. Phase corrected and
`weighted samples from the diversity paths are summed
`prior to the decision process. The squared magnitudes
`of the diversity path channel gains are summed to pro
`vide the proper threshold adjustment.
`
`1 Claim, 2 Drawing Sheets
`
`COMPLEX PILOT SYMBOL
`
`
`
`
`
`405
`
`T NOMINAL
`EvalA08
`THRESHOLD
`Ot2
`INTERPOLATION
`FILTER
`PILOT
`406
`SAVPLER
`COMPLEX CHANNEL
`CAIN ESTIVATE
`03
`
`412
`
`ADJUSTED
`
`COWPOSITE
`DOWN CONVERTER
`SIGNAL CENTERED
`
`
`
`
`
`
`
`
`
`
`
`
`
`
`
`RECEIVER
`PULSE-SHAPING
`FILTER
`
`SYuso
`SAPER
`
`
`
`INFO. SYBOLS TO BE COPENSATED
`
`PHASE CORRECTED
`SYBOL SCALED BY
`SQUARE OF CHANNEL AWPLITUDE
`
`
`
`DETECTED
`SYBOL
`
`IPR2020-00038
`MM EX1031, Page 1
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`

`

`U.S. Patent
`
`Aug. 18, 1992
`
`Sheet 1 of 2
`
`5,140,615
`
`TRANSMITTER-10
`-102
`VOICE
`
`107
`
`- YES
`
`110
`
`100
`- PROR ART -
`AVC. 7
`
`DETECTOR
`
`12 N.
`OUTPUT
`
`
`
`o o o
`
`201
`PILOT SYMBOLS
`2|| || || ET || || || 3 || || || ET || || 1
`2
`2
`2
`2
`2
`2
`2
`2
`2
`2
`2
`2
`2
`2
`2 oo o
`2
`2.
`2
`2
`2
`22222
`2
`2
`2
`2
`4|-T
`M=8
`DATA SYMBOLS
`202
`
`A (2.2
`
`AA 6. 3
`Q PILOT SYMBOL
`
`
`
`COMPLEX
`CHANNEL
`GAIN
`
`
`
`IMAG
`
`PILO SAMPLES
`
`w
`
`INTERPOLATED
`CHANNEL GAIN
`ESTIMATES
`REAL
`
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`U.S. Patent
`
`Aug. 18, 1992
`
`Sheet 2 of 2
`
`5,140,615
`
`COMPEX PILOT SYMBOL
`
`
`
`
`
`
`
`405
`
`ELA08
`INTERPOLATION
`PILOT
`FILTER
`SAPER 406
`COPLEX CHANNEL
`CAIN ESTIVATE
`403
`
`A77 6.4
`NOMINAL
`THRESHOLD
`Ot2
`
`COWPOSITE
`DOWN CONVERTER
`SIGNAL CENTERED
`A DC
`
`
`
`401
`RECEIVER
`PULSE-SHAPINC
`FILTER
`
`
`
`
`
`
`
`
`
`
`412
`
`ADJUSTED
`THRESHOLDS
`
`413
`
`DETECTED
`SYBOL
`
`SYuso
`SAPER
`407
`
`
`
`INFO, SYBOLS TO BE COPENSATED-
`
`PHASE CORRECTED
`SYBOL SCALED BY
`SQUARE OF CHANNEL AMPLITUDE
`
`PILOT
`01 o- INTERPOLATION
`FILTER
`PILOT
`SAPER
`
`601
`
`COPLEX
`CONJUGATE
`
`RX
`SIBOS
`
`DELAY
`DELAY
`
`(X)
`
`PILOT
`O1 O-INTERPOLATION
`602-N PILOT
`FILTER
`SANPLER
`
`COMPLEX
`CONJUGATE
`
`A 6.6
`
`NOMINAL
`THRESHOLD
`
`604
`
`
`
`RX2 - -
`SYBOLS
`
`- DELAY
`
`(X)
`
`G
`
`DECISION
`
`PILOT
`O1 O-INTERPOLATION
`603- PILOT
`FILTER
`SAPER
`
`sis -
`
`o2
`
`COPLEX
`CONUCATE
`
`(X)
`
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`1
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`5
`
`MAXIMAL RATO OVERSITY COMBINING
`TECHNIQUE
`This is a continuation of application Ser. No.
`07/536,839, filed Jun. 12, 1990, and now abandoned.
`TECHNICAL FIELD
`This invention relates generally to RF communica
`tion systems, and in particular to a diversity combining
`10
`technique used to combat the effects of fading in RF
`communication systems.
`BACKGROUND OF THE INVENTION
`RF (radio frequency) communication systems are
`subject to fading phenomena which produce noise and
`distortion in received information signals. One ap
`proach to overcoming the effects of fading is the use of
`a diversity receiver system.
`In a diversity system, more than one receive antenna
`20
`is employed. Even if a signal fade occurs at one antenna,
`there is a finite probability that a signal of relatively
`strong amplitude will still be appearing at one or more
`of the other antennas in the diversity system. The use of
`multiple antennas allows the receiver to operate on two
`25
`or more independent versions of the transmitted signal,
`and to combine or select these signals in such a way as
`to mitigate the effects of fading. The more independent
`the fading processes associated with each branch, the
`better the combiner performance.
`A method well known in the art for combining these
`branch signals is called maximum ratio combining, or
`max-ratio. In the max-ratio technique, the branch sig
`nals are adjusted so that they are in phase, and then they
`are weighted in proportion to their individual signal-to
`noise ratios before being summed. Of course, in utilizing
`this technique, a reference of some sort must be estab
`lished in order to phase compensate the received sig
`nals. This reference is commonly established through
`the transmission of a pilot tone. A pilot tone is simply a
`signal that is transmitted at a frequency adjacent to the
`modulated signal, from which the carrier phase and
`amplitude are estimated. Unfortunately, the use of a
`pilot tone carries with it the disadvantage of an increase
`in the required transmission bandwidth, as well as a
`45
`higher peak-to-average ratio of the transmitted signal.
`A pilot tone system is also susceptible to decorrelation
`between the pilot tone and the information bearing
`signal. This can lead to errors in the phase adjustment
`and weighting processes.
`Accordingly, a need arises for a method for max-ratio
`combining in a diversity system which does not suffer
`from higher bandwidth requirements nor the decorrela
`tion and peak to average problems associated with the
`separate pilot tone technique.
`SUMMARY OF THE INVENTION
`The above-described need is satisfied through a
`method for implementing diversity reception to coun
`teract the effects of channel fading on a transmitted
`information signal. At a transmitter, one or more prede
`termined pilot symbols are inserted, from time to time,
`into a quadrature amplitude modulated (QAM) infor
`mation stream, and an RF carrier is modulated with the
`QAM information stream to provide a transmitted sig
`65
`nal. Within each diversity path of a receiver designed to
`receive the transmitted information signal, the received
`signal is demodulated and appropriately sampled to
`
`5,140,615
`2
`provide a sampled, demodulated signal. The pilot sym
`bol samples and information symbol samples are sepa
`rated, and the pilot symbol samples are processed at
`predetermined pilot sample times to determine estimates
`of complex channel gain at each pilot sample time.
`Since the information symbol sample rate is higher than
`the pilot symbol sample rate, the complex channel gains
`are interpolated to compute complex channel gain esti
`mates for each information symbol sample time.
`Each information symbol sample is then multiplied by
`the complex conjugate of the corresponding complex
`channel gain estimate to provide a phase corrected and
`weighted estimate of the transmitted information sym
`bol, where the weighting is approximately the square of
`the amplitude of the complex channel gain. The square
`of the amplitude of the estimated complex channel gain
`is also calculated.
`Then, operating on the outputs of the diversity re
`ceive paths, the phase corrected and weighted informa
`tion symbol estimates are summed to provide a diversity
`combined information symbol estimate. By summing
`the squares of the amplitudes of the estimated complex
`channel gains, an estimate of composite weighting fac
`tor is derived. Using the composite weighting factor, a
`set of adjusted decision thresholds is derived, and esti
`mates of the originally transmitted information symbols
`are obtained through comparing the diversity combined
`information symbol estimate to the adjusted decision
`threshold.
`BRIEF DESCRIPTION OF THE DRAWINGS
`FIG. 1 illustrates a max-ratio combining technique of
`the prior art;
`FIG. 2 illustrates pilot symbols interleaved with data
`symbols in a 16-QAM information stream;
`FIG. 3 is a 16-QAM signal constellation indicating a
`predetermined pilot symbol;
`FIG. 4 is a block diagram of a receiver designed for
`operation with a 16-QAM system which phase corrects
`received symbols based on the complex channel gain
`estimate and achieves symbol detection through ad
`justed thresholds in the manner taught by the present
`invention;
`FIG. 5 illustrates interpolation of channel gain esti
`mates in between pilot symbol sample times; and
`FIG. 6 is a block diagram illustrating the maximal
`ratio diversity combining technique of the present in
`vention with three diversity receive paths.
`DETAILED DESCRIPTION OF THE
`INVENTION
`A max-ratio combining technique of the prior art is
`illustrated in FIG. 1 as generally depicted by the num
`ber 100. A voice signal (102) is applied to a transmitter
`(101) and the transmitted signal (103) is received over a
`plurality of diversity paths (104, 105 and 106). Each of
`these diversity input signals is scaled in a variable gain
`amplifier (107, 108 and 109) by an amount proportional
`to each individual signal power to noise power ratio.
`These diversity signals are adjusted so that they are in
`phase (co-phased). The co-phasing process in systems of
`the prior art is generally dependent upon a pilot tone
`included with the transmitted signal. After these co
`phased and weighted diversity signals are summed
`(110), a detector (111) yields the system's best estimate
`of the transmitted voice signal at the output (112).
`In a 16-QAM system, as contemplated by the present
`invention, the frequency domain pilot tone of the prior
`
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`5,140,615
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`4.
`art is abandoned in favor of a time domain pilot symbol.
`with the square of the amplitude of the estimated com
`FIG. 2 illustrates the insertion of pilot symbols (201) in
`plex channel gain (411). The phase corrected informa
`an input data symbol stream (202).
`tion symbols that form the output of the multiplier (410)
`In a four-level modulation system as contemplated by
`are then subjected to a decision process (413) incorpo
`the present invention, both in-phase and quadrature
`rating these adjusted thresholds to yield a best estimate
`information symbols may have any one of four possible
`of the transmitted information symbol at the output
`values. If in-phase and quadrature components are plot
`(414).
`ted in rectangular coordinates, this results in an array,
`In practicing the present invention with a plurality of
`or constellation as it is often called, of 16 possible val
`diversity receive paths, the phase corrected and
`ues; thus the term "16-QAM." This signal constellation
`10
`weighted information symbol estimates from separate
`is illustrated in FIG. 3. It is often convenient, in fact, to
`receive paths (601, 602 and 603 from FIG. 6) are
`consider in-phase and quadrature symbols as represent
`summed in a summer (605) to provide a diversity com
`ing a complex number with the in-phase axis (which is
`bined information symbol estimate. An estimate of com
`labelled I in FIG. 3) analogous to the real number axis,
`posite weighting factor is derived by summing the
`and the quadrature axis (labelled Qin FIG. 3) analogous
`squares of the amplitudes of the estimated complex
`to the imaginary number axis. An arbitrary, although
`channel gains in another summer (604). The composite
`necessarily predetermined, pilot symbol is also indi
`weighting factor is multiplied by the nominal thresholds
`cated in the figure.
`to derive a set of adjusted decision thresholds, and the
`FIG. 4 is a block diagram of a receiver path designed
`diversity combined information symbol estimate which
`to process QAM information symbols and pilot symbols
`20
`is the output of the first summer (605) is subjected to a
`using the method taught by the present invention. The
`decision process (606) utilizing the adjusted decision
`receiver path is generally depicted by the number 400.
`thresholds to obtain estimates of the originally transmit
`It is considered most advantageous to implement the
`ted information symbols.
`method taught by the present invention in a digital radio
`system employing a digital signal processor (DSP).
`What is claimed is:
`25
`1. In an RF communication system using quadrature
`This, of course, is not necessary, but it makes the imple
`amplitude modulation (QAM), a method for implement
`mentation much simpler. FIG. 4 shows a composite
`ing diversity reception to counteract effects of channel
`down-converted signal centered at DC as the input to a
`receiver pulse shaping filter (401). The input symbols
`fading on a transmitted information signal, the method
`comprising the steps of:
`are then sampled by a symbol sampler (402) at a prede
`30
`termined rate, then pilot symbols and information sym
`at a transmitter:
`bols are separated and directed along different paths.
`(a) inserting, from time to time, one or more prede
`Pilot symbols are directed along path 403 while infor
`termined pilot symbols into a QAM information
`mation symbols are directed along path 404.
`stream;
`The first element depicted along the pilot symbol
`(b) modulating an RF carrier with the QAM infor
`35
`path (403) is a pilot sampler (405) which samples pilot
`mation stream to provide a transmitted signal;
`symbols at predetermined times. The sampled pilot
`within two or more diversity receive paths:
`symbols are then multiplied by the inverse of the known
`(c) receiving the transmitted signal to provide a
`pilot symbol in a mixer (406) in order to determine how
`received signal;
`the pilot symbols have been affected by the communica
`(d) demodulating and appropriately sampling the
`tion channel. The resulting estimate of the channel ef
`received signal to provide a sampled demodu
`fect is known as the complex channel gain estimate. A
`lated signal;
`pilot interpolation filter (408) provides estimates of the
`(e) separating pilot symbol samples from informa
`complex channel gain at the information symbol sam
`tion symbol samples in the sampled demodulated
`pling times. This interpolation is necessary because pilot
`45
`signal obtained in step (d);
`symbols were inserted in the information symbol stream
`(f) processing pilot symbol samples at predeter
`only at widely spaced intervals, so the information sym
`mined pilot sample times to determine estimates
`bol sampling rate is correspondingly much faster than
`of complex channel gain at each pilot sample
`the pilot symbol sampling rate.
`time;
`Turning briefly to FIG. 5, the complex channel gain
`(g) interpolating complex channel gains deter
`is plotted in rectangular coordinates around orthogonal
`mined in step (f) to compute complex channel
`real and imaginary axes. FIG. 5 illustrates interpolated
`gain estimates for each information symbol sam
`channel gain estimates between the pilot sample times.
`ple time;
`Returning to FIG. 4, information symbols which are
`(h) multiplying each information symbol sample by
`processed through the information symbol path (404),
`55
`the complex conjugate of each corresponding
`are first subject to a delay (407) to compensate for pilot
`complex channel gain estimate to provide a
`sample processing that occurs in the pilot sample path
`phase-corrected and weighted estimate of the
`(403). Each information symbol sample is multiplied in
`transmitted information symbol, where the
`a multiplier (410) by the complex conjugate of the cor
`weighting is approximately the square of the
`responding complex channel gain estimate illustrated in
`amplitude of the complex channel gain;
`block 409. The phase corrected information signal,
`(i) calculating the square of the amplitude of the
`which is an output of the multiplier (410), is still
`estimated complex channel gain;
`weighted by channel effects. However, this weighting
`factor is closely approximately by the square of the
`then, operating on outputs of said two or more diver
`amplitude of the complex channel gain illustrated in
`sity receive paths:
`() summing the phase-corrected and weighted
`block 411. In a single receiver, the nominal thresholds
`predetermined by the four possible information symbol
`information symbol estimates to provide a diver
`levels are adjusted by multiplication in a multiplier (412)
`sity combined information symbol estimate;
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`5,140,615
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`(k) deriving an estimate of composite weighting
`factor by summing the squares of the amplitudes
`of the estimated complex channel gains;
`(1) using the composite weighting factor deter-
`mined in step (k) to derive a set of adjusted deci- 5
`sion thresholds; and
`(m) comparing the diversity combined information
`
`6
`symbol estimate provided in step (j) to the ad
`justed decision thresholds of step (1) to obtain
`estimates of the originally transmitted informa
`tion symbols.
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`as
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