United States Patent (19)
`Borth et al.
`
`11
`45
`
`Patent Number:
`Date of Patent:
`
`4,829,543
`May 9, 1989
`
`(54) PHASE-COHERENT TDMA QUADRATURE
`RECEIVER FOR MULTIPATH FADING
`CHANNELS
`75) Inventors: David E. Borth, Palatine; Chih-Fei
`Wang, Arlington Heights; Duane C.
`Rabe, Rolling Meadows; Gerald P.
`Labedz, Chicago, all of Ill.
`73 Assignee: Motorola, Inc., Schaumburg, Ill.
`21 Appl. No.: 128,976
`(22
`Filed:
`Dec. 4, 1987
`
`51) Int. Cl.' ...
`
`(56)
`
`References Cited
`U.S. PATENT DOCUMENTS
`4,475,214 10/1984 Gutleber ............................... 375/96
`4,484,335 11/1984 Mosley et al. ...
`... 375/96
`4,575,861 3/1986 Leureault.........
`... 37.5/15
`4,587,662 5/1986 Langewellpott ........................ 375/
`4,606051 8/1986 Crabtree et al. .
`... 375/86
`4,633,426 12/1986 Venier ..........
`... 375/96
`4,669,091 5/1987 Nossen .....
`... 375/14
`4,669,092 5/1987 Sari et al. .....
`... 375/14
`4,672,638 6/1987 Taguchi et ai.
`... 375/99
`
`4,691,326 9/1987 Tsuchiya m - as 375/1
`FOREIGN PATENT DOCUMENTS
`0174125 3/1986 European Pat. Off. .
`018401 18 6/1986 European Pat. Off. .
`OTHER PUBLICATIONS
`Andreas Polydoros, “A Unified Approach to Serial
`Search Spread Spectrum Code Acquisition'-Part II,
`A Match Filter Receiver, IEEE Transaction, vol. Com
`32, No. 5, May 1984.
`d
`Price et al.; "A Communication Technique for Multi
`path Channels', Proceedings of the IRE; vol. 46, Mar.
`1958; pp. 555-570.
`Turin; "Introduction to Spread-Spectrum Antimul
`tipath Techniques and their Application to Urban Digi
`tal Radio', Proceedings of the IEEE; vol. 68, No. 3,
`Mar. 1980; pp. 328-353.
`Eckert et al.; "The Fully Digital Cellular Radiotele
`
`y w a y
`
`w as w w
`
`w
`
`a
`
`&
`
`H04L27/22
`
`52 U.S. Cl. ........................................ 375/83; 375/96;
`329/112
`58 Field of Search ....................... 375/1, 115, 96, 39,
`375/77, 83; 364/604, 728, 819; 329/104, 112,
`110; 370/104
`
`phone System CD900'; Nordic Seminar on Digital
`Land Mobile Radiocommunication; Feb. 5-7, 1985,
`Espoo, Finland; pp. 249-259.
`Langewellpott et al.; "Performance Analysis of Radio
`Transmission in the Fully Digital Cellular Radio Sys
`tem CD900'; Nordic Seminar on Digital Land Mobile
`Radiocommunications; Feb. 5-7, 1985, Espoo, Finland;
`pp. 261-269.
`Leuenberger; "Digital Radio Systems Examined-Part I,
`General State of the Art'; MSN & CT; vol. 16, No. 1;
`Jan. 1986; pp. 81-92.
`Raith et al.; "Multi-Path Equalization for Digital Cellu
`lar Radio Operating at 300 KBit/S”; 36th IEEE VTG
`Conference, Jun. 1986; pp. 268-272.
`Inmos; "Cascadable Signal Processor TMS A100';
`Catalog Sheet, Jul. 1986.
`Stjernvallet al., “Radio Test Performance of a Narrow
`band TDMA System'; 37th IEEE VTG Conference;
`Jun. 1-3, 1987; pp. 293–299.
`Stjernvallet al.; "Radio Test Performance of a Narrow
`band TDMA System-DMS 90”; International Confer
`ence on Digital Land Mobile Radio Communications;
`Venice; Jun. 30-Jul. 3, 1987; pp. 310-317.
`Kammeyer; "Equalization Problems in a Digital FM
`Receiver'; Signal Processing vol. 9, No. 4; Dec. 1985;
`pp. 263-276.
`Primary Examiner-Robert L. Griffith
`Assistant Examiner-Stephen Chin
`Attorney, Agent, or Firm-Raymond A. Jenski; Rolland
`R. Hackbart
`ABSTRACT
`57
`A method and apparatus for phase-coherently demodu
`lating a multipath-impaired time division multiple ac
`cess QPSK data timeslot is disclosed. A quadrature
`separator generates multipath-impaired intermediate
`signals which, during a predetermined synchronizing
`sequence for the timeslot, are applied to a pair of syn
`chronizing correlators to generate quadrature multipath
`profiles. These profiles then are used to modify subse
`quently received QPSK timeslot signals at the separator
`outputs to coherently construct multipath compensated
`I and Q channel data.
`14 Claims, 5 Drawing Sheets
`
`315
`M/PATH CORREL
`
`
`
`M/PATH CORREL
`
`c(t)
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`U.S. Patent
`
`May 9, 1989
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`Sheet 1 of 5
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`4,829,543
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`
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`WIWO
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`U.S. Patent
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`May 9, 1989
`
`Sheet 2 of 5
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`4,829.543
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`US. Patent
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`May 9, 1989
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`U.S. Patent
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`May 9, 1989
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`Sheet 4 of 5
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`4,829,543
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`U.S. Patent May 9, 1989
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`Sheet 5 of 5
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`4,829,543
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`4-09
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`MAGNTUDE
`COMPARATOR
`
`415
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`SEQUENCE
`CONTROLLER
`
`MAGNTUDE
`COMPARATOR
`
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`1.
`
`PHASE-COHERENT TDMA QUADRATURE
`RECEIVER FOR MULTIPATH FADING
`CHANNELS
`
`O
`
`45
`
`4,829,543
`2
`channel profile including the significant multipath con
`tributions. Even though the data receiver may be mov
`ing, the channel profile can be assumed to undergo
`negligible changes in a given TDMA timeslot if the
`timeslot is sufficiently short in duration.
`SUMMARY OF THE INVENTION
`Therefore, it is one object of the present invention to
`compensate for multipath distortion introduced by a
`radio channel to transmitted digital information.
`It is another object of the present invention to create
`a channel profile to adapt the receiver demodulator to
`correct for the multipath distortion.
`It is a further object of the present invention to create
`the radio channel multipath profile model from a syn
`chronization sequence transmitted during part of one
`timeslot of a TDMA transmission and utilize that model
`during the entire timeslot period.
`Accordingly, these and other objects are achieved in
`the present invention which encompasses a time divi
`sion multiple access (TDMA) radio system receiver
`which utilizes a synchronizing sequence contained
`within an assigned timeslot of message data to adapt a
`multi-phase demodulator to the finite impulse response
`of the radio channel during the assigned timeslot. A first
`phase of the synchronizing sequence is correlated to a
`predetermined sequence to create a first correlator sig
`nal. Likewise, a second phase of the synchronizing se
`quence is correlated to the predetermined sequence to
`create a second correlator signal. The first and second
`correlator signals are then correlated with first and
`second phases of received message data and the result
`ing outputs of correlation are combined to yield first
`and second phase data.
`BRIEF DESCRIPTION OF THE DRAWINGS
`FIG. 1 is a block diagram of a data transmission sys
`tem employing quadrature digital transmission and re
`ception.
`FIGS. 2A and 2B are, together, a block diagram of a
`TDMA receiver which may receive QPSK signals.
`FIG. 3 is a block diagram of a TDMA receiver signal
`processor which may advantageously employ the pres
`ent invention.
`FIG. 4 is a block diagram of the peak detector circuit
`of the receiver of FIG. 3.
`FIG. 5 is a graph of the synchronization correlator
`outputs Cr(t) and Co(t) plotted against time and show
`ing a possible set of outputs including a correlation
`detection.
`DESCRIPTION OF THE PREFERRED
`EMBODIMENT
`A radio frequency system conveying a data system
`from a transmitter 101 to a receiver 103 is shown in
`FIG. 1. In the preferred embodiment, quadrature phase
`shift keying (QPSK) is employed to increase the
`throughput of the channel although other multi-dimen
`sional signaling may equivalently be employed. Fur
`ther, the well-known time division multiple access
`(TDMA) technique of sharing a limited channel re
`Source among a large number of users is employed in
`the present invention. Each of the users is assigned a
`brief period of time (a timeslot) during which a message
`may be transmitted to or received from the user. The
`advantages of such a TDMA technique over other tech
`niques (such as frequency division multiple access
`
`BACKGROUND OF THE INVENTION
`This invention relates generally to digital radio re
`ceivers and more specifically to receivers receiving and
`demodulating TDMA QPSK modulation in a multipath
`fading environment such as an environment where the
`receivers may be in motion. This invention is related to
`instant assignee's U.S. patent applications "Rapid Refer
`ence Acquisition and Phase Error Compensation for
`Radio Transmission of Data', filed on behalf of Labedz
`15
`et al. and "TMDA Radio System Employing BPSK
`Synchronization for QPSK Signals Subject to Random
`Phase Variation and Multipath Fading', filed on behalf
`of Borth et al. on the same date as the present invention
`and containing related subject matter.
`In a typical environment, a UHF or microwave radio
`channel exhibits a multipath structure in addition to
`Rayleigh fading. Thus, a radio receiver for a mobile or
`portable TDMA system operating at high speed data
`rates must accommodate reception of multiple replicas
`of the transmitted signal, each with a random magni
`25
`tude, phase, and time delay with respect to the transmit
`ted signal. Without corrective measures, the data mes
`sage can be obliterated by the multipath signals. As
`early as 1958, a receiver capable of accommodating
`these impairments was described for the use of either
`30
`Differential Binary Phase-Shift Keying (DBPSK) or
`noncoherent Frequency-Shift Keying (FSK). It em
`ployed a channel sounding method to estimate the chan
`nel impulse response or channel profile, and a transver
`sal equalizer having taps which were adjusted in re
`35
`sponse to the estimated channel profile. By 1960 the
`multipath channel had been exhaustively studied and
`simulated, and optimum reception had been defined, but
`largely avoiding phase coherent techniques. Such de
`modulation techniques do not permit the use of higher
`40
`spectral efficiency modulation methods which employ
`two-dimensional signaling techniques such as shaped
`Quadrature Phase-Shift Keying (QPSK) and its varia
`tions.
`By 1983, TDMA (Time Division Multiple Access)
`receivers for digital telephony using Binary Phase-Shift
`Keying (BPSK) phase coherent detection had been
`described in U.S. Pat. No. 4,587,662. In 1985 this was
`extended to include QPSK, but the receiver was only
`described in general terms.
`In 1986 an MSK receiver, with possible application to
`QPSK, was reported which could accommodate two
`rays of multipath and which used an adaptive equalizer
`employing both feedforward and feedback filtering.
`(See Krister Raith et al., “Multi-Path Equalization for
`Digital Cellular Radio Operating at 300 kbits/s', 36th
`IEEE Vehicular Conference, pp. 2682.272, May 1986).
`Although this adaptive equalizer apparently has never
`been thoroughly described in the literature, it is differ
`ent than the multipath correlation employed in the pres
`ent invention since it requires decisions to be made on
`the output in order to adjust the equalizer.
`Adaptive equalization generally operating continu
`ously on the data being received has been utilized in
`digital microwave receivers receiving continuous data
`65
`streams. Such continuous receivers can equalize over a
`relatively long period of time. TDMA, due to its burst
`like characteristics, demands rapid determination of the
`
`50
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`O
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`35
`
`4,829,543
`4.
`3
`Timing controller 231 functions as a phase-locked
`TDMA) are: (a) no duplexer is required for full duplex
`loop (PLL), using a stable timing reference to validate
`communications, (b) variable data rate transmission
`may be accommodated through the use of multiple
`the timeslot detect signal and provide a validated detect
`adjacent time slots, (c) a common radio frequency
`output signal. The validated timeslot detect signal is
`power amplifier may be used to amplify multiple chan
`applied to AND gate 233 along with a bit clock output.
`nels at any power level without the combining losses or
`The combined timeslot detect/bit clock signal is then
`intermodulation distortion present with FDMA, and (d)
`routed to the I and Q signal buffers 217 and 219, respec
`a capability of scanning other "channels' (timeslots)
`tively. Data signals are clocked into signal buffers 217
`without requiring separate receivers may be provided.
`and 219 using the combined detect/bit clock signal.
`The high data rate employed in the present invention
`In the implementation shown in FIGS. 2A and 2B, a
`(200 Kbps to 2 Mbps) exceeds the channel coherence
`conventional baseband synchronous decision feedback
`equalizer (DFE) 234 is employed for data signal recov
`bandwidth of the mobile radio channel for many urban
`and suburban environments. As a result, the channel
`ery. The DFE 234 basically consists of two parts: a
`exhibits a multipath structure in addition to the ex
`forward linear transversal filter 235 and a feedback
`pected Rayleigh fading. The receiver of the present
`linear transversal filter 237. The forward filter 235 at
`15
`invention enables TDMA quadrature signals to be co
`tempts to minimize the mean-square-error (MSE) due to
`herently received over a multipath fading channel. This .
`intersymbol interference (ISI), while the feedback filter
`237 attempts to remove the ISI due to previously de
`embodiment will demodulate a 2-megachip/sec QPSK
`radio signal, the only constraint being that the acquisi
`tected symbols.
`tion sequence be transmitted as a binary phase shift
`The decision feedback equalizer 234 structure is
`20
`keying (BPSK) signal with a predetermined phase rela
`adapted at least once each time slot in order to compen
`sate for the effects of the time-varying multipath profile.
`tive to the QPSK data.
`The equalized and quantized complex data output from
`FIGS. 2A and 2B are a block diagram of a TDMA
`receiver which may be employed to recover TDMA
`quantizer 238 is applied to multiplexer 239 for 2:1 multi
`quadrature phase shift keying data and is described in
`plexing together with the data clock and output as an
`25
`instant assignee's U.S. Pat. application No. 009,973
`output data word.
`"TDMA Communications System with Adaptive
`Returning to FIG. 1, in a QPSK communication
`Equalization' filed on Feb. 2, 1987 on behalf of David
`system, a transmitted signal x(t) may be expressed as:
`E. Borth and is incorporated by reference herein.
`The digital signal outputs of the A/D converters 209
`30
`and 211, respectively, are applied to in-phase (I) time
`slot correlator 213 and quadrature (Q) correlator 215,
`respectively, as well as to their respective signal buffers
`217 and 219. I correlator 213 performs a correlation
`function between all received bits of the input signal
`and a pre-loaded synchronization word (I sync word)
`corresponding to the in-phase time slot sync word.
`The output of I correlator 213 is a digital bit stream
`representing the sample-by-sample correlation of the
`received data with the stored synchronization word
`replica for the timeslot. The correlation function exhib
`its a peak when the I sync word is located in the re
`ceived sample data. In the same way, Q correlator 215
`performs a correlation function between the pre-stored
`quadrature Q sync word from memory 221 and the
`sampled quadrature (Q) input.
`The outputs of correlators 213 and 215 are applied to
`squaring blocks 223 and 225, respectively. The squaring
`block output signals represent the squared values of the
`separate I and Q correlation operations respectively.
`50
`The squaring block outputs are then applied to sum
`ming block 227. The I and Q correlation signals are
`summed together to form a squared envelope signal
`which represents the sum of squares of the correlation
`signal. The squared envelope of the correlation signal
`55
`makes an explicit determination of the phase ambiguity
`unnecessary. This, without resolving any ambiguity, a
`large amplitude signal output from summing block 227
`represents a possible start location for a particular
`timeslot.
`The output of summing block 227 is then routed to
`timeslot detector 229, wherein the summed correlation
`signal is compared with a predetermined threshold
`value. This threshold value represents the minimum
`allowable correlation value which would represent a
`65
`detected timeslot. If the summed output is greater than
`the threshold value, a time slot detect signal is gener
`ated and applied to system timing controller 231.
`
`x(t) = a(t) cos oct+b(t) sin oct
`()
`where a(t) and b(t) are the in-phase and quadrature
`information signals and cois the carrier frequency of the
`QPSK signal in radians/sec.
`A frequency-selective (ior delay-spread) channel that
`is, a radio channel subject to multipath interference,
`may be characterized by an equivalent channel impulse
`response given by:
`
`h(t) = doö(t - TO) -- a 16(t - Ti) -- a 28(t - T2) -- . . . =
`
`(2)
`
`O d5(t - T);
`
`45
`
`where a is the amplitude of the i-th resolvable path, ti
`is the (excess) path delay associated with the i-th resolv
`able path, and m-1 is the total number of resolvable
`paths.
`For a channel input given by equation (1), the output
`of the equivalent delay-spread channel having the im
`pulse response of equation (2) is essentially constant
`during any given timeslot, and given by:
`
`(3)
`
`- c.
`
`60
`
`O ax(t - T)
`
`o aia(t - Ticosco(t - Ti) --
`se
`b(t - T)sinc)(t - Ti).
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`4,829,543
`5
`6
`where signal a T(t) (not shown) is a predetermined syn
`It is this signal, y(t), which is input to receiver 103.
`chronization sequence with good aperiodic autocorrela
`When the local oscillator reference 106 in the receiver
`tion properties, such as one of the Barker sequences.
`has a phase offset of y with respect to the (direct-path)
`The uncorrected in-phase and quadrature receiver
`received QPSK transmission, the receiver local oscilla
`branch outputs corresponding to the synchronizing
`tor reference may be given by cos (o)et--y) and is essen
`transmitted signal xT(t) may be found by substituting the
`tially constant during a TDMA timeslot. (Although the
`signal of equation (8) in the received and low pass fil
`antenna is shown connected to the mixers 107 and 111,
`tered signals UI'(t) and UQ'(t) of equations (5) and (7)
`it is likely that additional signal processing will be re
`respectively, yielding:
`quired for higher frequency radio signals. If down-con
`10
`version to an intermediate frequency is used, the output
`frequency of local oscillator may be different). Let UI(t)
`denote the output of the mixer 107 in the uncorrected
`in-phase branch of the receiver and let UI'(t) denote the
`low-pass filtered version of UI(t) output from low pass
`filter 109. Similarly, let UQ(t) denote the output of the
`mixer 111 in the uncorrected quadrature phase branch
`of the receiver and let UQ'(t) denote the low-pass fil
`tered version of UQ(t) from filter 113. UI'(t) and UQ'(t)
`are subsequently input to signal processor 115 for reso
`lution into I and Q data and then coupled to data signal
`recovery 117.
`UI(t) is given by:
`
`p
`UIr(t) = i 2. O (t)ata T(t - Ti)cos(y – acri)
`
`15
`
`and,
`
`UQ'T(t) = i 2. O ()aat(t - ri)sin(y -- acti).
`
`(9)
`
`(10)
`
`UICt) = cos(a)ct -- y) 5, dia(it - Ti)cosoc(t - ri) +
`b(t. - Ti)sincisc(t - ti)
`
`= , fo aI()a(t - ti) (cos(y + cocti +
`cos(2act + y - acti) +
`()b(t - ti) (sin2.coct + y - octi -
`siny -- acti)).
`
`25
`
`(4)
`
`30
`
`35
`
`The low-pass filtered version UI'(t) of UI(t) is given
`by:
`
`Thus UI'(t) and UQ'(t) are defined during the training
`phase as “T” as shown in equations (9) and (10).
`Referring now to FIG. 3 which illustrates the pre
`ferred embodiment of the present invention in block
`diagram form, the signals UI'T(t) and UQ'T(t) are ap
`plied to synchronization correlators (303 and 305, re
`spectively) via conventional fast A/D converters 307
`and 309. In the preferred embodiment, synchronization
`correlators 303, 305 are 4 by 32 bit digital finite impulse
`response (FIR) filters programmed to provide signed
`weighted correlation outputs. Synchronization correla
`tors 303 and 305 are realized by an IMSA100 Cascada
`ble Signal Processor available from Inmos Corp., Colo
`rado Springs, Colo. The outputs of correlators Cr(t) and
`Co(t) which are, in simple terms, weighting factors for
`each i-th resolvable path, generated during reception of
`the acquisition sequence, may have the appearance as
`shown in FIG. 5 and are given by:
`
`(11)
`
`(12)
`
`UI'(t) = x. O ()cia(t - Ti)cos(y -- ar) -
`se
`
`(5)
`
`40
`
`Cr(t) = X. O ()acos(y -- acti)6(t - Ti)
`t
`
`and,
`
`b(t - ti)sin(y + c cri)).
`Similarly UQ(t) is given by:
`
`(6)
`
`UQ(t) = sin(cut -- y) 5, aia(t - Ti)cos()(t - Ti) --
`se
`b(t - Ti)sina (t - r)
`
`= 5, at()a( - T.) (sin{2i + y - o-ri) +
`sinty + octi) + ()b(t - ti) (cos(y + cotti) -
`cos(2coct + y - oct)
`
`(7)
`
`and UO'(t) is given by:
`-
`UQ() = ()a(att - risin() + o-r) +
`b(t - Ticos(y - acri).
`Considering the operation of the present invention in
`mathematical form, it is an important feature that the
`transmitted signal xT(t) during the synchronization (or
`training) phase of the equalizer 115 is a BPSK signal.
`When transmitted in the I phase it is given by:
`xt(t)=aT(t) cos at
`
`(8)
`
`Co(t) = io ()aisin(y -- oct)6(t - Ti).
`
`45
`
`50
`
`55
`
`60
`
`65
`
`The 6 function in equations (11) and (12) determine
`when to sample the in-phase and quadrature receiver
`branch outputs and the a factor provides a weighting
`for each i-th resolvable pass contribution. In the pre
`ferred embodiment, a sequence controller 311 is real
`ized using a conventional microprocessor (such as an
`MC68020 microprocessor available from Motorola,
`Inc.) and associated memory and timing dividers. The
`sequence controller 311 loads a predetermined normal
`ized replica of the acquisition sequence (32 each 4-bit
`words) stored in the memory of sequence controller 311
`into synchronization correlators 303 and 305 prior to
`the desired TDMA timeslot to be demodulated. TDMA
`frame timing is determined by the sequence controller
`311 employing a conventional framing algorithm to
`confirm and maintain timeslot acquisition,
`Synchronization correlators 303 and 305 each corre
`late the stored acquisition sequence against the last 32
`received A/D samples, and for each new sample per
`form another complete correlation. While receiving
`noise or random data, the outputs Cr(t) and Co(t) of
`synchronization correlators 303 and 305 are small num
`bers of either polarity, emerging at the same rate as the
`
`IPR2020-00038
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`

`4,829,543
`7
`A/D sampling rate (4 per chip interval). If the radio
`channel were free of noise and not degraded by multi
`path, then when an acquisition sequence has been re
`ceived and digitized and entered into the correlators
`303 and 305, Cr(t) and Co(t) would simultaneously man
`ifest a pair (or sometimes two adjacent pairs) of signed
`numbers significantly larger than those produced by
`noise or random data, such that the root sum of Squares
`of these numbers would be proportional to the magni
`tude of the received signal, and the phase angle y rela
`tive to the local reference oscillator is:
`
`8
`
`A = C(OUT() = i, (G)a(0)aicoscy + o-ri) -
`()b(0)aicos(y + acti)sin(y + acti)
`
`(14)
`
`B = Co(t) UI"(t) = o ()a(0)aicos(y -- c.)cti)sin (ly --
`
`se
`
`(15)
`
`10
`
`15
`
`(13)
`y=arc tan (Co(t)/Cr(t)).
`In the presence of multipath, each significant path
`will result in the presence of such a peak pair appearing
`on Cr(t) and Co(t), the signs and magnitudes of each
`pair of outputs at each peak defining the delay, phase
`angle, and amplitude contribution of that path of the
`total, fulfilling the equations (11) and (12). Thus, each
`sequence of numbers Cr(t) and Co(t) are bipolar multi
`path channel profile estimates, which resemble a classic
`multipath channel profile, except that they are bipolar.
`Each of the M/PATH correlators 312, 313, 315, and
`25
`317 are FIR filters of at least 32 taps. In the preferred
`embodiment, each M/PATH correlator is realized by
`an IMS A100 Cascadable Signal Processor (available
`from Inmos Corp., Colorado Springs, Colo.) conven
`tionally connected as a correlator. During the acquisi
`30
`tion sequence at the beginning of each desired timeslot,
`Cr(t) is shifted into the TAP control entry of M/PATH
`correlators 312 and 317, and C(t) is shifted into the
`TAP control entry of M/PATH correlators 313 and
`315. Peak detector 318 is shown in FIG. 4 and com
`35
`prises a root sum of squares approximator 401 and a
`threshold detector 403 having an output which signals
`the sequence controller 311 of the first significant ray of
`multipath. The sequence controller 311 then provides
`just enough additional reference port clocks to shift this
`40
`peak all but through the M/PATH correlators, thereby
`capturing C(t) and Co(t) in their respective M/PATH
`correlators. In the preferred embodiment, the root sum
`of squares approximator 401 is realized employing a
`magnitude adder 405 which adds Cr(t) and () Co(t)
`45
`and magnitude adder 407 which adds Co(t) and
`) Cr(t). The outputs of magnitude adder 405 and
`magnitude adder 407 are input to conventional magni
`tude comparators 409 and 411, respectively, where the
`root sum of squares approximation is compared to a
`50
`predetermined threshold to generate an output to the
`sequence controller 311 (via OR gate 413). This and
`other approximations to the square root of the sum of
`the squares may be found in, eg., A. E. Filip, "A Baker's
`Dozen Magnitude Approximations and Their Detection
`55
`Statistics," IEEE Transactions on Aerospace and Elec
`tronic Systems, vol. AES-12, pp. 86-89, January 1976.
`This output to the sequence controller 311 is shown as
`td in the example of FIG. 5. Thus, the peak detector 318
`reports the first significant peak to the sequence con
`60
`troller 311 which, in turn, starts the loading at T/S stop,
`to thus capture the channel profile in each of the
`M/PATH correlators.
`The four M/PATH correlators (312, 313, 315, and
`317 in FIG. 3) thus have the information available to
`65
`perform equations (14)-(17), below, whose results (A,
`B, C, and D) appear at the outputs of M/PATH correla
`tors 312, 313, 315, and 317 respectively.
`
`cocti) - (A)b(0)aisin(y + cot)
`
`acti)sin(y + o-ri) + ()b(0)aicos(y + coeti)
`
`Properly combining the quantities A through D, one
`obtains the following expressions for the in-phase and
`quadrature outputs of the receiver at time t=0:
`
`I = A -- D = Eo ()a(0)a(cos(y + acti} +
`sincy + oct))
`
`2:
`
`is
`
`o ()a(0)ct = In-phase data.
`
`e
`
`Q = C - B = S ()b(0)a;(cos(y + acti) +
`siny + acti)
`o ()b(0)a' = Quadrature-phase data.
`
`e
`
`(18)
`
`(19)
`
`Conventional adder 331 implements equation (18) to
`produce the recovered in-phase signal I and adder 335
`implements equation (19) to produce the recovered
`quadrature signal Q, which are replicas of the transmit
`ted I and Q channel data, respectively. The outputs I
`and Q are actually four sequential numbers per chip
`interval. It is possible to intergrate them and apply a
`simple threshold for a binary data stream, or to simply
`integrate them to provide relative weighting, both at
`the original rate, or to preserve their discrete sample
`form for use in somewhat more elaborate symbol or
`character correlation.
`It can be seen by following the general input equation
`(3) through to equations (18) and (19) that the informa
`tion contained in each of the paths of the multipath
`signal is coherently combined in the receiver, thereby
`permitting an effective time diversity gain in the re
`ceiver.
`In the preferred embodiment four M/PATH correla
`tors 312,313, 315, and 317 operate on 128 samples, or 32
`chip intervals so as to accommodate as much as an 8
`microseconds variation in the path delays, any one with
`respect to the others. This also imposes the requirement
`that the acquisition sequence be of no less than 9 micro
`seconds duration, preferably two to four times that
`long.
`Although the transmission of the synchronization
`sequence in only the I channel is employed in the pre
`ferred embodiment, an identical utilization of only the Q
`channel for transmission of the synchronization se
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`10
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`35
`
`40
`
`20
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`4,829,543
`10
`quence may easily be employed by one skilled in the art.
`first means for correlating said first phase of received
`Furthermore, it may be desirable to transmit the acquisi
`message data with said first correlator signal and
`tion sequence at some other angle relative to I and Q,
`said second correlator signal to produce first and
`for example, simultaneously and identically in both I
`second weighted signals, respectively;
`and Q for a 45 shift. Any angle can be accommodated
`second means for correlating said second phase of
`by operating on the multipath profile estimates Cr(t)
`received message data with said first correlator
`and Co(t) when applying them to M/PATH correla
`signal and said second correlator signal to produce
`tOrS.
`third and fourth weighted signals, respectively; and
`The outputs I and Q from the adders 331 and 335 may
`means for adding said first and fourth weighted sig
`subsequently be applied to a data signal recovery circuit
`nals and for subtracting said second weighted sig
`such as the conventional baseband synchronous feed
`nal from said third weighted signal.
`back equalizer described in the aforementioned U.S.
`4. A time division multiple access (TDMA) quadra
`patent application No. 009,973.
`ture phase modulation receiver which receives a mul
`Referring now to FIG. 5, a representative graph of
`tipath-impaired signal from a radio channel, including a
`the outputs Cr(t) and Co(t) is shown on one axis with
`15
`predetermined synchronizing signal portion and a mul
`time on the other axis. The outputs of the synchroniza
`tiphase data signal portion, comprising:
`tion correlators 303 and 305 have signed responses at
`means for orthogonally separating the received mul
`each clock pulse but none of the responses exceed the
`tipath-impaired signal into first and second inter
`established threshold magnitude until a correlation with
`mediate signals;
`the predetermined synchronization sequence a T(t) is
`means for detecting the predetermined synchronizing
`realized. As shown, a correlation is found at time td.
`signal portion from at least one of said first and
`In summary, then, the present invention describes a
`second intermediate signals;
`unique phase coherent method for demodulating a
`means for obtaining first and second radio channel
`QPSK radio signal that has been subject to a multipath
`profiles of the radio channel from said detected
`fading radio channel. In order that the equalization for
`25
`synchronizing signal portion; and
`reception of a radio signal subject to Rayleigh and mul
`means for combining both said obtained radio chan
`tipath fading be adapted for the channel, a training or
`nel profiles with the multiphase data signal portion
`synchronization signal is transmitted as one of the vec
`of both said first and second intermediate signals to
`tors of a quadrature phase modulated signal. The ran
`reconstruct phase coherent demodulated quadra
`dom amplitudes and phases of copies of the modulated
`30
`-ture signals.
`signal added to the signal by channel multipath are
`5. A time division multiple access (TDMA) quadra
`correlated and combined in accordance with a multi
`ture phase modulation receiver in accordance with
`path profile signal developed from the synchronization
`claim 4 wherein the synchronizing signal portion fur
`signal. Therefore, while a particular embodiment of the
`ther comprises a binary phase shift keying (BPSK) sig
`invention has been shown and described, it should be
`nal.
`understood that the invention is not limited thereto
`6. A time division multiple access (TDMA) radio
`since modifications unrelated to the true spirit and
`receiver demodulator which demodulates from a radio
`scope of the invention may be made by those skilled in
`channel a multipath-impaired quadrature phase shift
`the art.
`keying (QPSK) data signal including a predetermined
`It is therefore contemplated to cover the present
`acquisition sequence and a message in a TDMA times
`invention and any and all such modifications by the
`claims of the present invention.
`lot, the radio receiver demodulator comprising:
`means for