VOWllE 11 I NUllBER 2
`Tiie Pr11n·t l'nt11•CJ' fll !krts p' - M1erwbip
`111• 1 111 wl1ll V..._ Siil b* Dkb111111,
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`I 2 e1 .. l11!11he, "-'!
`S. P. Yeo, M. S. 1..-..g P. S. Koo;, T. S. YC!IO, llNl X. D.
`7Jtou
`G•ld Mite . . . , P111Ja1 w. llr M1t1" I
`......... ,, . . . . . . 1---, ,1 ....
`P. Mydi1uld llltd J. °"""'1w6ld
`C1q 1 . . a.Qma,J...._elB111• 1 11U .......
`n.111 om.www-Tlme 01
`' 6t-f6
`t *lal If'
`P. H. IMmu, Y. Sldnrany, """ R. Mlllta
`• , .... ,,. ........ u .......... v . . . .
`« ,,_ ...... n111 ., • ..,.,. .. , .. C..Wl'lc Ml-
`awlllfJ ............ Aatl
`'6-e
`••
`I. SlllM Mini tl1tll S. K. CllOwdJlllry
`................ ., ....... .........,.. Dlal1drlc
`w ......... c .................... Me1mJ.Slrtp Gnt&-
`... 8-73
`Y. L KlllnldttJv
`
`. . . . . I Jbtfl&:.........., 73-75
`
`. . . . . . . . . . .
`
`1
`
`.A. Sl1woM
`Wire Alli El Anl1J ~ 75-71
`I'll.. ac..,. .A. Sllanlillo, C Tlmt, tl1tll .A. Slaiwn1lk
`'Ille I t •• , ............... ., ........ Del CIJlnl
`Cllc hr CJll d n, 71-13
`Y. w..,.,.
`A ........,.. .,_,_........., ....... h• Ml-
`awlrlp .... AMmu u ................ Peed,
`D-14
`J. Olm tlllll K. WOfll
`
`FDRUARYI 1118
`
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`
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`Brw:llW .......... .,_.,
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`Opdml 'nrl•rn••• _. ...... 'rd Ip• 11r a
`SW....,_ Netwd, ,._,,,
`H. Pina L. Lb.I, """F. w-.
`
`.... ,._,.
`s.r. .,,,, 11:maet1w ., Ta <Ari u.... a111 c
`K. F. n.. G. B. MOflllll, Mil P. C L. yP
`......... ., ........... It w.... .. •• • •*
`,_., v ...... s...... ,,_112
`E. o. Kolnllw ut.li
`
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`Stodsst'le 1 htr c• 8)11
`s, 117-111
`G. ZWng L. 7illn& lllld Z. Chm
`
`A WIL!.Y·INTUSC1£NCE PUBLICATION
`John Wiley & Sons, Inc.
`NEW YOlllt I OllCHESTER BIUSMNE RlRONlU SINGAPORE
`
`Nall.11(2) 55-112 09")
`
`ZTE v. Fractus
`IPR2018-01461
`
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`
`

`

`.11UllM~,l MICROWAVE
`,,,,M"U''' AND Oi> I ICAL
`TECHNOLOGY
`LEI IERS
`
`VOLUME 11 I NUMBER 2
`
`FEBRUARY 5 1996
`
`EDITOR
`Kai Chang
`Texas A&M Univers ty
`College Station
`Texas
`
`EDITORIAL BOARD
`
`K. K. Agarwal, E·System s, USA
`J. Archer, CSIRO, Australia
`I. J . Bahl, ITT, USA
`P. Bernardi, University of Rome, Italy
`K. B. Bhasln, NASA Lewis Research Center, USA
`K. J. Button, MIT National Magnet Labs. USA
`H. J. Caultleld, Alabama A & M Universily, USA
`J. Chrostowski, National Research Council,
`Canada
`R. A. Cryan, University of Northumbria, UK
`A. A. de Salles, CETUC·PUC, Brazil
`U. Efron, Hughes Research Labs. USA
`M. Ettenberg, David Sarnoff Research Center,
`USA
`H. A. Fetterman, UCLA. USA
`L. Figueroa, Boeing Co .. USA
`T. K. Flndakly, Hoechst Celanese Corp .. USA
`T. T. Fong, Hughes Aircraft Co., USA
`N. N. Fomln, Moscow Technical University, Aus·
`sla
`V. Fouad Hanna, CNET PAB I STS. France
`P. B. Gallion, ENST, France
`F. Gard lol, Ecole Polytechnique Federal, Switzer·
`land
`H. Ghafourl·Shlraz, University of Birmingham,
`England
`J. Goel, TRW. USA
`P. F. Gold1mlth, Cornell University, USA
`K. C. Gupta, University of Colorado, USA
`G. I. Haddad, University of Michigan, USA
`A. C. HanHn, Consultant. USA
`A. Hardy, Tel Aviv University, Israel
`P. R. Harczfeld, Drexel University, USA
`W. J. A. Hoefer, University of Victoria, Canada
`M. Horno, University of Sevilla, Spain
`
`H. C. Huang, Shanghai Science Technology Uni·
`varsity. China
`C. Jackson, TRW, USA
`R. Jansen, Industrial Microwave and RF Tech
`nlques Inc .. Germany
`S. Kawakami, Tohoku University, Japan
`M. A. Karim, University of Dayton, USA
`E. L. Kollberg, Chalmers University of Technol-
`ogy, Sweden
`J. A. Kong, MIT. USA
`Y. Konishi , Uniden Corporation, Japan
`S. K. Koul, Indian Institute of Technology. India
`H. J. Kuno, Hughes Aircraft Co .. USA
`C. H. Lee, University of Maryland, USA
`J. N. Lee, Naval Research Labs, USA
`R. Q. Lee, NASA Lewis Research Center, USA
`S . W. lee, University of Illinois, USA
`T. LI, Bell Telephone Labs. USA
`C. Lin, Bell Communication Research. USA
`J. C. Lin, University of llhnois, USA
`W. Lin, Chengdu Institute of Radio Engineering,
`China
`H. ling, University of Texas, USA
`I. V. Lindell, Helsinki University of Technology.
`Helsinki, Finland
`Y. T. lo, University of Illinois. USA
`J. M. McMahon, Naval Research Labs, USA
`K. A. Michalak!, Texas A & M University, USA
`T. Mldford, Hughes Aircraft Co., USA
`J. W. Mink, North Carolina State Universily, USA
`Y. Naito, Tokyo Institute of Technology. Japan
`R. Nevels, Texas A & M University, USA
`A. I. Noslch, National Academy Science. Ukraine
`
`J. Ojeda-Castaneda, lnsbtuto National de As
`trofis1ca, Mexico
`K. Peterman, Technical Univers1ly. Berlin, Ger
`many
`J. Ra, KAIST, Korea
`
`A. Rosen, David Sarnoff Research Center. USA
`
`G. Salmer, Universit6 des Sciences et Tech
`niques de Ulle-Flanders·Artois. France
`
`F. K. Schwering, US Army CECOM. USA
`
`A. K. Sharma, TRW, USA
`L. C. Shen, University of Houston, USA
`o. W. Smith, British Telecom Research Labs.
`England
`
`B. E. Spielman, Washington University In St.
`Louis. USA
`C. Sun, Cahforn1a Polytechnic Slate Un1vers1ty,
`USA
`
`H.F. Taylor, Texas A& M University, USA
`
`C. S. Tsai, University of California at Irvine. USA
`
`H. Q . Tserng, Texas Instruments. USA
`
`J. B. Y. Tsui, Wright-Patterson AFB. USA
`o. Wada, Fuptsu Labs, Japan
`R. W. Wang, Academia Sinlca, China
`
`A. G. Wllllemson, University of Auckland. New
`Zealand
`
`J. C. Wiltse, Georgia Technology Research lnsh
`lute. USA
`J. Wu, National Taiwan University, Taiwan
`
`E. Yamashita, University of Electro·Commun1ca-
`hons. Japan
`S. K. Yao, Optech, USA
`
`H. W. Yen, Hughes Research Labs, USA
`
`F. T. S. Yu, Pennsylvania State University, USA
`
`Microwave and Optrcat Technology Letters (ISSN: 0895·2477) 1s published
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`Copyright o 1996 John Wiley & Sons, Inc. All rights reserved . No part of
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`original short papers and letters on theoretical, applied, and syslem results
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`in the following areas:
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`Antennas and Propagation
`Texas 77843-3128. Telephone (409) 845-5425.
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`

`

`THE RESONANT FREQUENCY OF
`RECTANGULAR MICROSTRIP ANTENNA
`ELEMENTS WITH VARIOUS SUBSTRATE
`THICKNESSES
`
`KEY TERl\IS
`
`Mehmet Kara
`Weapons Systems 01V1s1on
`Aeronautical & Mant1me Research Laboratory
`Defence Science and Technology Organisation
`UNIVERSITY OF
`P O Box 1500
`MICHIGAN
`sa11sbury SA 5108. Austraha
`JAN 2 6 1Q
`Re1<1111111t freq11e11(1'. rnriom s11bstrt11e. patch a111e1111a. micrM~P
`ENGINEERING
`AUSTRAC r
`For11111/as based 011 tmn\111i.~sio11-li11e. rnl'ity. a11d~1~~~Xll'all models
`to determine the l'<'.\'OJ1a111 Ji·eq11e11cies of a 1-ecta11g1dar microstrip a11te1111a
`t'lrmel// lw1•t• been .1·111died and lhrir rnlidity as.l'essed. Their rnriations
`11·en' c.rperi111c11111/(r 1·erijit'd by ct11a(l':i11g 11 ~·ct ofnell'/y designed a111e111111
`d e111e111s wi1h s11/J.11mtes s11tisji·i11g 1he cri1eri11 Ji ~ 0.0815>.." for 1.21 ~
`r,. .!> 10.2, when' >..11 is 1/ie .fi·ee-spac" 1ml'ele11gth. h the 1hick11ess, and s,
`1/ie relatire per111illil'i1y of th<' dielecflic s11bs1ra11: . .£> 1996 Jolin J.l'ilt•y &
`So11s. Inc.
`
`1. INTRODUCTION
`The resonant frequency of microstrip antcnna clements must
`be determined accurately, as they have narrow bandwidth
`a nd can only operate effectively in the viciniry of the reso(cid:173)
`nam frequency.
`Factors for determining the frequency at which resonance
`occurs include
`
`(a) The voltage standing-wave ratio (VSWR). referred to
`the input terminals of the antc:nna. is at a minimum.
`This corresponds lo a minimum in the magnitude of
`the rcneetion coefficient.
`(b) The input impedance. referred to the input terminals,
`is real (Z;,, = R;11 ) . which means the input impedance
`has no reactive part. Generally. this point is very close
`to the freque ncy where the resistance reaches a maxi(cid:173)
`mum. T herefore the resonant frequency may also be
`defined as the point at which the resistance reaches a
`maximum, inde pe nde nt of the value of reactancc.
`
`This a rticle is primarily conce rned with antenna elements
`that arc matched to their transmissio n-line feeds. In this case,
`the frequency at which the input impedance is real is equal to
`the frequency at which the VSWR is at a minimum.
`Several methods have been proposed and used to deter(cid:173)
`mine performance properties of microstrip antenna clements
`[ 1- 17]. T hese methods have different levels of complexity,
`require vastly different computational efforts, and can gener(cid:173)
`ally be divided into two groups: simple analytical methods
`and rigorous numerical methods. Simple analytical methods
`can give a good intuitive explanation of antenna radiation
`in
`rigorous
`properties. Exact mathematical formulatio ns
`methods involve extensive numerical procedures, resulting in
`round-off e rro rs, and may also need final experimental ad(cid:173)
`justmenrs to the theoretical results. They are also time con(cid:173)
`suming a nd no t easily included in a computer-aided-design
`system. Basically, the re is no clear-cut rule as to which one of
`these is the bes t to use; the firs t guideline would be the
`thickness of the substrate.
`
`Based on this observation, simplified analytical me thods
`are used in th is work. Formulas based on transmission-line,
`cavity. and magnetic-wall models to determine the resonant
`frequencies of a rectangular microstrip antenna element have
`been investigated. Their respective regions of validity in the(cid:173)
`ory and applicability for a given antenna element have also
`been established. For the above-specified range of substrates.
`a transmission line model ha-; been verified and successfully
`u ed to calculate the resonant frequencies of rectangular
`microstrip antenna clements without involving complicated.
`time-consuming. and difficult numerical methods. These re(cid:173)
`sults arc then compared with specified design frequencies.
`
`2. ANALYSIS
`The config uration of a probe-fed rectangular microstrip an(cid:173)
`tenna cle ment i~ shown in Figure 1. The transmission-line
`(1 - 4]. the cavity (5- 10]. and the magnetic-wall models [I I]
`have bcen used f'o r calcu lating the resonant frequencies. and
`the re have been varia tions compared with measurements.
`The basic formulas for computing the resonant frequen(cid:173)
`cies arc given in the fo llowing sections.
`
`2. 1. For11111fas Based 011 the Transmission-Line Model. To cal(cid:173)
`culate the resonant frequency of a rectangular microstrip
`antenna dcment. the a ntenna is regarded as two parallel
`radiating slots [3] with dimcnsions IV and h having constant
`field aperture distributions and separated by the clement
`length /. of approximately half the wavelength in the dielec(cid:173)
`tric.
`In reality the electric field at the open end of the patch is
`distorted by an abrupt termination at the edges, resulting in
`fringing electric field!>. This fringing effect is incorporated
`into the formula for the resonant frequenC)'.
`A formula for the resonant frequencies of rectangular
`antenna clements with thin substrates was given by Bahl (2)
`and Derncryd (4] as
`
`•
`Cu
`./, = 2(L + 2 .. HV )yse(W) .
`
`( I )
`
`is the velocity of electromagnetic waves in free
`whe re t 0
`space and ..:,.(W) is the effective dielectric constant, which is
`o btaim:d from (18]
`
`f:, + 1
`I
`s, -
`e (W) - - - + ~====
`?_./l + 10/t/ W
`2
`•'
`
`(2)
`
`l •
`
`-.!
`--,h
`
`GROUND PLANE
`
`/
`
`/ ·--(cid:173)
`
`DIELECTRIC ~w
`SUBSTRATE
`
`Figure 1 Configura tion of a rectangular microscrip antenna ele(cid:173)
`ment with dimensional parnmetcrs
`
`MICROWAVE AND OPTICAL TECHNOLOGY LETIERS I Vol. 11 , No 2. February 5 1996
`
`55
`
`ZTE v. Fractus
`IPR2018-01461
`
`ZTE
`Exhibit 1004.0003
`
`

`

`TABLE 2 The 0.9-GHz Cavity Resonator
`
`Dimensions
`
`Measured
`FDTD
`
`a (cm)
`
`21.9075
`22.0000
`
`b(cm)
`
`8.89
`9.00
`
`c (cm)
`
`22.065
`22.000
`
`Type
`
`Derivation
`
`Q
`
`ReMJnant Frequency
`
`Empty Cavity
`Empty Cavity
`Tunable Cavity
`Tunable Cavity
`
`Analytical
`Numerical
`Measured
`Numerical
`
`23,171
`23.174
`N/ A
`20.229
`
`963.58 MHz
`967.50 MHz<error < 0.5<"()
`N/A
`900.20 MHl
`
`TABLE 3 The 1.4-GHz Cavity Resonator
`
`Dimensions
`
`Measured
`FDTD
`
`a (cm)
`
`13.081
`13.000
`
`b(cm)
`
`8.859
`9.000
`
`c (cm)
`
`15. I B
`15.250
`
`Type
`
`Derivation
`
`Q
`
`Rc~ommt Frequency
`
`Empty Cavity
`Empty Cavity
`Tunable Cavity
`Tunable Cavity
`
`Analytical
`umerical
`Measured
`umerical
`
`22.982
`22.992
`N/ A
`20,698
`
`l.51520Hz.
`1.5185 GHz (error < O.Y;)
`1.481 7 G Hz
`1.-164 GHz (error < I .2"( )
`
`REI' ERENCES
`I. K. A. Zaki and C. Chen. ··Loss Mechanism in Dielectric-Loaded
`Resonators:· I F.££ Trans. Micrvwoce 771e01y Tech . Vol. MIT-33,
`Dec. 1985. pp. 1-148- 1452.
`2. J. Krupk<1, "Pro perties of Shielded Cylindrical Quasi-TE11,,m·mode
`Dielectric Resonators:· IEEE Tram . .11icrou·ace 171eory Tech. Vol.
`Mrr-36. April 1988. pp. 774-779.
`3. J. E. Lebaric and D. Kajfez. "Analy~i~ of Dielectric Resonawr
`Cavities using the Finite Integration Technique," /£££ Tm11s.
`Microwace Theo1~· Tech .. Vol. MIT-37, Nov. 1989. pp. I 740 17-18.
`4. A. Navarro, M. J. Nunez. and E. Martin. ··Finite Difference Time
`Domain FFf Method Applied to Axial ly Symmetrica l Electromag(cid:173)
`ne tic Resonant Devices . ., IEE Proc. Pt. fl. Vol. L"\7. No. 3. June
`1990, pp. 193 196.
`5. C. Wang. 13. O. Gao, and C D. Deng. "Accurate ~tudy of
`Q-Factor of Resonator by a Finite-Difference T ime-Domain
`Method."" IEEE Tmm . . \ficro1mre 771eot)" Tech .. Vol. l\IIT--13. July
`I 995. pp. 152-1- 1529.
`6. R. F. 1 larringwn. Time-/-111r111011ic Efec1ro111ag11etic Fic:ld5,
`McGraw- Hill. Inc .. New York. 1961.
`
`Receired 8-f8-95
`
`Microwa,·e and Optical Technology Letters. 11 / 2. M 66
`c 1996 John Wiley & Sons. Inc.
`CCC 0895-2477 / 96
`
`To compute the resonant frequencies, Lhe FDTD mcLhod
`was used. The cavities were excited by a sine wave wiLh a
`Gaussian envelope. Using the Fourier transform, the timc(cid:173)
`domain data were transformed to the frequency domain. The
`resonant frequencies were discerned as distinctive peaks that
`corresponded to the various modes present in the cavity.
`Once the resonant frequencies were found, the cavity was
`excited with a continuous sinusoidal wave al that resonant
`frequency. and the method previously described was used to
`obtain the Q factor.
`The Q factors and resonant frequencies computed with
`the FDTD algorithm arc shown in Tables 1 and 3 for the 0.9
`and 1.4-GHz resonators. respectively. For both empty cavi(cid:173)
`ties, the numerical results are compared with analytical val(cid:173)
`ues. and the agreement is less than 1 %1 for both the Q and
`resonant frequ ency. For the tunable cavity at 0.9 GHz, the
`numerical values arc shown in Table 2, but no analytical or
`measured results were available. For the tunable cavity at 1.4
`G Hz, measurements of the resonant frequency were avail(cid:173)
`able, and the FDTD value is within 1.20: error of the
`measurements. The measured resonant frequency was ob(cid:173)
`tained from the scattering parameters measured with a net(cid:173)
`work analyzer. The numerical results for the Q of the tunable
`cavities in both Tables 2 and 3 show the right trends. because
`the Q decreases from that of the empty cavity. as would be
`expected due to the additional loss of the tuning disk and
`truss.
`
`IV. CONCLUSIONS
`A technique for computing the quality factor of complex
`resonators was demonstrated. The advantages of this ap(cid:173)
`proach are that the FDTD method can be employed to model
`resonators of arbitrary shape that contain dielectric and
`metallic materials such as tuning disks. and the results arc
`obtained directly from the time-domain fields through time
`averaging. A similar approach can be used to account for
`dielectric losses.
`
`EXPERIMENTAL INVESTIGATIONS ON
`THE IMPEDANCE AND RADIATION
`PROPERTIES OF A THREE-ELEMENT
`CONCENTRIC MICROSTRIP
`SQUARE-RING ANTENNA
`
`I. Saha Misra and S. K. Chowdhury
`Department of Electronics & Telecommunication Engineering
`Jadavpur U111vers11y
`Calcutta 700 032 India
`
`KEY TER,\IS
`Co11ce111ric micro.wrip square ring a11tc1111a. impedance 1111d 111diatio11
`ba11dll'id1'1. fe<'d lorn1io11
`
`ABSTR<\CT
`£\peri111e111af i11res1ig/l/io11 011 an electro111ag11e1ical(1· co11pfed co11ce111ric
`111icros1rip Sf11111re-ri11g a111e111w (C\ISRA J has been presemed. (}11r preri(cid:173)
`ous work 011 co11ce111ric 111icros1rip a111111/11r-ri11J~ r<'sonmors receafed 1ha1
`rile i111peda11t'e and mdia1io11 bandll'idrh c1111 be improred co11sidemh~1'.
`A three-efe111e111 C/llS!?A has bee11 desig11ed and its meas11re1f impedance
`and mdiaiio11 pallrms hare been comparl'd with those vf 11 single
`sq11are-ri11g a111e1111t1 haring a di111e11sio11 equal /0 1he largest i:/e111e11t of
`1fre CMSl?A. Tfre 1'f](·c1 of change of feed foca1io11 has also been s111died.
`Results show 1/1111 1fre wtal perce111 ba11d11'id1/i (B W) for rile w11ce1111ic
`111icros1rip square ring <1111e111w is larger wi1/i respec1 10 1/ie si11 .. ~fe-ri11g
`a111e111111, 1111d 1Jii5 effecr is cery much pro111i11e111 ll1 a partirnf11r feed
`focarion. s. 1996 John Wife:i· & So11s. l11c.
`
`INTRODUCTION
`One of the most important aspects of any antenna is its
`bandwidth. The main drawback of microstrip antennas (MA)
`is their narrow bandwidth. The bandwidth of an antenna can
`be increased proportionately either by increasing the sub(cid:173)
`strate thickness or by reducing the dielectric constant [I).
`However. the increase in substrate thickness is generally
`limited by excitation of surface waves, and there arc practical
`
`66
`
`MICROWAVE AND OPTICAL TECHNOLOGY LETTERS I Vol. 11 , No. 2. February 5 1996
`
`ZTE v. Fractus
`IPR2018-01461
`
`ZTE
`Exhibit 1004.0004
`
`

`

`limitations in decreasing the value of the dielectric constant.
`Thus, the BW obtained from conventional MA is not suffi(cid:173)
`cient for many purposes. Another important aspect of MA is
`a variety of feeding techniques that can be applied to them.
`Electromagnetic coupling is an attractive one. due to its
`multilayered structure. which allows the antenna to be inte(cid:173)
`grated with its feed circuitry [2]. Another rea<>on for electro(cid:173)
`magnetic coupling is that it has been used in configurations
`that significantly enhance the BW of a patch antenna [3].
`Our previous works on electromagnetically coupled con(cid:173)
`centric microstrip ring antennas [4. 5] show that these struc(cid:173)
`tures give wide bandwidlh. The present article deals with a
`concentric microstrip square-ring antenna (CvfSRA) contain(cid:173)
`ing three clements. The variation of input impedance at
`differe nt bands or frequencies have been measured by an HP
`
`841 OB network analyzer and compared with those of an
`electromagnetically coupled microstrip single square-ring an(cid:173)
`tenna. The variation of input impedance has also been mca-
`
`f•equency "•P 0 01 Griz
`- • - Cenle• feed
`-·-•-·- o-45 ctn owoy ftotn cen•ef
`- -o ... - Corner feed
`
`' r"'centerfeed'
`I
`I
`~ : 1
`0 ·45Cm
`
`' <Corner .ee
`'
`'
`d
`'
`
`Figure 1 F.lcc1romagnctically red concentric square-ring. amcnna
`-.ho\\ ing diffcrcnl recd pol.it ion~
`
`Figure 3
`loca1io11\
`
`Impedance plot of \ingle-ring. amenna at differem feed
`
`(g]: .T
`
`0.
`
`.1
`
`, ,
`
`I
`
`I
`
`I
`
`-1 1- I -lWof(cid:173)
`t-d n -1
`(a) w
`
`'
`
`I
`
`(b)
`
`Figure 2
`
`(a) Thn:e-element concentric micros1rip :.qu:m:-ring an1cnna (b) single square-ring antenna
`
`TABLE 1 Comparison of Percent BW Between Single Square-Ring and Concentric Square-Ring Antennas
`
`Single Square Ring
`
`Concentric Square Ring
`
`Feed loca1ion
`
`Ccmer feed
`
`0.45 cm away
`rrom center
`
`Corner feed
`
`Frequency range
`in GHz
`
`2.7 - 2.74 = 0.11-l
`
`2.698 - 2.76 = 0.068
`
`2.638 - 2.692 = 0.054
`
`2.805 - 2.852 = 0.047
`
`'r Bandwidth
`
`IA7
`
`2.49
`
`2.02
`
`1.66
`
`Frequency range
`in Gllz.
`2.655 - 2.685 - o.m
`5.3 1 - 5.435 = 0.125
`6.55 - 6.85 - 0.30
`2.65-1 - 2.694 = 0.0-1
`7.5 '"' 0.86
`6.M
`
`2.612 - 2.65 = !l.038
`2.74 - 2.773 = o.o:n
`6.74 - 7.2 = 0.46
`7.33 - 7.53 - U.20
`
`er Bandwichh
`
`l.12
`2.32
`.J.47
`
`1.49
`12.16
`
`1.44
`1.19
`6.6
`2.69
`
`MICROWAVE AND OPTICAL TECHNOLOGY LETIERS I Vol. 11, No. 2, February51996
`
`67
`
`ZTE v. Fractus
`IPR2018-01461
`
`ZTE
`Exhibit 1004.0005
`
`

`

`sured at three different feed locations (Figure I). These
`measurements indicate that the I : 2 VSWR bandwidth ob(cid:173)
`tainable by this structure is about 14%.
`
`taining the following relation:
`
`w,, f- I
`d ,, f- 1
`; = -
`- = - - = 1.15.
`W,,
`d,,
`
`as indicated in Figure 2(a).
`
`DESIGN OF THREE-ELEMENT CMSRA
`The three-element CMSRA is shown in Figure 2(a}. We have
`first chosen the innermost square-ring antenna with side
`ti = 1.0 cm and width w = 0.2 cm. T he spacing between the
`adjacent clements and their widths are then chosen. main-
`
`f ,eQ vency step 0 0 I GH t
`~- Ce:uer 'eed
`-
`- ·-• ~·- 0 ~s cm owoy fro.,.., ic:e111e·
`- o - - Corti er feed
`-
`
`. ¢1
`
`,0
`
`- · · - ·- 0 4S cm owoy from cenrer
`
`la)
`
`(b)
`
`where the suffix n re presents the n th patch number.
`The ring widths and spacings increase from the innermost
`element to the outermost element. Maintaining this relation.
`the outermost square ring has side a = 2.42 cm and wid th
`w = 0.267 cm. The single-element square-ring antenna inves(cid:173)
`tigated has an identical dimension [Figure 2(b)). The three(cid:173)
`clement CMSRA and single-element square-ring have been
`fabricated on PTFE s ubstrate having dielectric constant e, ~
`2.55 and thickness 0.159 cm. Both of them are fed electro(cid:173)
`magnetically by a so-n microstrip line fabricated on a sub(cid:173)
`strate with identical dielectric constant and other properties.
`
`EXPERIMENTAL RESULTS
`The measured input impedance loci for the single square ring
`and concentric ring have been plotted in ' Figures 3 and 4,
`respectively. The I : 2 VSWR circle has also been drawn on
`the corresponding Figures. T he compared I: 2 VSWR BW
`for the single square ring and concentric ring at different
`feed locations is given in T able I. From this table, it is seen
`that the total I: 2 YSWR BW is increased for the thrce(cid:173)
`demcnt CMSRA a!> compared to that or the single ring. This
`effect is prominent at the feed location OAS cm away from
`the center [Figure 4(b)).
`
`RADIATION PATTERN
`The radiation patterns for the single square-ring and thrce(cid:173)
`clement CMSRA have been measured for diff..:rent feed
`locations and are plotted in Figure 5 and 6. respectively,
`covering the entire bandwidth of the antenna. Comparison of
`the radiation p<ittern of a single ring and CMSRA (Figures 5
`and 6) show~ that over the entire bandwidth the nature of the
`radiation pattern is qualitatively similar to that of the single
`ring operating at the fundamental mode. Again from Figure 6
`it is seen that radiation patterns for CMSRA remain un(cid:173)
`changed with the change of feed location.
`
`CONCLUSION
`From the above in\'estigations it may be said that the concen(cid:173)
`tric microstrip square-ring antenna has a multiple band e ffect
`with increase in total percent bandwidth with respect to the
`single square ring having the largest physical dimension of
`the CMSRA. Changing the location of the feed the total
`percent b<rndwidth
`increases considerably, and with
`the
`
`Corne• feed - ·- 2·65 GHt
`O·a5 cm owoy from center ---- 2. 7 GHz
`o
`-\0°
`
`C e1~1:d - - 2. 7 GHz
`
`Impedance plot of concentric square-ring antenna (a) for
`Figure 4
`lower band of frequencies. (b) for higher band of frequencies for the
`feed 0.45 cm away from center
`
`Figure 5 E-planc radiation pattern of single square-ring antenna at
`different feed locations
`
`68
`
`MICROWAVE AND OPTICAL TECHNOLOGY LETIERS I Vol. 11 , No 2. February 5 1996
`
`ZTE v. Fractus
`IPR2018-01461
`
`ZTE
`Exhibit 1004.0006
`
`

`

`- •- 2·65GHz
`---- 6 .75 GHz
`
`2-65 GHz
`-•-
`6-6 GHz
`-
`-
`- - - - 6 · 9 GH z
`- ·- 7.3 GHz
`
`-•- 2.62 GHz
`--6.8~GHz
`- --- 7.12 GHz
`- - 7.4 GHz
`
`0
`
`(a)
`
`(h)
`
`(c)
`
`Figure 6 E-planc radiation pattern of concentric square ring an(cid:173)
`tenna (a) for center fe..:d (b) for the feed 0.45 cm away rrom center
`and (c) for corne r feed
`
`changed feed location radiation patterns of the CMSRA
`remain unaltered. The most attractive feature of this struc(cid:173)
`ture is the increase of impedance and radiation bandwidth
`without losing the microstrip antenna's advantage of small
`size.
`
`ACKNOWLEDGMENT
`T he authors would like to thank CSIR. Government of India
`for financial assistance for this research.
`
`REFERE:\1CES
`I. G. Kumar and K. C. Gupta . .. Broad-Band Microstrip Antennas
`Using Additional Resonators Gap-Coupled to the Radiating
`Edges," !£££ Trans. A11te1111as Propagat .. Vol. AP-32, Dec. 1984.
`pp. 1375 1379.
`2. B. Belentcpc ... Modelling and Design of Electromagnetically Cou(cid:173)
`pled Microstrip Patch Antennas and Antenna Arrays,.. IEEE
`Antenna Propagat. Magazine, Vol. 37, Feb. 1995, pp. 31- 38.
`3. D. M. Pozar and S. M. Yoda. " Rigorous Analysis of a Microstrip
`
`Feed Patch A ntennas:· IEEE Trans. Amen11as Propagat., Vol.
`AP-35, Dec. 1987. pp. 1343- 1350.
`4. I. Saha Mi~ra and S. K. Chowdhury, "Experiments on the
`Impedance and Radiation Properties of Concentric Microstrip
`Ring Resonator," £lec1ro11 Leu., Vol. 16, March 1995, pp. 421 -422.
`5. I. Saha Mi:.ra and S. K. Chowdhury ... Design of Concentric
`Microstrip Ring Antennas:· IEEE A111e1111a Propagm. Maga::ine. to
`be published.
`
`Receired 9-15-95
`
`Microw;l\'c and Optical Technology Leuers. 11/ 2, 66-69
`-e 1996 John Wiley & Son~. lac.
`CCC 0895-'.!477 /96
`
`RADIATION BEHAVIOR OF PLANAR
`DOUBLE-LA YER DIELECTRIC
`WAVEGUIDES COMBINED WITH A
`FINITE METAL-STRIP GRATING
`
`Victor I. Kallnlchev
`Radio Engineering Department
`Moscow Power Engineering Institute (Technical University)
`14 Krasnokazarmennaya, Moscow 111250. Russia
`
`KEY TER:\IS
`,\fr111l-11np grt11111.i:. do11Me-l11yer dielectric 1rnreguide. )wface mode. leaky
`mode. mdit11ion
`
`ABSTRACT
`Tire radiation and srauering dwracteristics of do11ble-layer planar dielec(cid:173)
`tric ll'at'<'g11ides integrated ll"itlr 11 metal-strip grating are ana~r::ed using a
`111·0-dim1•nsio11al model l/1111 t1 11wro11·-strip cwre111 approximation. Some
`nw11c11rnl tlat11 concemin8 the high(1· directio1111/ ante1111a t1pplications of
`these V/111('//llVS in the 111illi111cter-11·an• range are presented. 'l;: 1996 John
`Will')·<~ Suns. Inc.
`
`1 . INTRODUCTION
`Research performed during the last several years showed that
`radiating structures based on a dielectric waveguide inte(cid:173)
`grated with a metal-strip grating are very promising as low(cid:173)
`cost. low-profil e. and easy-to-fabricate microwave and mil(cid:173)
`limeter-wave printed antennas [I. 2). The most commonly
`used geometry of such antennas is based on a single-layer
`grounded dielectric slab or a dielectric-inset waveguide [3- 5).
`ln this work. the subject of investigation is a radiating
`structure formed by a double-layer planar dielectric wavc(cid:173)
`guitk combined with a finite metal-strip grating [Figures J(a)
`and l(b)]. The lower layer may be either an auxiliary dielec(cid:173)
`tric substrate(&~> I) or an air gap (e~ = 1). It is assumed
`here that the upper layer guides a slow surface wave: that is.
`we suggest that &1 > e~.
`The goal of the work is to develop an effective approach
`to the analysis of such double-layer structures and to obtain
`some data concerning their radiation and scattering charac(cid:173)
`teristics in millimeter-wave range. In particular, it would be
`interesting to compare the radiation behavior depending upon
`whether the grating is located on the bottom or top surface of
`the upper guidcway layer. Moreover, the re is a further aspect
`of in terest to us. As was established by Jackson and Oliner,
`the higher-order below-cutoff modes (leaky modes) existing
`
`MICROWAVE AND OPTICAL TECHNOLOGY LETIERS I Vol. 11. No. 2. February 5 1996
`
`69
`
`ZTE v. Fractus
`IPR2018-01461
`
`ZTE
`Exhibit 1004.0007
`
`