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`PCT/US2005/026766
`
`four preamble carrier-sets is described here. For those
`
`skilled in the art, different values for the FFT size, the
`
`left and right guard band sizes, or the number of preamble
`
`carrier-sets may be used.
`
`5
`
`[0048]
`
`In the case of four-sector configuration in which
`
`each cell contains four sectors, one way to generate
`
`preambles is to divide the entire 1024 subcarriers into four
`
`equal subset, arranged in an interlaced manner.
`
`Effectively, there are four preamble carrier-sets. The
`
`10
`
`subcarriers are modulated, for example, using a level
`
`boosted Phase Shift Keying (PSK) modulation with a CAZAC
`
`sequence cyclically shifted with a code phase defined by
`
`IDcell and Segment, which are the base station identity.
`
`More specifically, the four preamble carrier-sets are
`
`15
`
`defined using the following formula:
`
`PreambleCarrierSetm
`
`m+4*k
`
`( 18)
`
`20
`
`where PreambleCarrierSetm specifies all subcarriers
`allocated to the specific preamble, m is the number of the
`preamble carrier-set indexed as 0, 1, 2, or 3, and k is a
`
`running index. Each segment of a cell is assigned one of
`
`the four possible preamble carrier-sets in this particular
`
`example.
`[0049] To further illustrate, let the 1024-FFT OFDMA sampling
`
`25
`
`rate be 20 MHz at the Nyquist rate. The basic preamble time-
`domain syntbol rate is lOMHz. The frequency=domain components
`are composed of a Chu sequence described in Equations (1) and
`
`(2) of length 128 that is zero-inserted to length 512 by
`
`30
`
`inserting CAZAC symbols one for every four frequency bins.
`
`In
`
`the following, it can be established that a time-domain CAZAC
`sequence at the symbol rate (lOMHz)
`introduces a CAZAC
`sequence in frequency domain after spectrum folding.
`Its
`frequency-domain CAZAC sequence can be computed using a 512-
`
`35
`
`FFT operation instead of a 1024-FET operation.
`
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`sequences do not maintain the CAZAC property. For example,
`
`the PAPR is about 4.6 dB when the phase rotation shown in
`
`FIG. 1B is 8 = n/3. To achieve lower PAPR, the phase 8 can
`
`be adjusted to n/4. Although the "folded spectrum" is no
`
`5
`
`longer an exact CAZAC sequence in the frequency domain, the
`
`resulting time domain waveform has a low PAPR of 3.0dB.
`[0055] This technique to preserve CAZAC sequence
`
`characteristics of the folded frequency spectrum in both
`
`frequency and time domains is now further described below.
`[0056] Following on the above example, the above described
`
`10
`
`construction of the CAZAC sequence in FIGS.
`
`lA and lB is
`
`used to reconstruct the 1024 subcarriers using the 4:1 zero(cid:173)
`
`inserted 512-element frequency-domain CAZAC sequence of a
`
`128-element Chu sequence such that, after the spectrum
`
`15
`
`folding due to the down sampling at the mobile station
`
`receiver, the folded 512 spectral components form the
`
`frequency-domain CAZAC sequence of the Chu sequence.
`
`[0057] Let c~u denote the time-domain 512-element CAZAC
`
`sequence and its frequency-domain CAZAC sequence be denoted
`
`20
`
`as gchu (512 elements) and expressed as
`
`.1rn2
`
`gchu(4n+k)= e'rn, n=0,1, ... ,127'
`othenvise
`0,
`
`{
`
`(26)
`
`where k denotes the fixed preamble carrier-set.
`
`cchu and gchu
`
`form a time-frequency pair and their relationship is
`
`2 5
`
`expressed as
`
`( 2 7)
`
`[0058j
`
`In IEEE P802.16e/D3, the 1024-FFT OFDMA has 86 guard
`
`30
`
`subcarriers on the left-hand side and 87 on the right-hand
`
`side. The DC (direct current) subcarrier resides on index
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`512. The construction procedures of assembling gL and gR of
`
`the left- and right-hand sides 1024-FFT OFDMA preambles are
`
`g R (1 : 86) = g Chu (1 : 86)
`
`5
`
`gR(87: 425) = e-j;r
`
`13gchu(87: 425)
`
`(28)
`
`(29)
`
`(30)
`
`(31)
`
`(32)
`
`(33)
`
`_
`
`/'A""'£'".~1""'\.
`
`g R \. '4-L.U : .J 1 L.) = V
`
`1\
`
`gL(1:86)=0
`
`g L (87: 425) = ejtr/J g Chu (87: 425)
`
`gL(426: 512) = gChu(426: 512)
`
`10
`
`In addition, if the DC component is not used, for example in
`IEEE 802.16 OFDMA system, then
`
`The final reconstructed 1024-FFT frequency components of the
`
`preamble symbol is
`
`(34)
`
`15
`
`20
`
`25
`
`30
`
`and its final reconstructed 1024 time-domain preamble
`sequence at Nyquist rate is
`
`c = IFFJ; 024 ( q) .
`
`(35)
`
`(36)
`
`[0059] After spectrum folding due to subsampling at symbol
`rate in the time domain, the resulting folded frequency
`spectral components of even-numbered samples are, based on
`
`Equation (24),
`
`g(1: 512)- gL(l: 512) + gR(l: 512)
`
`(37)
`
`The overlapped area has the following relationship
`
`(38)
`
`17
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`[0060] Equations (28)-(33) suggest that the CAZAC property is
`preserved. Note also that overlapped area of odd-numbered
`samples has the following relationship according to Equation
`(25):
`
`(39)
`
`Therefore, the reconstructed time sequence has the lowest PAPR
`for the even-numbered sampled sequences and very low PAPR for
`the odd-numbered sampled sequences that only slightly deviate
`from the exact CAZAC sequences due to the guard bands
`
`requirement. The nominal PAPR of the time-domain sub-sampled
`sequences is less than 3dB at all different code-phases. The
`frequency components of the reconstructed 1024-FFT in the
`preamble sequence have constant amplitudes and thus may be
`used to facilitate the channel estimation.
`[0061]
`performed as follows: The IDCell and Segment allocation to
`
`In one implementation, fast cell searching can be
`
`different sector are done via assigning different CAZAC code
`
`phases of cyclic shift of the gdu sequence and forming the
`
`5
`
`10
`
`15
`
`20
`
`time-domain sequence in the same manners described in
`Equations (28)-(36).
`[0062] FIG. 4 shows an example of the subcarrier
`allocations of the preamble sequence in segment 0.
`[0063] FIG. 5 shows the corresponding amplitude of the
`25 waveform in the time domain. Because the frequency-domain
`spectral components form a CAZAC sequence, a new sequence
`formed by cyclically shifting the sequence of the spectral
`components, in the time domain (subsampled) also forms a
`CAZAC sequence. Due to the well-defined zero-
`autocorrelation properties, identifying code-phase and
`
`30
`
`thereby identifying IDcell and segments can be made with
`optimal decision. The cyclic shifting of the order of
`different components in the PN sequence permits the MSS to
`retain one copy of the PN sequence without other shifted
`sequences. A simple look-up table may be used to provide
`
`35
`
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`5
`
`10
`
`15
`
`20
`
`25
`
`30
`
`the relationships between all sequences based on the cyclic
`shifting and the corresponding base stations and the
`associated cell sectors. Therefore, the present technique
`enables fast cell searching.
`
`[0064] A CAZAC sequence has been used for channel sounding
`whereby the CIR (channel impulse response) can be uniquely
`determined because of the zero-autocorrelation property of
`the CAZAC sequence. In OFDMA or OFDM systems, we can use it
`not only to identify CIR but also to achieve fine timing
`synchronization whereby we can exclusively remove GI (guard
`interval) so as to minimize ISI.
`
`[0065] FIG. 6 shows the time waveform of the result of
`matched filtering of the near-CAZAC sequence (spaced by
`symbols) without channel distortion and FIG. 7 shows the
`
`result of matched filtering of the near-CAZAC sequence in a
`multipath fading environment. The waveforms are CIRs of the
`tested RF multipath environment.
`[0066] For a sensible and low-cost TCXO, the clock
`
`precision is usually about 5ppm for both the base station
`and the mobile station in some systems. At 10GHz the
`frequency offset becomes 50kHz. For a 11kHz FFT spacing it
`spans 5 subcarriers in both directions.
`[0067] The near-CAZAC sequence in the frequency domain can
`be used to simplify identification of peak positions of the
`cross-correlation. For example, for a sensible and low-cost
`TCXO,
`the clock precision is usually about Sppm (BS+SS) . At
`lOGHz carrler frequency the frequency offset becomes 50kHz.
`For an 11kHz FFT spacing it spans 5 subcarriers in both
`directions. We can assign code phase for different sectors
`that have different IDCells and segments by at least 10 code
`
`phase apart that accommodates ±5 subcarrier drifts due to
`
`large frequency offset, then we can easily perform frequency
`offset cancellation to within 11kHz. Further fine
`correction utilizes pilot channel tracking.
`
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`[0068] The PAPR of the current preamble design is 4.6dB.
`The PAPR can be further reduced by selecting different phase
`factor in Equations (29) and (32). For example, if we change
`
`the phase factor in Equations (29) and (32) from e~ 13 to
`
`5
`
`e~ 14 as shown in Equations (40) and (41), then PAPR is
`reduced to 3.0dB by compromising the CAZAC performance.
`
`(40)
`
`(41)
`
`10
`
`[0069] Only a few implementations are described.
`Modifications, variations and enhancements may be made based
`on what is described and illustrated here.
`
`20
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`What is claimed is:
`
`CLAIMS
`
`1. A method for communications based on OFDM or OFDMA,
`comprising:
`
`5
`
`selecting an initial CAZAC sequence;
`
`modifying the initial CAZAC sequence to generate a
`
`modified sequence which has frequency guard bands; and
`using the modified sequence as part of a preamble of a
`
`downlink signal from a base station to a mobile station.
`
`10
`
`2. The method as in claim 1, wherein the initial CAZAC
`
`sequence is a Chu sequence.
`
`3. The method as in claim 1, wherein the initial CAZAC
`
`15
`
`sequence is a Frank-Zadoff sequence.
`
`4. The method as in claim 1, further comprising:
`
`using an order of frequency components of the preamble
`sequence to identify a base station transmitter; and
`
`20
`
`using different orders of frequency components of the
`
`preamble sequence based on a cyclic shift of the orders of
`
`frequency components to identify different base station
`
`transmitter.
`
`25
`
`5. The method as in claim 4, further comprising using
`different orders of frequency components of the preamble
`
`sequence based on a cyclic shift of the orders of frequency
`
`components to further identify different cells sectors in
`each cell of a base station.
`
`30
`
`6. The method as in claim 1, wherein the modifying of
`the initial CAZAC sequence comprises:
`selecting frequency components in the initial CAZAC
`
`sequence to create the frequency guard bands; and
`
`21
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`PCT/US2005/026766
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`setting amplitudes of the selected frequency components
`
`in the initial CAZAC sequence to zero to create frequency
`
`guard bands.
`
`5
`
`7. The method as in claim 6, wherein the modifying of
`
`the initial CAZAC sequence further comprises:
`
`adjusting a phase of a selected group of adjacent
`
`frequency components in the initial CAZAC sequence whose
`
`amplitudes are not changed.
`
`10
`
`15
`
`8. The method as in claim 1, further comprising:
`
`sub sampling the preamble at a mobile station receiver
`
`to create a frequency overlap and to minimize a variation in
`
`amplitude.
`
`9. A method for communications based on OFDM or OFDMA,
`
`comprising:
`
`selecting a CAZAC sequence of a length L in frequency
`
`which includes spectral components in first, second and
`
`20
`
`third sequential portions in frequency;
`
`modifying the CAZAC sequence to produce a first
`
`modified sequence by setting amplitudes of spectral
`
`components in the first portion of the CAZAC sequence to
`zeros and adding a first phase shift on spectral components
`
`25
`
`of the second portion of the CAZAC sequence, without
`
`changing the third portion;
`
`modifying the CAZAC sequence to produce a second
`
`modified sequence by setting amplitudes of spectral
`
`components in the third portion of the CAZAC sequence to
`
`30
`
`zeros and adding a second phase shift spectral components of
`
`the second portion of the CAZAC sequence, without changing
`
`the first portion;
`combining the first and second modified sequences to
`form a combined sequence in frequency of a length 2L,
`35 wherein the first portion from the first modified sequence
`
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`is positioned next to the third portion from the second
`modified sequence in the combined sequence; and
`performing an inverse fast Fourier transform on the
`combined sequence to generate a first preamble sequence in
`time for OFDM or OFDMA communication.
`
`10. The method as in claim 9, further comprising
`setting widths of the first and third portions of the CAZAC
`sequence to achieve desired OFDMA guard bands.
`
`11. The method as in claim 9, further comprising
`setting an amplitude of a DC subcarrier to zero when the DC
`
`subcarrier is not used.
`
`12. The method as in claim 9, further comprising making
`the first phase shift and second phase shift to be opposite
`to each other.
`
`13. The method as in claim 9, further comprising:
`prior to generation of the first and the second
`modified sequences, performing a cyclic shift of frequency
`components of an initial CAZAC sequence to produce the CAZAC
`sequence which is subsequent used to generate the combined
`sequence; and
`
`using an order of the spectral components of the CAZAC
`sequence to identify at least an identity of a base station
`which transmits the first preamble sequence as part of a
`
`downlink signal.
`
`5
`
`10
`
`15
`
`20
`
`25
`
`30
`
`14. The method as in claim 13, further comprising using
`the cyclic shift of frequency components of the initial
`CAZAC sequence to generate different orders of the frequency
`components in frequency to identify at least different base
`stations and different cell sectors of cells of the
`35 different base stations.
`
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`15. The method as in claim 9, further comprising:
`
`performing a cyclic shift of time components of the
`
`first preamble sequence to generate a second preamble
`
`sequence.
`
`16. The method as in claim 15, further comprising using
`
`the cyclic shift of time components of the initial CAZAC
`
`sequence to generate different orders of the time components
`
`to identify at least different base stations.
`
`5
`
`10
`
`17. The method as in claim 16, further comprising using
`
`the cyclic shift of time components of the initial CAZAC
`
`sequence to generate different orders of the time components
`
`to represent, in addition to the different base stations,
`
`15 different cell sectors of cells of the different base
`
`stations.
`
`18. The method as in claim 9, wherein the initial CAZAC
`
`sequence is a Chu sequence.
`
`20
`
`19. The method as in claim 9, wherein the initial CAZAC
`
`sequence is a Frank-Zadoff sequence.
`
`20. A method for communications based on OFDM or OFDMA,
`
`25
`
`comprising:
`
`sub sampling a preamble signal in a downlink signal
`
`received at a mobile station receiver to create a frequency
`
`overlap and to minimize a variation in amplitude, wherein
`
`the preamble signal is generated from an initial CAZAC
`
`30
`
`sequence to preserve properties of the initial CAZAC
`
`sequence and has frequency guard bands; and
`
`extracting an order of signal components in the
`
`preamble signal to identify at least a base station at which
`
`the downlink signal is generated.
`
`35
`
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`21. The method as in claim 20, wherein the initial
`
`CAZAC sequence is a Chu sequence.
`
`22. The method as in claim 20, wherein the initial
`CAZAC sequence is a Frank-Zadoff sequence.
`
`5
`
`25
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`EVOLVED-0002061
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`wo 2005/011128
`
`PCT/ AU2004/001036
`
`1
`
`METHOD AND SYSTEM FOR COMMUNICATION IN A MUL TJPLE ACCESS
`NETWORK
`
`1 0
`
`RELATED APPLICATIONS
`This application claims priority to Australian Provisional Patent Application
`5 No. 2003903826, filed 24 July 2003, entitled· "An OFD~ Receiver Structure", the
`specification thereof being incorporated herein by reference in its entirety and for
`all purposes.
`FIELD OF INVENTION
`The present invention relates to the field of wireless communications. In
`particular,
`the present
`invention
`relates
`to
`improved multiple access
`communications.
`In one form, the invention relates to an improved signal
`processh1g method and apparatus for a multiple access communication system.
`It will be convenient to hereinafter describe the invention in .relation to the use of
`an iterative method of determining the reception of a signal in a multi user packet
`based wireless OFDM
`(Orthogonal Frequency Division Multiplexing)
`communication system, however, ·it should be appreciated that the present
`invention may not be limited to that use, o.nly. By way of further example, in other
`forms the present invention may relate. to recursive filtering for joint iterative
`decoding in a variety of systems and functions such as linear multiple access
`channel decoders; iterative equalisation. iterative joint channel estimation and
`detection/decoding,
`iterative space-time processing,
`iterative multi user
`interference cancellation and Iterative demodulation.
`RELATED ART
`Throughout this ·specification the use of the word "inventor" in singuiar form
`25 may be taken as reference to one {singular) or more (plural) inventors of the
`present invention. The inventor has identified the following related art.
`Most wireless communications systems are based on so-called multiple
`access
`techniques
`in which,
`information such as voice and data are
`communicated. . This is a technology where ·many simultaneously active users
`share the same system resources in an ·organised manner.
`In most cases,
`sharing resources in a multiple access system means that if more than one user
`is active, then .all active users interfere with each other. Traditionally, such
`
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`2
`
`interference has been considered to be part of the inevitable noise that .corrupts
`transmissions.
`Such interference increases with the number of ac~ive users_ and thus, the
`performance quality in terms of how many users (capacity) that can share the
`resourCE!S simultaneously becomes limited .
`. Figure 1 shows an exemplary multiple access scenario that may occur in
`Wireless Networks. The radio terminals 102, 104 and 100b transmit signals that
`are received at network access point 1 OOa. In general n~t all of these signals are
`·intended for radio terminal 100a. They maybe signals from devices that belong to
`other networks, presumably in unlicensed radio spectrum. In any case there are
`ordinarily some users of interest that belong to the network to which 1 OOa .
`provides access. The Network aims to make arrangements for all of these
`signals to be effectively transmitted. Commonly the users may be required to
`share the radio resource by, for example, transmitting on different frequencies or
`at different times. Such techniques may be wasteful in terms of the expensive
`radio resource.
`The radio terminal 102 may have an associated user 103 who generates
`and· receives information (in the form of voice, video, data etc). Similarly, the
`radio terminal 102 is associated with a user. In the case of a vehicular user 105,
`the vehicle (such as bus, train. or car) may generate and receive data to be
`communicated over the network. This data may.also be generated and received
`by the passengers and/or operators of the vehicle. , The network access point
`'
`100b may also wish to communicate with radio terminal100a as may be the case
`in wireless· backhaul or multihop net.Norks. ln this respect, it is also possible that
`the other users' radio terminals 102, 104 may form part of any multihopping
`network.
`One way to improve capacity Is to introduce error control coding. Applying
`coding all~ws performance to· be improved by only allowing a few of all possible
`combinations of code symbols to be transmitted. Another way is to expioit the
`information contained in the interference. This is known as joint multiuser
`detection. · In systems where both these techniques are used, a decoding strategy
`may be applied which is termed Iterative decoding. Here, a multiuser detector
`first provides an estimate of the transmitted symbols in terms of reliability
`
`\
`
`5
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`3
`
`information. This information is forwarded to decoders that also provide reliability
`information. based on the input from the detector. ·Information is then exchanged
`in an iterative fashion until there are no further improvements. This decoding
`strategy may increase capacity significantly, getting very close to theoretical
`capacity limits at a complexity level within reach of practical implementation.
`However, an optil1"al multiuser detector is prohibitively complex for practical
`implementation, as the inherent compleXity grows. exponentially with the number
`of active users. Instead~ linear multiuser detection based on linear filtering may
`be applied. where the corresponding complexity only grows linearly with the
`number of active users. The inventor has identified that for practical reasons
`related art .linear filters for iterative joint multiuser decoding are based on the
`received signal and the most recent information from the decoders as input to the
`filter. These filters have bee~ designed based on various optimality criteria.
`Where multiple users share common communications resources. access
`to channel resources may be addressed by a multipie access scheme, commonly
`executed by a medium access control (MAC) protocol. Channel resources such
`as available bandwidth are typically strictly limited in a wireless environment. It is
`therefore desirable to use these resou.rces as efficiently as· possible. Allowing
`multiple users to share common resources creates a risk for disturbances and
`interference caused by colliding access attempts. Such disturbances are usually
`referred to as multiple· access interference.
`In wireless local area network
`(WLAN) systems the MAC attempts to schedule transmissions from Stations in
`order to avoid collisions. Sometimes the MAC fails, and Stations access the
`channel iesouices simultaneously .. An example of this situation is illustrated in
`Figure 2, which shows the transmission of packets from a first transmitter station
`1 a second transmitter station 2 and, a representation of received packets at the
`access point shown on the lowermost line. Physical layer receivers may fail to
`recover such collided packets. As the traffic load on the network increases, this
`probiem becomes a significant iimiting factor in terms of ne&york capacity and
`quality of service.
`A different problem, leading to similar effects, is caused by the multipath
`nature qf communication channels associated with. for example, a WLAN. The
`multipath channel causes several delayed replicas of the same signal to arrive at
`
`5
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`1 0
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`4
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`the receiver. This, in turn, creates self-interference similar in nature to multiple
`access. interference discussed above.
`In this case, the problem becomes a
`. limiting factor for the required power to achieve acceptable performance, which
`translates into limitations on the coverage of the WLAN. An example of a direct
`and a reflected version of the original signal arriving at the receiver is ·shown in
`Figure 3, where the direct and reflected transmissions of the packet are illustrated
`on the top two lines as shown. The presence of self interference is .indicated by
`shading in the received signal, represented by the access point on the lowermost ·
`line as shown. Transmission range may be affected by the interference
`10 mechanisms described above and also by the sophistication of the diversity
`signal processing at th~ Receiver. Physical Layer receiver designers therefore
`strive to ensure that effective use is made of all available time. frequency and
`space diversity (the latter_~ay be provided through the use of multiple antennas).
`The inventor has also· identified that when· synchronizing transmitted
`packets over wireless cOnnections ·each packet ordinarily has a preamble of
`several repetitions of the same short signal. A received packet signal may be
`correlated with a delayed version of itself where commonly the delay equals the
`duration of the repeated signal component In the preamble.· This correlation may
`be .implemented repetitively over a given sample sequence. The output power of
`the resultant correlation may then be combined with the average power of the raw
`received signal to define a decision statistic; The point at which the decision
`statistic exceeds a given threshold is selected as the time of arrival of the packet.
`However, there are drawbacks with this technique in: as much as signal
`distortions may be ampiified or accentuated by the processing, invoived with the
`synchronization process producing uncertainties in the determination of packet
`timing.
`Generally, In packet based communication systems it is important to
`reduce latency of a receiver or, in other words, provide as little delay as possible
`between arrival of signals and the decoding of the bits contained in those signals.
`30 Moreover, receiver processes are unable to determine the variation of a ·radio
`channel over the time of a packet length and the associated effect on the
`waveform of the transmitted signal. This may lead to lower than optimum data
`rates due to poor1y tracked packets that are otherwise intact being discarded.
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`In OFDM packet based communication systems channel impairments may
`occur, which contriQute to changing both the channel over which an OFDM signal
`travels and also
`the received signal
`itself. · Collectively,
`these channel
`impairments comprise variations in the transmission channel due to mu1tipath
`.
`.
`fading and, variations to OFDM symbols due to frequency and time offsets
`caused by receiver inaccuracies and phase offsets due to combined transmission
`and reception processes. These channel impairments may vary from OFDM
`symbol to OFDM symbol, in .other words, they may not be invariant over the
`length of a packet.
`Traditionally, channel impairments are countered by
`estimates made using a packet preamble and maintained by pilot symbols
`throughout the received packet; which. may assume invartance over the packet
`length. Other methods use data estimates to aid for example with channel .
`estimation and these are implemented In the frequency domain and may result in
`power loss by discarding a cyclic prefiX for each received symbol. Generally,.
`there is no ·use made of all available received information to address channel
`impairments in such packet based communication systems.
`With regard to space 9iversity, for multiple receiving antennae in wireless
`data packet communication systems related art. schemes provide decisions on
`the synchronization of a received signal on the basis of per antenna and then a
`20 majority vote, otheiWise the received. measurements are added prior to the
`decision. These approaches do not address the variation of signal statistics.
`across the number of antennae resulting in degraded synchronizaticm accuracy
`.and increased packet loss.
`In EP 1387544 it is noted that tima synchronisation Of a receiver to the
`incoming signal is essential for effective decoding of that signal. In many packet
`based applications a special preamble is inserted by the transmitter at the s~rt of
`every packet transmitted in order to assist the 'receiver with its timi"ng estimation
`task. In OFDM _systems the transmitter imparts a special structure on the signal
`cauea a cyciic prefix. This cyciic prefiX is inserted for ·every OFDM symboi. A
`cyclic prefix is a replica of a small portion 9f the. last section of a signal inserted at
`the start c:>f the signal. ·There are many OFDM symbols transmitted sequentially in
`most forms of communication. In EP 1387544 the cyclic prefiX, in the form of a
`guard interval as a cyclic continuation of the last part of the active symbor, is
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`employed to time synchronise the receiver instead of a preamble. In EP 1387544
`a two step time synchronisation approach is disctosed, namely a pre-FFT and
`post-FFT time synchronisation algorithm. These are complementary techniques
`and may be used together. The pre-FFT technique consists of a "delay and
`correlate" algorithm applied to fifld the cyclic prefix of the OFDM symbols. This is
`achieved by setting the delay in the "delay and correlate'' algorithm to the
`distance betw'een the cyclic prefix and the region from which it was . eo pied. The
`output of the correlator is then filtered using an auto-regression filter comprising a
`recursive Infinite-Impulse Respo.nse · (IIR) filter to determine an average of the
`correlation across OFDM symbols. A second filtering, by way of smoother 44 in
`Fig 2· of EP 1387544, is then applied to discard samples outsid.e of the maximum
`delay measurable, namely, the cyclic prefix duration. However, EP 1387544
`relates to a system which makes use of a streaming signal and not readily
`adapted for the random arrival of packets.
`In the case of streaming signal, the
`signal is always there but the fine timing ~ssociated with the OFDM symbol
`boundaries must be determined.
`In US 6,327,314 (Cimini, Jr. et al) the problem of tracking the radio channel
`in a hostile propagation environment is addressed for wireless communications
`systems using OFDM and one or more antennae for reception. The solution
`disclosed by Cimini Jr. employs decoder and demodulator outcomes to generate
`a training or, reference signal, to drive the estimation of the channel for use in
`decoding the next symbol. The decoding, demodulation and channel estimation
`loops run according to the paradigm that the channel estimate may use .all
`outcomes up to· and inciuding the symboi tc;- be decoded. Each OFDM symbol is
`decoded once. The raw channel estimate is obtained by multiplying the received
`OFDM symbol·with the training symbols. These training symbols may be from a
`decoding step. The raw channel estimate, corresponding to one OFDM symbol,
`is stored in· a database. Each time a. new OFDM sym~ol is to be processed all
`raw estimates in the database are employe" to. yield the channei estimate at the
`processing wavefront.
`In this disclosure the raw channel estimates are stored
`and a smoothing step is executed every time the data base is accessed, which
`entails a relative degree of complexity.
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`In US 6,477.210 (Chuang et al) the problem of tracking the radio channel
`in a hostile propagation environment
`is also addressed
`for wireless
`communications systems using OFDM and one or more antennae for reception.
`The solution provid~d in this disclosure augments that disclosed in US 6,327,314
`by more clearly disclosing t~e processing flow and adding a backward recursion
`to the processing. The backward recursion includes the steps of demodulation,
`decoding and channel estimation, as in the forward recursion, but the processing
`commences from th~ end of the packet. Chuang et al is restricted to Maximum
`Likelihood decoding systems such as Viterbl decoders. There are many other
`types of FEC systems that do not employ ML decoding (e.g. Soft Qutput
`Decoders such as A-Posterior Probability techniques} and, moreover, for which
`Chuang is ~ot adapted to operate within.
`In a paper by Czylwik, A., entitled "Synchronization for systems with
`antenna diversity". IEEE Vehicular Technology Conference, Vol. 2, 19~22 Sep.
`1999~ pp 728-732 the time and frequency synchronisation of a receiver is
`considered.
`In order. to successfully decode a packet the receiver must
`determine the packet time of arrival. Errors in this estimate may result in signa