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`
`US 2{)t)40(J47284/\1
`
`(19) United States
`(12; Patent Application Publication am Pub. N0.: US 2004/0047284 A1
`3
`Eidson
`(43) Pub. Date:
`Mar. 11 2004
`
`(S4) TRANSMIT DIVERSITY FRAMING
`STRUCTURE FOR MULTIPATH CHANNELS
`
`I’lIl')IlCaJtl0I1 Clas-sificatinn
`
`(76)
`
`Inventor: Dunaltl Brian Eidson. San Diego, (TA
`(US)
`C
`ct-
`Add ~:
`S:I:1[:;:(iFa]l;1&::tsky,rLl;:q.
`CONEXANT SYSTEMS’ INC‘
`4311 Iambome Road
`Newport Beach’ CA 926606095 ms)
`
`(21) APPL No_:
`
`103893272
`
`(32)
`
`Fihgd:
`
`Mm-_ 14, 2003
`
`Related U.S.AppIicati{]n Data
`
`(63)
`
`(.‘entinuation~in—part of application No.
`liled on Mar. 13, 2002.
`
`lUt'(!99,550,
`
`Int. Cl.’ .................................................... .. H041 lla’00
`(51)
`
`(52) U.S. Cl.
`.......... .. 370x203
`
`(57)
`ABS'l‘RAC'l‘
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`Systems and tecltntgues are dtsclosecl lor framln-g and pro-
`cesstng smgle—earr1er andfer
`()i~DM transm11—d1verstty
`Iransmtsstons lh rough delay—sprea:i channels, as well as Izor
`deframtng and processing the eurrespontltng merged signals.
`received from a plurality of antennas, to estimate the trans-
`mitted information.
`'I'in'te-dornairt processing techniques
`may_1)e used l’or‘l)oti1 types of transmissiotis to create
`multiplexed dual signal-unit pairs. particularly when cyclic
`prefixes are needed to reduce delay-spreading effects.
`Repetitive pilot words may be employed in burst preambles
`andfor in payloads of transtnission bursts to minimize or
`provide flexibility in the amount of bandwidth that is con-
`sumed to generate good channel response estimates under
`changing channel conditions.
`
`Block Pair
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`
`Patent Application Publication Mar. 11, 2004 Sheet 1 of 7
`
`US 2004/0047284 A1
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`Patent Application Publication Mar. 11, 2004 Sheet 2 of 7
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`US 2004/0047284 Al
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`Patent Application Publication Mar. 11, 2004 Sheet 3 of 7
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`US 2004/0047284 A1
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`Patent Application Publication Mar. 11, 2004 Sheet 4 of 7
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`US 2004/0047284 A1
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`Patent Application Publication Mar. 11, 2004 Sheet 5 of 7
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`US 2004/0047284 A1
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`Patent Application Publication Mar. 11, 2004 Sheet 6 of 7
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`Patent Application Publication Mar. 11, 2004 Sheet 7 of 7
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`
`US 2004/0047284 A1
`
`Mar. 11, 2004
`
`TRANSMIT DIVERSITY FRAMING STRUCTURE
`FOR MUI..TII’ATH CHANNELS
`
`[0001] This application is a continuation-in-part of U.S.
`patent application Ser. No.
`l0lU99,556, “Transmit Multi-
`plexing and Receive Processing for Delay Spread Chan-
`nels," fI.le(l Mar. 13, 2002, the entire contents of which is
`hereby incorporated by reference herein.
`
`BACKGROUND OF THE INVENTION
`
`[0002]
`
`1. Field of the Invention
`
`[0003] This invention generally relates to wireless com-
`munications links, and, more specifically. to frame structures
`for diverse antenna transmissions.
`
`[0004]
`
`2. Related Art
`
`[0005] Virtually all wireless communication channels are
`limited in their ability to accurately communicate data by the
`signal—to—noise ratio (SNR) of the wireless channel. Antenna
`diversity is one category of techniques that may be used to
`enhance the effective SNR ofconimunications channels, and
`thus enhance the ability to accurately transmit data.
`
`[0006] Antenna diversity can be incorporated at the trans-
`mitter, or at the receiver, or both. However, for the cost-
`sensitive subscriber station, diversity is much cheaper if it is
`instantiated at the base station transmitter (where its benefits
`and costs may be shared by all subscribers), rather than at
`every subscriber station. In installations where a great deal
`of diversity is required for reliable service, multiplicities of
`diversity may be achieved if the base station transmitter and
`the subscriber
`station receiver both possess diversity.
`Mechanisms to realize transmit diversity are of great utility
`for wireless communications.
`
`[0007] Many transmit diversity techniques have been pro-
`posed in the literature. One such technique is transmit delay
`diversity. At the transmitter, delay diversity is achieved by
`using two antennas that transmit the same signal, with the
`second antenna to transmitting a delayed replica of that
`transmitted by the first antenna. By so doing. the second
`antenna creates diversity by establishing a second set of
`independent multipath elements that may be collected at the
`receiver. If the multipath generated by the lirst transmitter
`fades, the multipath generated by the second transmitter may
`not, in which case an acceptable SNR will be maintained at
`the receiver. This technique is easy to implement, because
`only the composite TXO+TXl channel is estimated at the
`receiver. Transmit delay diversity does not
`require the
`receiver to have special a-priori knowledge that the trans-
`mitter is using this type of diversity, because the receiver's
`equalizer compensates automatically for the additional mul-
`tipath diversity induced by the second transmit antenna.
`
`[0008] Both OFDM and single carrier modulation can
`easily implement a delay diversity scheme. The biggest
`drawback to transmit delay diversity is that it increases the
`effective delay spread of the channel, and can perform
`poorly when the multipath introduced by the second antenna
`falls upon, and interacts destructively with, the multipath of
`the first antenna,
`thereby reducing the overall
`level of
`diversity.
`
`[0009] Another transmit diversity technique of lo\.v-to-
`moderate complexity is described in “A simple transmit
`diversity scheme for wireless communications," S. Alam-
`
`outi, IEEE Journal on Select/lrerr.s in Comrrttmicarioms, vol.
`16, no.8 Oct. I998, pp. i451-1458. This technique provides
`two-way maximal ratio-combining diversity. Unfortunately,
`the Alamouti transmit diversity scheme cannot be directly
`applied to systems experiencing delay spread, because it
`relies on an ability to isolate pairs of multiplexed symbols
`from each other, that is, the receiver must be able to process
`each pair of symbols without significant interaction from
`other pairs of symbols.
`In delay-spread channels, where
`symbol energy not only overlaps other symbols, but indeed
`may span hundreds of symbols, such absence of interaction
`cannot be relied upon. A transmit diversity technique that
`overcomes some of the limitations of the foregoing is
`described herein.
`
`[0010] Transmit diversity techniques rely upon estima-
`tions of the symbol content of received signals. Estimating
`and compensating for transfer characteristics of the wireless
`channel, in turn, generally improves the symbol estimates.
`Irrespective of the basic transmit diversity technique used,
`techniques to enhance the symbol estimation will improve
`the overall ability of a communication system to accurately
`transfer data. Accordingly, there is a need for techniques to
`enhance the data transmission elfectiveness ofbasic transmit
`diversity multiplexing techniques.
`SUMMARY
`
`Processing techniques and framing structures to
`[0011]
`enhance the effectiveness of transmit-diversity wireless
`communications, and systems employing such techniques
`and structures, are disclosed herein that may be used to
`enhance the effectiveness of communications transmitted by
`diverse antennas, particularly when the transmission chan-
`nels have delay-spread characteristics. Multiplexing tech-
`niques to provide a plurality of signals for a corresponding
`plurality of transmit antennas are disclosed, as well as
`corresponding receiver combining and equalization tech-
`niques. Data structures are also disclosed for use in con-
`junction with diversity multiplexing techniques, particularly
`for delay-spread channels. Framing and processing tech-
`niques are disclosed that are applicable to single-carrier
`andfor OFDM transmit—diversity transmissions and recep-
`tions through delay-spread channels.
`
`[0012] One embodiment is a method of transmitting dual
`signal-unit pairs from diverse antennas. It includes process-
`ing a plurality of N-point signal units each into a plurality of
`forms using time-domain techniques, and prepending a
`cyclic prelix on each of the resulting forms, before trans-
`mitting the prefixed forms of the signal units in concurrent
`pairs from the diverse antennas.
`
`[0013] Another embodiment is a method of interpreting
`received signals that were transmitted in multiplexed forms
`from diverse transmit antennas. The method includes iden-
`tifying received pilot words that include cyclically prefixed
`first and second pilot signal units. The method further
`includes ignoring the cyclic prefixes and combining forms of
`the lirst and second pilot signal units with lirst and second
`expected pilot symbol units to derive first and second
`channel estimates. The method further includes deriving
`estimates of transmitted payload signal units by combining
`forms of the channel estimates with forms of received
`payload signal units.
`
`[0014] A further embodiment is a method of transmitting
`dual signal-unit pairs from diverse antennas. The method
`
`9
`
`

`
`US 2004/0047284 A1
`
`Mar. 11, 2004
`
`includes deriving a plurality of pilot signal units from
`portions of pilot data expected by a receiver, establishing a
`variant fonn of each signal unit, and cyclically prefixing the
`signal units and their variants. The method also includes
`transmitting, substantially concurrently, appropriate pairs of
`these various cyclically prefixed signal units. The method
`yet further includes transmitting a repetitive pilot signal unit
`by transmitting an appropriate signal unit pair one or more
`additional times, without cyclic prefixes. immediately after
`they have been transmitted with a cyclic prefixes.
`
`[0015] Yet another embodiment is a system that may be
`used for transmitting dual signal-unit pairs from diverse
`antennas. The system includes first and second antennas and
`a signal-unit derivation block configured to derive N-point
`signal-units cl" time-domain samples from modulated source
`information. It also includes a diversity multiplexer block
`configured to multiplex pairs of the derived N-point signal
`units into multiplexed dual signal-unit pairs, each having
`first and second N—point mulliplexec|—signal—units (“MSUs")
`for the first antenna, and first and second N-point MSUs for
`the second antenna, where the first N-point MSU for the first
`antenna is related to the second N—p0int MSU for the second
`antenna by complex conjugation and modulo-N sample time
`inversion, and the second N-point MSU for the first antenna
`is related to the first N-point MSU for the second antenna by
`complex conjugation, negation, and modulo-N sample time
`inversion. The system also includes a first output processing
`block configured to cyclically prefix the first and second
`N-point MSUs for
`the first antenna, and to process the
`prefixed MSUs for sequential transmission from the first
`antenna; and a second output processing block configured to
`cyclically prefix the first and second N-point MSUs for the
`second antenna, and to process the prefixed MSUs for
`sequential transmission from the second antenna substan-
`tially concurrently with the sequential transmission from the
`first antenna.
`
`[0016] Yet a furtlier embodiment is a receiver system that
`may be used for receiving paired multiplexed signals trans-
`mitted from plural antennas. The system of this embodiment
`includes a receive and alignment block ccnfigu red to receive
`and
`align prefixed multiplexed—signal—units {“MSUs")
`received sequentially in a frame structure having a preamble
`portion and a payload portion, and a cyclic prefix removal
`block configured to remove cyclic prelixes lirorn received
`MSUs. The system also includes a pilot word identification
`block configured to identify,
`in accordance with relative
`position within the frame structure, J concatenated copies of
`a first received pilot MSU, RPD, followed by P concatenated
`copies of a second received pilot MSU. RP],
`that were
`transmitted based upon a first expected pilot signal—unit EPU
`and a second expected pilot signal-unit EP1, and a channel
`estimation block configured to combine a representation of
`RFC, and a representation of RP, with complex conjugated
`forms of EP.,ar1d El’, to create a first channel estimate HE“
`and to combine the representations of RPO and RP, with
`forms of EPO and EP.
`that are not complex conjugated to
`create a second channel estimate HE3.
`
`BRIILEF DIESCRIPTION OF TIIE DRAWINGS
`
`[0017] Embodiments ofthe present invention will be more
`readily understood by reference to the following figures, in
`which li.ke reference numbers and designations indicate like
`elements.
`
`[0018] FIG. 1 illustrates temporal organization of a block
`pair.
`
`[0019] FIG. 2 illustrates concatenation of block pairs for
`transmission.
`
`[0020] FIG. 3 is a matrix showing a dual block pair signal
`multiplexing relationships.
`
`[0021] FIG. 4 is a signal
`Feedback Equalizer.
`
`flow diagram of a Decision
`
`[0022] FIG. 5 shows a Unique Word variation of the
`multiplexing of FIG. 3.
`
`[0023] FIG. 6 illustrates a burst
`Unique Word preamble.
`
`transmission using a
`
`[0024] FIG. 7 shows Pilot Words disposed in a general
`payload transmission.
`
`[0025] FIG. 8 illustrates a use of cyclic prefixes with the
`multiplexing of FIG. 3.
`
`[0026] FIG. 9 illustrates using Unique Words for cyclic
`prefixes in block pairs.
`
`[0027] FIG. 10 shows a dual block pair multiplexing
`transmission format.
`
`[0028] FIG. 11 illustrates communication system features
`including plural receivers.
`
`[0029] FIG. 12 is a block diagram of transmit diversity
`processing for sing1e—carricr applications using time domain
`multiplexing.
`
`[0030] FIG. 13 illustrates dual block pair signal multi-
`ptexing relationships for OFDM.
`
`[0031] FIG. 14 is a block diagram of transmit diversity
`processing for OFIJM applications using frequency domain
`multiplexing.
`
`[0032] FIG. 15 shows a modification of the processing of
`FIG. 14 to use time domain multiplexing and avoid some
`FFTs.
`
`[0033] FIG. 16 is a block diagram showing OFDM trans-
`mit diversity receive processing.
`
`[0034] FIG. 17 illustrates an example of preamble fram-
`ing for OFDM transmit diversity.
`
`DETAILED DESCRI PITON
`
`[0035] Multiplexed data may be transmitted over two or
`more antennas, as described more fully below, to enhance
`the reliability of communication with a receiver (or receiv-
`ers). The techniques described are particularly clfective
`when the communication is conducted over multipath delay
`spread channels. Two-antenna diversity can double the
`effective diversity level of a system operating over such
`channels. Of course, multipath.
`in itself.
`is
`a
`form of
`diversity. Thus, for example, with a system operating over a
`single channel with three multipaths and having (there£ore)
`a diversity level of three; use of two transmit antennas (with
`one receiver) as described below could increase the diversity
`level to 6 {or more). Entbodiments that further employ two
`or more receivers may increase the diversity gains even
`further.
`
`10
`
`10
`
`

`
`US 2004/0047284 A1
`
`Mar. 11, 2004
`
`L»)
`
`Single Carrier Transmnit Diversity
`
`[0036] This description assumes a communication system
`that is configured to transfer selected signals from a trans-
`mission system having a plurality of transmit antennas to a
`receiving system having one or more antennas. The desired
`transmission signals are assumed to at least partly take the
`form of symbols defined in the time domain. FIG. 1 illus-
`trates that a selected number B ofsymbols may be organized
`as a symbol block, such as a "Block 0"l02 or a "Block
`'l"'l04. ideally, a delay spread guard period 106 is provided
`before each block (eg, 102 and 104). For convenience,
`delay spread guard periods will typically have the same
`length, D, which may be measured in symbol lengths or
`time. The length, D, of these delay spread guard regions
`would typically be longer than the delay spread span of the
`channel.
`ln one embodiment,
`these delay spread guard
`regions would have a ‘cyclic prefix’ format; ie. they would
`be composed of the last D symbols of the block that follows
`them. A combination of sequential blocks, such as the
`illustrated Block 0 and Block 1, each preceded by a delay
`spread guard having a known length that may range down to
`zero, fonts a block pair 100. Note that the two blocks within
`a block pair are logically paired together, but physically
`separated from themselves (or other data) by delay spread
`guard regions 106. in general, any particular block, such as
`the "Block 0"102, may be the same or dilferent from the
`other block (e.g., "Block l"104) of its block pair 100. It will
`generally be useful, for minimizing bit error rate (BER)
`characteristics, if the channels from each antenna to the
`receiving. system do not change significantly between the
`beginning and end of transmission of a block pair. Note that
`the symbols within a block should be adjacent, but
`that
`blocks composing a block pair do not have to be adjacent,
`although they are pictured that way in FIGS. 1 and 2.
`
`[0037] FIG. 2 illustrates a sequence of block pairs, includ-
`ing block pairs (m—1) 202, (m) 204, (m+1) 206 and (m+2)
`208, which have been concatenated for transmission as an
`extended payload from a particular antenna. In general, any
`block pair transmitted from a particular antenna may be
`different from or identical to any other block pair transmitted
`from the same antenna at another time.
`
`[0038] Block Pair Transmit Multiplexing
`
`[0039] FIG. 3 indicates a block multiplexing structure that
`a two-antenna transmitter may use to transmit the informa-
`tion of each of two sequences in two related forms over
`block pairs 302 and 304 that resemble the block pair 100
`(see FIG. 1). {s0[n]} is a signal set describing a first block
`306 of the block pair 302 transmitted by Transmit Antenna
`0, while {s,[n]} is a signal set that describes a second block
`308 of the block pair 302. {sO[n]} and {s,[n]} each describe
`a sequence of length B symbols, 0% n-cl?-1, that represents
`information that is to be delivered to a receiver via the two
`transmit antennas.
`
`[0040] A block 310 is the first block of the block pair 304
`transmitted by Transmit Antenna 1, while block 312 is the
`second block of the block pair 304 transmitted by Transmit
`Antenna 1. The first block 306 of the block pair 302 conveys
`the same information as the second block 312 of the block
`pair 304, but in a ditlerent form. As compared to the block
`306,
`the block 312 is a
`time-inverted sequence of the
`complex conjugate of the symbols that form the symbol
`sequence
`Similarly, the first block 310 of the block
`
`pair 304 is a time—invened sequence of the negative complex
`conjugate of the symbols of the sequence {s,[n]}.
`
`[0041] Thus, Transmit Antenna 1 transmits blocks having
`the same information as is transmitted by related blocks of
`Transmit Antenna 0, but in reverse time order, and the blocks
`of Transmit Antenna 1 have a sequence of symbols that are
`the {positive or negative} complex conjugate of the symbols
`of the related blocks of Transmit Antenna 0 in a sequence
`that is also time-reversed cyclically about zero, modulo-B.
`
`It should be understood that the signal sets {so[n]}
`[0042]
`and {s,[n]} are only nominally in an “unmodified” form.
`Either or both {s0[n]} and {s,[n]} may, of course, be related
`to other symbol sequences that are the actual symbols which
`are being sent. Thus, either of these signals may in fact be,
`for example, a time-inverted, negated andfor complex-com
`jugated version of the actual desired symbols. This merely
`reflects; the generality of the signal sets {so[n]} and {s][n]},
`and does not affect the relationship between blocks that are
`diagonally positioned within the space-time matrix ofblocks
`shown in FIG. 3.
`
`[0043]
`
`Transmit and Receive Signal Relationships
`
`[0044] Define S,,(ej“’), S,(ej‘°}, Nn(el"’), N1(ei"’), I-In{e’_‘”),
`and Il1(cj“’)
`as
`the Discrete-time Fourier
`transforms
`(DTFTS), respectively, of the symbol sequences
`and
`{s,[n]} (each normalized, without loss of generality, so that
`their average symbol energy is 1); additive white, zero-
`mean, oz-variance noise sequences {n0[n]} and {n,,[n]}; and
`the channel impulse responses {h0[n]} and {h,[n]} that are
`associated with each transmit antenna Due to the multi-
`
`plexed transmit signals, each received block of payload
`symbols (which is typically separated from other blocks by
`delay spread guard intervals) includes information from both
`blocks of a received block pair. Accordingly, the information
`from individual blocks can only be extracted by combining
`information from the two blocks of a block pair. The
`received signals associated with each individual block of a
`block pair, interpreted in the frequency domain, are:
`
`Rate”) = Hote*“:e'oteJ"~'t— rn:e*~ts;teI~t+N.,te=”t
`R|[£”""] = .‘-tat:-"”1S'.te""I+ .’fjtel'”]S.§t¢-""'t+ N1 te"“’t_
`
`Entn-
`
`1
`
`[0045] Note that l)'l'FTs are used for frequency domain
`descriptions for generality, but this is in no way limiting: a
`B—point (or, more generally, an implementation using a
`K-point) Discrete Fourier Transforrn (D171) would uni-
`formly sample the DTFT response over the interval o:e[0,
`21), yielding B (or K) samples.
`
`[0046] The received signal is a merger of the concurrently
`transmitted blocks, so processing facilities should identify
`received block pairs (through their time relationship with a
`preamble, for example), and combine forms of the received
`block pairs with forms of the channel response estimates.
`Assuming that
`the frequency domain channel
`responses
`Ho(ei"’) and l-l,(e5"’) are known (or estimated}, a received
`block pair R0 and R1 may be filtered and combined accord-
`ing to the frequency domain combining scheme
`
`11
`
`11
`
`

`
`US 2004/0047284 A1
`
`Mar. 11, 2004
`
`Cote”: =H'5tc"“’tRnte”"] +H'1{e""tR]t&:"")
`cheer = —H.ter~iir.3i'c~: + rrg.iet”iR1te~:'
`
`E111 3
`
`El?” lfzifln, =
`
`l
`Di are 1 '
`
`Eqn. 8
`
`Processing the block pair. R0 and R1, thus includes
`[0047]
`filtering various forms of the received blocks with a form of
`a channel estimate, and in the frequency domain the filtering
`may consist of multiplying a form of a channel estimate by
`a form of a received block. The appropriate filtered results
`are then combined (in the frequency domain case, added) to
`produce a pair of combiner outputs C0 and C1. Using the
`expanded representation of the received blocks provided by
`Eqn. 1, the combiner outputs are seen to be
`
`tL'n[¢-3'”) = [II-r..tt-F“ ,1]: + |.'-tl [at-"”J]:]5'ntr:"“'] +
`tr5qe*“tNo:c'“‘i+mtc*"tN;tc1‘*‘t
`
`Erin 3
`
`(qr;-‘"1 = [|rr.,;ei~_:f + |r+,ter~;|°}s.t.»i~a _ '
`H.teew5teP1+rr;,u-*":N.teJ”i
`
`[0048] Applying the following definitions:
`
`Dte’“}fil|Hote”"}I2+|H|tF’"'l|2l
`and
`
`N5-ate“! = rrgte"“iNgtc*"1 + I-t1te""_iN;te"‘"i
`
`Nc1te"“'t= _,miePiN.;:e"“i + rr5te""iN1te"“i'
`
`Eat‘ 4
`
`Erin. 3
`
`the expressions for the frequency domain reprcsenv
`[0049]
`tation of the combiner outputs C,-,(e"") and C](eJ‘°) become
`
`Cate“! = D[e“”l.5'otF’”t+Ncc,tc"“t
`
`1"=rirI- 6
`
`C; LN“! = Dt¢*"“"t.S‘.t¢=""t + Nr.-1tr"“t
`
`In view of Eqn. 6, estimates of S0(ej°') and S1(ei‘")
`[0050]
`may be obtained using any equalization technique that will
`substantially remove the influence of l'J(eJ‘”).
`
`[0051] Equalization
`
`[0052] Many equalization techniques exist. A linear equal-
`izer may be used in an effort to eliminate D(eJ‘”) through
`pre-multiplication by an appropriate equalizer characteristic
`that is generally inverse to D{ej"’):
`
`éfllepltiiircrr = Elcwiiirararcfllelml
`
`31 :c'“1,,.,,,,,, = an-=‘“ 1,,-,,,,,,Ct te““’t
`
`Eqnl
`
`if
`
`[0053] Linear equalizer functions may take various forms.
`As examples, the linear equalizer solution obtained using a
`zero forcing (ZF) optimization criterion would be
`
`[0054] whereas a linear equalizer solution obtained using
`a Minimum Mean Squared Error (MMSE) optimization
`criterion would be
`
`-
`‘5“”“"ii-if =
`
`I
`
`E . 9
`in
`
`[0055] Because D is defined in Eqn. 4, frequency domain
`processing facilities may therefore equalize the combiner
`outputs by dividing each output by a quantity that reflects a
`sum of the individual channel response magnitudes. In the
`case of MMSE, the divisor sum further includes a term 01
`that retlects a normalized reciprocal of the signal to noise
`ratio [SNR) measured for the received signal. The aforesaid
`equalization results may be directly interpreted as estimates
`S0(eJ'”) and §1(eJ“’), whereupon {soft} equalized estimates of
`the symbol sequences {s0[n]} and {s,[n]} may be obtained
`directly by computing inverse D'IT"Ts (I-DTFTS) of the
`estimates.
`
`Instead of performing frequency domain process-
`[0056]
`ing, the facilities may perform combining and equalization
`in the time domain using block lcngth—B circular convolu-
`tions (denoted by the operator ‘GD, where
`
`gnlnlfhamr = Wamrlfll '31-"nltil
`3-. tnia... = amt»: es. tn] ‘
`
`cola] = itfiltfl-it}InodtBl]®.n3[rr] +
`in [H] 1% r[ [[3 — 1:] modtflj]
`r‘. [n] = —Fr[[n] ®r'E,|(B — rrtmodiflt] + I
`ir[,[(B — I'll rnod(Bt] ®r. [rt]
`
`E9” 10
`
`Eqn 1]
`
`and the lower-case variables are time domain rep-
`[0057]
`resentations of upper-case frequency domain variables.
`Thus, in the tine domain the combiner outputs may be based
`on forms of received blocks filtered by forms of channel
`estimates, just as in the frequency domain.
`In the time
`domain case, the various forms of the received blocks. and
`of the channel estimates, may difier by not only complex
`conjugation (positive or negative), but also by time-reorder-
`ing of the symbol sequences, and in particular by cyclic time
`reversal of the time sequence of the corresponding block
`symbols. The equalization filtering may be performed by
`circular convolution between block-length sequences. The
`equalizer e“"°"[n] may be derived from an l—D’I‘[-‘T of
`E(ei-»),im, if frequency domain information on the channel
`impulse responses is immediately available. If not, the time
`domain responses may be frequency transformed to generate
`I I0(e’“') and IIl(e]‘”), and then I_1(e"")Hrm, May be derived, for
`example, by further processing according to Eqn. 8 or Eqn.
`9.
`
`[0058] FIG. 4 Illustrates equalizer elements In an equal-
`izer subsystem that may be less prone, than a linear equal-
`
`12
`
`12
`
`

`
`US 2004/0047284 A1
`
`Mar. 11, 2004
`
`izer, to emphasize noise in frequency bands where notches
`occur. FIG. 4 shows a signal to be equalized 402 progressing
`through a typical Decision Feedback Equalizer (DFE) hav-
`ing two equalizer processing elements: a feedforward (lin-
`ear)
`filter 404, and a decision feedback filter 406 that
`subtracts symbol decisions made in the time domain. The
`linear feedforward filter element 404 generates intermediate
`equalized results
`
`-continued
`
`“Item = _\:ot¢»"”tR,te!'"2}—S,ti-"“ti?:.[e!'“t _
`|Sg[£JWJ] +[.S‘.ter~tl'
`.s'ot¢~**‘tN;:e“t—5.:¢~*“iNorc"t
`
`|Sot¢=J"u 1|: 4- |.S‘.teJ‘"t[2
`
`74,1;-1"-'t= r_':eIw;,.,_c.;uv_-M:
`Ztte""t= Eio""iF,c1:e*"i"
`
`Eqn. 12
`
`[0065] This leads to calculations that may be performed by
`the receiver signal processing facilities to obtain channel
`estimates, including the frequency domain MMSE estimates
`
`and an I-DTFT may be applied to the equalized
`[0059]
`results to yield time domain sequences z0[n] and z,[n]. Each
`ofthese sequences may then be separately operated upon in
`the time domain by the decision feedback filter 406, and may
`use identical feedback coeflicients f[n] to provide an equal-
`ized signal 408 in the form of estimates {§Q[n]} and {§.,[n]}.
`
`[0060] A MMSE criterion-optimizing DFE may use the
`feedforward filter
`
`.
`
`3’
`
`5' PR F‘
`""R H“
`|sotemt- + |5';(¢rJW 1| + r!’-
`.5‘ te-""13 “I-5 ME to”)
`A
`Hit-?"")mrsr = ‘g‘:
`tS'o:eJ'"tl‘ + IS: te"*'t|‘ + U’-
`
`1"=t1rL 16
`
`[0066]
`
`and zero-forcing estimates
`
`MMSE
`Eli-?"“"JF;
`
`Flew]
`= .
`
`E9” 13
`
`_ S[,tr*""lRnte"‘“l+5[le"'“!Rltc"’]
`F! (W:
`°' ”’
`Isou.»r»i:’+ts‘.te»~i|*
`.
`.8‘
`""13? 9”“ -5 Ma Pa"
`1,-_~me,.,,)zF= ole
`it
`J
`tie
`ole
`|s.,ieJ~i|3 +|5'.t¢-MI’
`
`Erin 1?
`
`[0061] where F(ej"’) is the DTFT of l[rr]. A zero-forcing
`solution would be identical to the MMSE solution, except
`that it would not possess the f found in Eqn. 13, which
`reflects the normalized reciprocal of the receiver SNR.
`
`[0062] The frequency domain filtering described in Eqn.
`12 may,
`in the alternative, be implemented in the time
`domain. The signal processing may filter the combiner
`outputs by circularly convolving them with an equalization
`sequence eFF[n]:
`
`zolnltinmr = €'rFl"l ®Fol"l
`-It l”llr'rmr = 9n’ lfll®f‘| lfll .
`
`Etltt 14
`
`[0063] where e,_.,{n] is the I-DTFT of l3(e"“’)FF(in either its
`MMSE or zero-forcing forms).
`
`Channel Estimation
`
`It is, of course, very helpful to obtain good channel
`[0064]
`estimates, since the estimates of the transmitted sequences
`may depend upon the channel estimates at a number of
`stages. Manipulation of Eqn.
`1 reveals that the frequency
`domain channel characteristics can he expressed as
`
`[0067] The channel estimates oi'Eqns. 16 and 17 are based
`upon arbitrary symbol data sequences {s0[n]} and {s,[n]}.
`Unfortunately,
`the arbitrary symbol data sequences must
`generally be estimated themselves, thus compounding any
`inaccuracies. It may be useful to avoid relying exclusively
`on such estimates for further deriving estimates of the
`channel response.
`Prearnbles and Pilot Words
`
`[0068]
`
`[0069] The channel estimation process will be less reliant
`on received symbol estimates when known sequences of
`Symbols are transmitted at identifiable times. These known
`sequences may be referred to generally as “pilot symbols,"
`although it should be understood that such sequences might
`also appear in preambles, or in other forms. In many cases,
`an identical sequence will be consistently t