(12) United States Patent
`Laroia et al.
`
`(10) Patent N0.:
`(45) Date of Patent:
`
`US 6,985,433 B1
`Jan. 10, 2006
`
`US006985433B1
`
`METHODS AND APPARATUS FOR
`DETERMINING MINIMUM CYCLICPREFIX
`DURATIONS
`
`W0
`
`WO 00/01084
`
`1/2000
`
`OTHER PUBLICATIONS
`
`(54)
`
`(75)
`
`Inventors: Rajiv Laroia, Basking Ridge, NJ (US);
`Junyi Li, Pine Brook, NJ (US)
`
`(73)
`
`Assignee: Flarion Technologies, Inc., Bedminster,
`NJ (US)
`
`(*)
`
`Notice:
`
`Subject to any disclaimer, the term of this
`patent is extended or adjusted under 35
`U.S.C. 154(b) by 852 days.
`
`(21)
`(22)
`
`(60)
`
`(51)
`
`(52)
`(58)
`
`(56)
`
`Appl. No.: 09/689,273
`
`Filed:
`
`Oct. 12, 2000
`
`Related US. Application Data
`
`Provisional application No. 60/233,000, ?led on Sep.
`15, 2000.
`
`Int. Cl.
`(2006.01)
`H04] 11/00
`US. Cl. ..................................... .. 370/208; 370/339
`Field of Classi?cation Search .............. .. 370/203,
`370/208, 210, 310, 328, 332, 334, 339, 343,
`370/344, 464, 465, 480; 375/130, 132—134,
`375/260
`See application ?le for complete search history.
`
`References Cited
`
`U.S. PATENT DOCUMENTS
`
`5,915,210 A
`6,061,405 A
`6,362,781 B1 *
`6,535,550 B1 *
`6,687,307 B1 *
`
`6/1999 Cameron et al.
`5/2000 Emami
`3/2002 Thomas et a1. ........... .. 370/344
`3/2003 Cole ........................ .. 375/222
`2/2004 Anikhindi et a1. ........ .. 370/344
`
`FOREIGN PATENT DOCUMENTS
`
`EP
`
`0 982 906 A
`
`1/2000
`
`R. PerveZ and M. NakagaWa, “Parallel Coded Optical Mul
`ticarrier Frequency Division Multiplexing—A Potential
`Step ToWards High Speed, High Capacity and High Reli
`ability in Optical Transmission Systems”, IEICE Transac
`tions on Communications, V. E79 B, No. 11, pp. 1677-1686,
`Nov. 1996.
`J. Vankka, M. Kosunen, J. Hubach, and K. Halonen, “A
`Cordic-based Multicarrier QAM Modulator”, Global
`Telecommunications Conference—Globecom ’99, General
`Conference (Part A), pp. 173-177.
`
`* cited by examiner
`
`Primary Examiner—John PeZZlo
`Assistant Examiner—Donald L Mills
`(74)Att0rney, Agent, or F irm—Straub & Pokotylo; Michael
`P. Straub
`
`(57)
`
`ABSTRACT
`
`Methods and apparatus for generating and transmitting
`frequency division multiplexed signals are described. Each
`transmitted FDM signal is generated by combining, e.g.,
`multiplexing, a plurality of individual analog subcarrier
`signals together. Individual analog subcarrier signals are
`generated by processing one or more digital signals, e.g.,
`symbols plus a cyclic pre?x for each symbol, corresponding
`to the subcarrier to generate an analog subcarrier signal there
`from. A cyclic pre?x incorporated into subcarrier signals is
`designed to be of suf?cient length that it covers delays
`introduced by ?lters in the individual subcarrier signal paths
`and common signal path. The incorporated cyclic pre?x has
`a minimum duration determined as a function of several
`group signal dealys including a common signal path group
`signal delay and a communications channel group signal
`delay.
`
`7 Claims, 9 Drawing Sheets
`
`902
`
`CALCULATE CSPD
`F0
`
`N CSDPs I
`ANALYZE
`CALCULATED
`"V905
`CSFDS AND
`IDENTIFY LDNGEST
`CSFD
`
`CALCULATE
`CSPCD
`CSFCD
`
`908
`
`TERMINE
`TRANSMISSION
`CHANNEL DELAY
`(TCD)
`
`TCD
`
`LCSPDv
`CONVOL E
`CSPCD and
`TCD TO
`AL ULATE THE
`DURATION OF THE
`CYCLIC PREFIX
`
`912
`
`STORE
`CALCULATED
`
`OUTPUT
`CALCULATED
`CVCLIC PREFIX
`DURATION
`
`STOP
`CYCLIC PREFIX DURATION
`CALCULATION ROUTINE
`
`AT&T, EXH. 1018, Page 1 of 16
`
`

`
`U.S. Patent
`
`Jan. 10, 2006
`
`Sheet 1 0f 9
`
`US 6,985,433 B1
`
`AT&T, EXH. 1018, Page 2 of 16
`
`

`
`U.S. Patent
`
`Jan. 10, 2006
`
`Sheet 2 0f 9
`
`US 6,985,433 B1
`
`3804
`
`SINUSOID SIGNAL GENERATOR
`Constellation
`
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`0m
`olnt
`OFDM I
`I
`0 O I
`s mbolk
`y
`> I 307
`
`308
`l
`Sinusoid Signal
`GeneratorCircuit for
`Sub-carrier k, fk
`
`[Bk] and Elk are selected from one of the
`constellation points by element 307
`
`FIG. 3A
`
`305
`8
`SINUSOID SIGNAL GENERATOR 3120
`Constellation
`G Cosine Signal‘
`enerator ircuit for
`-
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`Pom“ Pmmz
`Sub-carrierk,(fk-fc)
`OFDM I
`I
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`S
`S
`S mbo|k
`inusoid ignal
`_ y
`307
`Generator Circuit for
`Sub-carrier k, (fk-fc)
`
`[Bk] and GK are selected from one of the
`constellation points by element 307
`
`3312
`
`FIG. 3B
`
`[Bflcosmnfkvm?
`'
`
`[Bk]cos(2rrfkt)+9k)
`—>
`
`AT&T, EXH. 1018, Page 3 of 16
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`

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`U.S. Patent
`
`Jan. 10, 2006
`
`Sheet 3 0f 9
`
`US 6,985,433 B1
`
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`AT&T, EXH. 1018, Page 4 of 16
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`

`
`U.S. Patent
`
`Jan. 10, 2006
`
`Sheet 4 of 9
`
`US 6,985,433 B1
`
`TS
`
` STAGE 2
`
`AT&T, EXH. 1018, Page 5 of 16
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`AT&T, EXH. 1018, Page 5 of 16
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`

`
`U.S. Patent
`
`Jan. 10, 2006
`
`Sheet 5 of 9
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`US 6,985,433 B1
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`AT&T, EXH. 1018, Page 6 of 16
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`

`
`U.S. Patent
`
`Jan. 10, 2006
`
`Sheet 6 0f 9
`
`US 6,985,433 B1
`
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`AT&T, EXH. 1018, Page 7 of 16
`
`

`
`U.S. Patent
`
`Jan. 10, 2006
`
`Sheet 7 0f 9
`
`US 6,985,433 B1
`
`9302 A
`
`830
`
`START
`CYCLIC PREFIX DURATION
`CALCULATION ROUTINE
`
`904
`
`I
`CALCULATE CSPD
`FOR EACH OF THE
`N SUBCARRIER
`SIGNALS
`
`I,
`
`CALCULATE
`CSPCD
`
`910
`I
`I
`DETERIvIINE
`TRANSMISSION
`CHANNEL DELAY
`(TCD)
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`
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`908
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`N CSDPs
`
`v
`ANALYZE
`CALCULATED
`N906
`CSPDS AND
`IDENTIFY LONGEST
`CSPD
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`LCSPD
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`FIG. 9
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`CONvOLvE LCSPD,
`CSPCD and TCD TO
`CALCULATE THE N912
`DURATION OF THE
`CYCLIC PREFIX
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`DURATION IN
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`DURATION
`
`918
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`I
`
`STOP
`CYCLIC PREFIX DURATION
`CALCULATION ROUTINE
`
`AT&T, EXH. 1018, Page 8 of 16
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`

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`U.S. Patent
`
`Jan. 10, 2006
`
`Sheet 8 of 9
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`US 6,985,433 B1
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`AT&T, EXH. 1018, Page 9 of 16
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`AT&T, EXH. 1018, Page 10 of 16
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`

`
`US 6,985,433 B1
`
`1
`METHODS AND APPARATUS FOR
`DETERMINING MINIMUM CYCLICPREFIX
`DURATIONS
`
`RELATED APPLICATIONS
`
`This application claims the bene?t of US. Provisional
`Application Ser. No. 60/233,000 ?led Sep. 15, 2000.
`
`FIELD OF THE INVENTION
`
`The present invention is directed to methods and appa
`ratus for communicating information and, more particularly,
`to methods and apparatus for generating and transmitting
`frequency division multiplexed signals.
`
`10
`
`15
`
`BACKGROUND
`
`In Frequency Division Multiplexing (FDM) communica
`tion systems, the available spectral bandWidth W is divided
`into a number of spaced sub-carriers, f1, .
`.
`. , fN, Which are
`used to transmit information. Speci?cally, information bits
`are ?rst mapped to complex FDM symbols B1, .
`.
`. , BN. The
`signal to be transmitted, S(t), is constructed by individually
`modulating those symbols onto the sub-carriers over an
`FDM symbol duration, that is,
`
`20
`
`25
`
`Where lBkl and Elk are the amplitude and the phase of complex
`symbol Bk, respectively, and t is the time variable. Orthogo
`nal Frequency Division Multiplexing (OFDM) is one par
`ticular example of FDM.
`FIG. 1 illustrates a knoWn system 100 for generating and
`transmitting an OFDM signal S(t). In the knoWn system 100,
`a digital signal processor (DSP) 112, generates a sequence of
`baseband discrete complex samples of S(t), Which are then
`converted to an analog continuous signal through use of a
`digital-to-analog converter 114. The analog signal generated
`by the D/A converter 114 is passed through a loW-pass ?lter
`(LPF) 115, mixed to the carrier frequency by mixer 116,
`ampli?ed With a poWer ampli?er 118, and ?nally transmitted
`over the communication channel 120.
`In the knoWn system, information to be transmitted on
`sub-carriers is combined in the digital domain so that by the
`time digital to analog conversion occurs distinct sub-carrier
`symbols do not exist, e.g., separate symbols corresponding
`to different sub-carriers are not available to be subject to
`separate and distinct digital to analog conversion operations
`and/or separate analog signal processing operations.
`One major draWback of the knoWn OFDM signal genera
`tion technique is the high peak-to-average ratio of the
`transmitted signal to be ampli?ed. Loosely speaking, the
`peak-to-average ratio is the ratio of the maximum and the
`average poWers of a signal. In general, the signal reception
`capability depends on the average poWer of the signal.
`HoWever, to avoid nonlinear distortion such as signal clip
`ping, the poWer ampli?er at the transmitter normally has to
`operate linearly across the full dynamic signal range of the
`generated signal. This usually requires use of a class A
`poWer ampli?er. As a result of the linear nature of the poWer
`60
`ampli?er 118, the poWer consumption of the poWer ampli?er
`mainly depends on the maximum transmission poWer.
`Hence, the peak-to-average ratio is an important measure of
`poWer consumption given the quality requirement of signal
`reception.
`In the OFDM system 100, the analog signal to be ampli
`?ed is the sum of many sinusoid Waveforms, e.g., sub-carrier
`
`2
`. , BN are
`.
`signal. Assuming complex OFDM symbols B1, .
`independent random variables, the analog signal at a given
`time instant Will tend to be a Gaussian distributed random
`variable, Which is Well recogniZed to have a large peak-to
`average ratio. Hence, the transmission of the OFDM signals
`generally consumes a signi?cant amount of poWer, Which is
`very undesirable, e.g., for mobile transmitters using battery
`as poWer supply. Various methods have been proposed to
`reduce the peak-to-average ratio of the OFDM signals. The
`basic ideas in these methods is to arrange complex symbols
`B1, .
`.
`.
`, BN appropriately to minimiZe the peak to average
`ratio. HoWever, in such methods, the fundamental structure
`of signal transmission of combining sub-carrier signals ?rst
`and then poWer amplifying the combined signal is normally
`the same as shoWn in FIG. 1.
`Thus, in the existing methods, sub-carrier signals are ?rst
`combined in the digital domain and then poWer ampli?ed.
`This tends to result in large poWer consumption as the
`combined signals in general do not have a good, e.g., loW,
`peak-to-average poWer ratio. In vieW of the above discus
`sion, there is a need for improved frequency division mul
`tiplexed signal generation and transmission techniques
`Which alloW for loWer peak-to-average poWer ratios and
`therefore improved energy ef?ciency during poWer ampli
`?cation stages of signal generation. It is desirable that at
`least some of the neW methods and apparatus be suitable for
`use With OFDM signals.
`
`BRIEF DESCRIPTION OF THE FIGURES
`
`FIG. 1 illustrates a knoWn system for generating and
`transmitting OFDM signals.
`FIG. 2 illustrates a system for generating and transmitting
`signals implemented in accordance With the present inven
`tion.
`FIGS. 3A and 3B illustrate sinusoidal signal generators
`Which can be used to generate an analog passband signal
`from DIGITAL OFDM symbols input thereto.
`FIG. 4 illustrates a poWer ampli?cation module imple
`mented in accordance With one embodiment of the present
`invention.
`FIG. 5 illustrates a poWer ampli?cation module imple
`mented in accordance With another embodiment of the
`present invention.
`FIG. 6 illustrates a multi-stage combiner circuit imple
`mented in accordance With the present invention.
`FIG. 7 illustrates an additional system for generating and
`transmitting OFDM signals implemented in accordance With
`an embodiment of the present invention Which uses ?lters in
`the signal transmission path.
`FIG. 8 illustrates a computer system Which can be used
`for calculating cyclic pre?x duration values in accordance
`With the present invention.
`FIG. 9 illustrates an exemplary cyclic pre?x duration
`calculation routine.
`FIG. 10 illustrates a communications system Wherein
`poWer ampli?cation of the transmission signal is performed
`in addition to poWer ampli?cation of individual subcarrier
`signals.
`FIG. 11 illustrates another embodiment of the communi
`cations system of the present invention With circuitry for
`generating cyclic pre?xes shoWn.
`
`SUMMARY OF THE INVENTION
`
`In accordance With the present invention, poWer con
`sumption associated With generating and transmitting fre
`
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`55
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`AT&T, EXH. 1018, Page 11 of 16
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`

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`US 6,985,433 B1
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`10
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`15
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`25
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`35
`
`3
`quency division multiplexed signals, e.g., OFDM signals, is
`reduced as compared to the known system discussed above.
`These results are achieved by performing poWer ampli
`?cation on analog sub-carrier signals on an individual basis
`prior to combining them to form an OFDM signal to be
`transmitted. As the sub-carrier signals tend to be substan
`tially in the form of sinusoid Waveforms, the ef?ciency of
`poWer ampli?cation of the individual sub-carrier signals Will
`generally be higher than that of performing poWer ampli?
`cation on a combined OFDM signal.
`While the present invention is described throughout the
`present application in the context of various exemplary
`OFDM embodiments, it is to be understood that the methods
`and apparatus of the present invention are applicable to a
`Wide range of FDM communications systems and are not
`limited solely to OFDM applications.
`In accordance With the present invention, at each OFDM
`symbol duration, a sinusoid signal, e.g., analog signal, is
`generated for each sub-carrier, With the phase and the
`amplitude of the sub-carrier being set by a complex OFDM
`symbol at the beginning of the OFDM symbol duration. In
`one embodiment, the sinusoid signals are bandpass, Wherein
`the signals can be ?rst generated in the baseband and then
`mixed to the carrier frequency or alternatively can be
`generated directly in the bandpass. In another embodiment,
`the sinusoid signals are baseband, Wherein the signals are to
`be mixed to the carrier frequency in a later stage. In one
`embodiment of the invention, the sinusoid signal is gener
`ated by an analog signal generator, Where the phase and the
`amplitude of the sub-carrier Waveform are set by an OFDM
`sub-carrier symbol. In another embodiment of the invention,
`individual sub-carrier sinusoid signals are generated from
`separate sequences of digital samples passing through digi
`tal-to-analog convertor devices.
`The complex OFDM symbol transmitted on each sub
`carrier is generated by a digital device and is to convey
`information bits to be communicated. In one embodiment of
`the invention, the OFDM symbols corresponding to a sub
`carrier at different symbol durations are preferably of con
`stant or near constant amplitude, thereby leading to a con
`stant-amplitude phase-modulated sinusoid signal for each
`sub-carrier. In such an embodiment, the ampli?ers used for
`individual sub-carrier signals may have a ?xed gain. HoW
`ever, even When ?xed gain ampli?ers are used, the gain
`applied by different ampli?ers corresponding to different
`sub-carriers may differ from one another.
`According to the invention, the sinusoid signals repre
`senting the sub-carrier signals are poWer ampli?ed by using
`linear and/or nonlinear stages individually and, in most
`cases, in parallel. In one embodiment of the invention, Where
`a constant-amplitude phase-modulated sinusoid signal is
`generated for each sub-carrier, the poWer ampli?cation is
`done With high-efficiency nonlinear poWer ampli?ers such
`as class C poWer ampli?ers, or done With linear poWer
`ampli?ers of small peak setting. Acombination of linear and
`non-linear poWer ampli?ers may be used for individual
`sub-carrier signals if desired.
`According to the invention, the poWer ampli?ed sub
`carrier signals are added using one or multiple stages of
`analog combining devices. Analog multiplexers are
`examples of combiner circuits suitable for use in combining
`the poWer ampli?ed signals in accordance With the present
`invention.
`The individual sub-carrier signals as Well as the transmit
`ted signal (TS), generated by combining the sub-carriers
`signals, may be passed through ?lters to limit out-of-band
`spectral emissions. According to the invention, one or more
`
`40
`
`45
`
`55
`
`65
`
`4
`?lters are put in various places in the sub-carrier paths and/or
`in the transmission signal path. Suitable locations for such
`?lters include after the combining devices used to generate
`the transmission signal, betWeen the combining stages, and
`in the individual sub-carrier signal paths. In the individual
`sub-carrier signal paths ?lters may be placed e.g., after the
`poWer ampli?ers of individual sub-carrier, before the com
`bining devices, and/or before poWer ampli?cation.
`To facilitate signal reception at the receiver, according to
`the invention, the cyclic pre?x added in the transmitted
`OFDM signal should effectively cover the majority of the
`transient responses due to various transmit components. To
`insure that cyclic pre?xes are of the proper duration and thus
`length, in accordance With one feature of the present inven
`tion signal delays, e.g., group delays, in the signal paths
`Which are traversed are determined, e.g., through the use of
`a computer system. Signal delays in this context may include
`transient responses introduced by various components, e.g.,
`?lters and/or ampli?ers. In this manner, signal delays such
`as those introduced by setting the phase and amplitude of
`each sub-carrier and ?ltering, in addition to the dynamic
`response introduced by the communication channel are
`taken into consideration When determining the duration of
`the cyclic pre?x.
`
`DETAILED DESCRIPTION OF THE
`INVENTION
`
`FIG. 2 illustrates an exemplary frequency division mul
`tiplexer signal generation and transmission system capable
`of generating and transmitting OFDM signals, implemented
`in accordance With one exemplary embodiment of the
`present invention. As illustrated in FIG. 2, information bits
`to be transmitted on various sub-carriers are ?rst mapped to
`complex OFDM symbols B1, .
`.
`. , BN, e.g., one symbol per
`sub-carrier for each symbol period, by a digital symbol
`generator (DSG) 202. According to the invention, each
`OFDM symbol Bk (Where 1<k<N) is then modulated to a
`corresponding sub-carrier fk using a corresponding sinusoi
`dal signal generator 204, 204‘ of signal generator module
`203, thereby generating an analog sinusoid signal for one
`symbol duration for each sub-carrier. The symbol duration is
`equal to the inverse of the spacing betWeen tWo adjacent
`sub-carriers, plus the duration of a cyclic pre?x portion to be
`discussed beloW. Each complex OFDM symbol to be trans
`mitted is used to convey information bits to be communi
`cated. In one embodiment of the invention, the OFDM
`symbols corresponding to each sub-carrier at different sym
`bol durations are of constant or near constant amplitude,
`thereby leading to a constant or near constant-amplitude
`phase-modulated sinusoid signal for each sub-carrier. In
`such a case, the amplitude of different sub-carrier signals
`may differ With the maximum amplitude of a particular
`sub-carrier remaining constant or nearly constant over time.
`According to the invention, the signals (SS1—SSN) of all
`the sub-carriers are poWer ampli?ed individually. In several
`embodiments the ampli?cation of individual sub-carrier
`signals is performed in parallel, e. g., by poWer ampli?cation
`module 205. The poWer ampli?cation module 205 includes
`N poWer ampli?cation circuits 206 thru 206‘, one for each of
`the N sinusoidal sub-carrier signals (SS1 thru SSN). In other
`embodiments sub-carrier circuitry is used on a time shared
`basis With ampli?ed sub-carrier signals being buffered While
`the ampli?cation circuitry is reused to amplify another
`sub-carrier signal. It is also possible in some embodiments
`to use sub-carrier circuitry for some sub-carriers on a time
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`shared basis While sub-carrier circuitry for other sub-carriers
`is not reused on a time shared basis.
`The power ampli?cation is performed using linear and/or
`nonlinear stages, e.g., one or more poWer ampli?cation
`circuits per sub-carrier signal. Because the signal of each
`sub-carrier is substantially a sinusoid Waveform, and in the
`exemplary embodiment is of constant or near constant
`amplitude, high efficiency poWer ampli?cation devices may
`be used as ampli?cation circuits 206, 206‘. As Will be
`discussed further beloW, in one particular embodiment,
`poWer ampli?cation is done With nonlinear poWer ampli?
`ers, e.g., high-ef?ciency class C poWer ampli?ers. In another
`embodiment poWer ampli?cation is done With linear poWer
`ampli?ers, Wherein the peak to be handled by the poWer
`ampli?er can be set small due to the constant or near
`constant signal amplitude thereby minimiZing poWer con
`sumption. Various combinations of linear and non-linear
`poWer ampli?ers is also possible.
`According to the invention, the analog poWer ampli?ed
`sub-carrier signals (PAS1—PASN) are added by one or more
`combining devices, e.g., analog multiplexers, Which are
`used to implement combiner circuit 208. The combined
`signal TS generated by combiner circuit 208, is transmitted
`over the communication channel 210.
`In order to control the out-of-band spectral emission of
`the transmitted signal, according to the invention, as Will be
`discussed in detail beloW, ?lters maybe used at various
`places in the signal processing path shoWn in FIG. 2.
`Various components in the communication system 200,
`such as the signal generators 204, 204‘ used to generate the
`sinusoid signals SS1—SSN and ?ltering circuits to be dis
`cussed beloW, can introduce transient responses into the
`transmitted signal TS. Those transient responses can con
`volve With the dynamic response introduced by the com
`munication channel 210 When the signal reaches a receiver.
`In order to facilitate signal reception at the receiver, the
`length of the cyclic pre?x is chosen, in accordance With one
`feature of the present invention, to cover the majority of the
`combined transient and dynamic responses.
`FIGS. 3A and 3B illustrated sinusoid signal generators
`304, 305 for generating an analog passband sinusoid signal
`for a single sub-carrier, k Which may be any one of the N
`sub-carriers. In the FIG. 2 system, for every symbol dura
`tion, a sinusoid signal is generated for each sub-carrier,
`Where the phase and the amplitude are given by the OFDM
`45
`symbol that is to be conveyed on that sub-carrier. The
`sinusoid signal generators 304, 305 may be used as any one
`of the generators 204, 204‘ illustrated in FIG. 2.
`The sinusoid signals generated by the generators 304, 305
`are bandpass. These signals can be generated directly in the
`bandpass as in the FIG. 3A embodiment or can ?rst be
`generated in the baseband and then mixed to the carrier
`frequency as in the FIG. 3B embodiment. Alternatively, the
`sinusoid signals can be output as baseband signals, Wherein
`the signals are to be mixed to the carrier frequency in a later
`stage, that is, after poWer ampli?cation.
`The signal generator module 304 illustrated in FIG. 3A
`includes a sWitching element 307 for extracting the values
`lBkl and Elk Which are then used by the sinusoid signal
`generator circuit 308 to generated the bandpass signal lBklcos
`(2J'cfkt+0k). N of the signal generators 304 may be used to
`implement the signal generator module 203.
`Similar to that shoWn in FIG. 3A, a bandpass signal
`|Bk|cos(2rcfkt+0k) can be generated as shoWn in FIG. 3B from
`a pair of baseband sinusoid signals |Bk|cos(2rc(fk—fC)t+0k)
`and |Bk|sin(2rc(fk—fC)t+0k), Which are mixed to the carrier
`frequency by mixer 312 to generate the passband signal
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`|Bk|cos(2rcfkt+0k). The signal generator 305 includes sWitch
`ing element 307 for extracting the values lBkl and Elk Which
`are then used by the cosine signal generator circuit 310 and
`sinusoid signal generator circuit 312 to generate the base
`band signals |Bk|cos(2rc(fk—fC)t+0k) and |Bk|sin(2rc(fk—fC)
`t+0k), respectively.
`The phase and the amplitude of the Waveforms are set by
`the OFDM symbol at the beginning of the OFDM symbol
`duration. From one symbol duration to another, the baseband
`sinusoid signals are generated from different sets of OFDM
`symbols. In the cases Where the OFDM symbols for a
`sub-carrier have the constant amplitude, for example, When
`the OFDM symbols are generated With phase-modulation
`methods, only the phase is set from one OFDM symbol
`duration to another, thereby resulting in a constant-ampli
`tude phase-modulated sinusoid signal for each sub-carrier.
`In FIG. 3A, each digitally generated OFDM symbol is
`supplied to sWitch device 307 Which controls the phase and
`the amplitude of the corresponding sinusoid or cosine signal
`generator 308, 310, 312 at the beginning of the OFDM
`symbol duration from a given constellation set, and main
`tains the values for the entire symbol duration. The sWitch
`device 307 operates at the OFDM symbol rate and generates
`discrete outputs Whose range is determined by the constel
`lation siZe of the OFDM symbols.
`In reality, the signal generators 304, 305 are not be able
`to change the phase and the amplitude instantly. Instead, a
`transient period exists at the beginning of the OFDM symbol
`duration, during Which the actual signal generated by the
`signal generator 304, 305 is not a constant-amplitude phase
`modulated sinusoid as desired. In one embodiment to elimi
`nate or minimize any resultant adverse impact, the cyclic
`pre?x added to the OFDM symbol duration is made as long
`or longer than the transient period.
`In another embodiment of the invention, not shoWn in
`FIG. 3, a digital device, such as a digital signal processor, is
`used to generate a sequence of discrete samples of the
`sinusoid signal from the OFDM symbol. Those discrete
`signal samples are passed through a D/A device to generate
`the required sinusoid Waveform.
`FIG. 4 illustrates a signal poWer ampli?cation module 400
`implemented in accordance With one embodiment of the
`present invention. The ampli?cation module 400 may be
`used in place of the poWer ampli?cation module 205 in the
`system 200.
`As discussed above, according to the invention, the sinu
`soid signals of the sub-carriers are poWer ampli?ed by
`means of linear and/or nonlinear stages individually and in
`parallel prior to being combined.
`In the cases Where a constant or near constant-amplitude
`phase-modulated sinusoid signal is generated for each sub
`carrier, the poWer ampli?cation is done With high-ef?ciency,
`poWer ampli?ers Which may be non-linear. In one version of
`the FIG. 4 embodiment, poWer ampli?ers 1 thru N, 402, 402‘
`are non-linear class C poWer ampli?ers. In both the FIG. 2
`and FIG. 4 embodiments, one poWer ampli?er is used for the
`sinusoid signal of each sub-carrier. In the case of using
`nonlinear poWer ampli?ers as in the FIG. 4 example, the
`output of each poWer ampli?er 402, 402‘ may include
`high-order harmonics. In the FIG. 4 embodiment, ?lters 404,
`404‘ are included after each poWer ampli?er 402, 402‘ to
`eliminate or reduce high-order harmonics from the ampli?ed
`sub-carrier signals.
`In the alternative embodiment illustrated in FIG. 5, linear
`poWer ampli?ers 502, 502‘, one per sub-carrier, are used to
`implement a poWer ampli?cation module 500 Which may be
`used in place of the module 205 illustrated in FIG. 2. In the
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`FIG. 5 embodiment, the peak to be handled by the power
`ampli?er can be set to be small to reduce poWer consump
`tion, since the input sinusoid signal representing one of the
`sub-carrier signals (SS1—SSN) should have a good peak-to
`average poWer ratio.
`FIG. 6 illustrates a combiner circuit 600 Which may be
`used in place of the combiner circuit 208 illustrated in FIG.
`2. In accordance With the present invention, the outputs of
`the poWer ampli?ers representing poWer ampli?ed signals
`(PAS1—PASN) for each sub-carrier are added together. This
`may be done using a single combining circuit (summer) or
`using multiple combining circuits, each With a relatively
`small number of inputs, arranged in stages. The combiner
`circuit 600 illustrated in FIG. 6 incorporates tWo states 602,
`606 although more stages are possible. In the ?rst stage 602,
`small subsets of the poWer ampli?er output signals PAS are
`?rst added together to produce intermediate signals. For
`example signals PAS 1 trough PASX are combined by the ?rst
`combiner 1 604 to form a ?rst intermediate signal. The
`signals PASY through PASN are combined by combiner Y
`604‘ to form another intermediate signal. Additional com
`biners in the ?rst stage may also produce intermediate
`signals. The intermediate signals generated by the ?rst sated
`602 are then combined in the second stage by combiner 608
`to form the combiner circuit output signal TS.
`As discussed above, ?lters may be used at various points
`during the processing of signals by the system 200, e.g., to
`control out-of-band spectral emissions illustrated in FIG. 2.
`FIG. 7 illustrates a system 700 Which is similar to the system
`200 but includes optional ?ltering circuitry. The FIG. 7 ?lter
`circuitry includes ?rst and second ?lter modules 702, 710,
`Which each include one ?lter for each of the N sub-carrier
`signals. In addition, the ?lter circuitry includes a ?lter 720
`Which is used to ?lter the signal TS generated by the
`combiner circuit 208 prior to transmission over the commu
`nications channel 210.
`As shoWn in FIG. 7, ?lters 704, 704‘ can be placed after
`the sinusoidal signal generators 204, 204‘, betWeen the
`poWer ampli?cation circuits 206, 206‘ and combiner circuit
`208, and after the combiner circuit 208. In addition, one or
`more ?lters can be placed betWeen combining stages, e.g.,
`betWeen stages 602, 606 in a multi-stage combiner circuit
`such as the one illustrated in FIG. 6.
`Filters elements in a communications path including the
`communications channel itself can introduce group delay
`Which is signal delay Which is dependent on frequency. As
`a result the different frequency components of a pulse
`launched into a communications path Will arive at the
`destination With slightly different delays. Both frequency
`dependent attenuation and the group delay cause dispersion
`on the transmission line, i.e., the spreading in time of a
`transmitted pulse.
`Thus, ?lters in the paths of subcarrier signals can intro
`duce group delays into the subcarrier signals While ?lters in
`the path of the combined signal TS can introduce group
`delays into the transmission signal TS. Additional group
`delays may be introduced into the signal TS by the com
`munications channel 210. As discussed above, the cyclic
`pre?x duration may, and in various embodiments of the
`present invention is, selected to be of suf?cient duration that
`it Will cover the group delays introduced by the ?lters in the
`subcarrier signal paths, the combined signal path and the
`transmission channel.
`FIG. 8 illustrates a computer system 800 Which