LOW NOISE MICROWAVE INTEGRATED RECEIVER FOR 64 QAM DIGITAL RADIO
`
`A. Giavarini/F. Marconi
`
`ABSTRACT
`
`64 QAM digital radio requires a very high linear operation not only in the
`transmitter section, but also in the receiver.
`The paper presents a low noise microwave integrated receiver able to cover
`each CCIR radio frequencies band from 4 to
`11 GHz with the linearity
`performances required for 64 QAM digital radio application.
`
`1.
`
`INTRODUCTION
`
`expenses
`
`High capacity digital radios with multilevel modulation methods require a
`large dynamic operating range, therefore an automatic gain control network
`used
`in
`be
`the microwave
`receiver
`in order to
`must
`section
`of the
`guarantee the required linearity performance even during strong up fading
`phaenomena as often observed in many trial systems.
`A low noise figure guarantees a high system gain value.
`instantaneous
`bandwidth
`The
`wide
`reduces
`the
`maintenance
`drastically decreasing the number of spare part units.
`integrated
`Microwave
`technology
`permits
`circuit
`the
`mechanical dimensions of the units.
`Substantially we can say that linearity performance,
`low noise figure,
`bandwidth and -mechanical
`instantaneous
`wide
`compactness
`are
`the main
`characteristics required for a modern microwave receivers line up.
`
`reduction
`
`of
`
`the
`
`2.
`
`MICROWAVE RECEIVER
`
`Figure I shows the microwave receiver block diagram: it consists of a low
`noise amplifier with an automatic gain control (AGC) network and an image
`rejection mixer followed by an intermediate frequency preamplifier.
`
`LOCAL OSCILLATOR
`
`>
`
`_x
`
`LOW NOISE
`PREAMPL.
`WITH AGC
`
`IMAGE
`REJECT.
`MIXER
`
`Fig. 1
`>bMijcrowave receiver
`diagram
`block
`
`IF PREAMP.
`
`2.1 Image Rejection Mixer
`
`To avoid the band-pass filter between preamplifier and mixer, an image
`rejection mixer has been designed: its block diagram is shown in figure 2.
`
`* GTE Telecomunicazioni S.p.A. - Cassina de' Pecchi - Milano - Italy
`
`168
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`
`

`

`Balanced mixer 1
`
`It consists of two balanced mixers and three hybrids: the first for the RF
`signal in quadrature, the second for the local oscillator in phase and the
`third for the IF signal in quadrature that combines the output of the two
`IF mixers.
`output
`the
`IF
`this
`way
`In
`response to the RF signal
`is
`present only at one of the IF
`IF
`while
`the
`hybrid
`outputs
`output response to the image
`signal is present at the other
`IF hybrid output.
`diodes
`Schottky
`barrier
`Low
`balanced
`for
`used
`been
`have
`mixers: a typical 10 dB noise
`achieved
`been
`has
`figure
`IF
`70 MHz
`dB
`including
`2
`a
`preamplifier noise figure.
`
`RF
`
`input
`
`fig. 2 - Image rejection mixer principle schema
`
`2.2 Low Noise Preamplifier
`With the purpose of achieving the required linearity performance together
`with the best noise figure of the complete receiver, a two stage low noise
`preamplifier with the automatic gain control has been designed.
`0.3 micron gate length GaAs FET devices have been used obtaining a typical
`noise figure of the first stage of 1.1 dB at 6 GHz and 1.9 dB at 11 GHz.
`or
`AGC microwave network can be placed in between the two FET stages,
`after them according to the block -diagram of figure 3 in which, for the
`two different frequency range of 6 and 11 GHz, gain and noise figure are
`given.
`
`NF2dB
`
`R'
`
`RFIN
`NF= 1,9dB
`
`OF N ~~~~,
`
`I' ~~~~
`
`6) O~~F OUT
`
`IsSTAGE LNA
`=1t dO
`G
`NF = 11 ddB
`IP .21dBm
`
`AUTOMATIC
`GAIN
`CONtROL
`2.5-d
`11.
`
`2ndSTAGEtNA
`G -13dB
`NF z dB
`P1dOm
`
`T~~~~~~~~~~~~~I-U
`
`>
`
`,
`
`(N
`
`FOUT
`
`(3a)
`
`OF IN
`
`NF2AdB
`
`RF IN
`NF.23dB
`
`RF OUT
`
`ATOMATIC Zd STAGE LNA
`ist AGE LNA.
`G =1OdB
`=9.5dB
`GAIN
`G
`N =-22dB
`CONTROL
`NF = 1,9dB
`IP =AdBm ILd1S-265dB
`IP =2BdBn
`
`~6)
`
`,
`
`FU
`
`(30)
`
`Fig. 3 - Preamplifier with AGC block diagram
`3 b): 11 GHz arrangements
`3 a): 6 GHz arrangements
`
`Figure 4 shows a third order intermodulation level versus input power for
`the two different solutions in the 6 and 11 GHz frequency range.
`It can be see that the best compromise between noise figure and third
`is to choose, for the 6 GHz, AGC network in
`order intermodulation level
`between the two FET stages and for the 11 GHz after them.
`
`169
`
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`
`

`

`AP[
`
`\
`
`22f.-fi i
`
`ft
`
`2fz-f,
`
`tiP
`(d 8)
`100
`
`80
`6 0
`40.
`
`20
`
`A P
`(d )3
`
`120t
`
`A
`
`100t
`804
`6 0-
`
`F
`
`2f.-t
`
`ft
`
`N.
`
`f1
`
`2f.-f,
`
`11 GHz
`
`-o -40 -30 -20 -10
`
`-P
`
`a
`
`Pi(dBm)
`
`6 GHz
`
`\
`
`40
`
`@
`
`f
`
`w
`
`@
`
`t
`
`'
`
`@
`
`-50 -40 -30 -20 -10
`
`~--
`
`Pi (dBm)
`
`Fig. 4 - Third Order Intermodulation Level Versus Input Power in the
`6 and 11 GHz Frequency Range
`dotted lines: with
`network between the two FET stages
`AGC
`continuous lines: with AGC network after the two FET stages
`
`-
`-
`
`2.3 Automatic Gain Control Network
`
`-
`
`The automatic gain control network consists of a parallel type PIN diodes
`attenuator in a balanced or single configuration as shown in figure 5.
`
`IN
`
`Ro
`
`I N
`
`IT
`
`OUT
`
`R:PIN
`
`DIODE RESISTANCE
`(b)
`
`R- PIN DIODE RESISTANCE
`(a)
`Fig. 5 - Balanced (a) and single (b) Configuration PIN diode attenuator
`
`Parallel
`configuration
`increasing the
`because,
`been
`has
`choosen
`type
`attenuation,
`PIN
`increases:
`diode
`minimizes
`current
`that
`the
`intermodulation distortion introduced by the diode itself at a high input
`power level.
`The circuit has been designed in order to obtain a high dynamic range
`attenuation with lower PIN diode current variation with respect to the
`commonly used PIN diode attenuators. This has been achieved by means of
`admittance step (*) in the transmission line when PIN diodes are located.
`Figure 6 shows a parallel type circuit schema.
`
`zt~~~
`I;ZoX/4
`
`-I
`
`l
`
`it
`X./4
`
`Zo R
`
`~~~
`
`rI
`
`REFLECiION COEFFICIENT
`
`Fig. 6
`
`Zo
`
`type
`Parallel
`PIN
`Attenuator Circuit Schema
`
`Diode
`
`(*)
`
`Italian patent application N.22923 A/85
`
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`
`

`

`It can be shown that the attenuation of this kind of circuit is given by:
`1
`Zt2 + Zo
`R
`
`(1 - Irl 2)
`
`0
`
`1
`+ Zo
`
`+ 1
`R
`
`Zt
`R
`
`I
`
`I + Zt2
`R. Zo
`
`A [dB] = 10 10g9o
`
`-
`
`+
`
`1
`Zt2 +Zo
`R
`
`1
`Zt2 +Zo
`R
`
`-1
`R
`
`Z Z
`
`o
`
`1 R
`
`+
`
`Zo
`
`with:
`
`where:
`
`PIN diode resistance
`t
`Transmission line impedance
`Input/Output characteristic impedance
`Zo
`attenuation versus Zt with Zo = 50 Q, parameter: PIN
`Figure 7 shows the
`diode resistance.
`
`Rz
`
`A(
`
`dB)
`
`Fig. 7
`
`Attenuation Versus
`Diode
`Resistance
`.values
`
`ZT for PIN
`different
`
`100
`
`ZT
`(n)
`
`3.
`
`MIC REALIZATION
`
`All the microwave circuits are microstrip designed. High purity alumina
`substrates 25 mils thick have been used and the circuits are manufactured
`in thin film technology with Ti, Pd, Au metal conductor system which is
`resistant to environmental corrosion, solderable, thermobondable, accepts
`resistors,
`and
`of tantalum nitride
`stabilization
`cycling. for
`thermal
`maintains good end-of-life conductivity.
`
`171
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`
`

`

`Requirements for precise fine lines and small spaces (50 pm) are presently
`well met by selective plating the gold directly onto the Pd barrier layer
`using positive photoresist sprayed on as thickly as the final gold.
`Figures 8 and 9 shown respectively 6 GHz on 11 GHz receiver pictures.
`
`Fig. 8 - 6 GHz Receiver
`
`Fig. 9 - 11 GHz Receiver
`
`4.
`
`OVERALL
`MICROWAVE
`PERFORMANCES
`
`RECEIVER
`
`EXPERIMENTAL
`
`RESULTS
`
`AND
`
`SYSTEM
`
`Figure 10 shows a measured overall microwave receiver noise figure for
`designed in different frequency ranges.
`Figure 11
`different receivers,
`shows third order intermodulation product levels measured at maximum input
`power level.
`
`NF (dB)
`
`:3
`
`22.5F
`
`RF-eVI13dBm
`
`I dB/Div
`
`ZMIHz/DIv
`
`r
`
`4
`
`6
`
`I
`6
`
`I
`
`10
`
`I-_
`12
`
`F(GHz)
`
`I
`
`_____
`
`_____
`
`Fig. 10 -
`
`Typical
`overall
`microwave
`receiver
`noise f;igure versus
`frequency
`
`Fig. 11 --
`
`Typical
`third
`i ntermodul ati on
`duct at maximum
`power level
`
`order
`pro-
`i nput
`
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`

`

`In figure 12 receiver gain variation versus input power level is shown.
`Lastly figure 13 shows BER as function of received signal level (RSL) when
`an AGC circuit is used in the RF section (continuous line) or in the IF
`It can be shown that an attenuator located in the
`section (dotted line).
`RF section gives an improvement of about 20 dB at maximum RSL for any BER
`threshold with respect to an attenuator located in the IF section.
`Therefore a higher operating dynamic range is obtained as required for 64
`QAM digital radio application.
`
`GAIN
`(dB)
`34
`
`0
`
`-I
`
`Fi 9. 12
`
`gain
`Receiver
`versus
`ti on
`power level
`
`varia-
`i nput
`
`-5
`
`-6
`
`-7
`
`-2
`
`-3
`
`-4
`
`-8
`Vagc (v )
`Il
`t
`I
`I
`-50 -45 -40 -35 -30 -25 -20 -15
`INPUT POWER LEVEL (dBu)
`
`Fi g. 13
`
`versus
`RSL
`BER
`in
`circuit
`IF
`line)
`dotted
`or
`section
`continuous
`
`with AGC
`section
`in
`RF
`line)
`
`BER
`io-.1
`ur-4-
`so4-
`io~-
`so'-.
`
`I
`
`I
`
`-30
`
`-20
`
`-10
`
`0
`
`RSL (dBm)
`
`5.
`
`ACKNOWLEDGEMENT
`
`The authors would like to thank Dr. R. Macchi and Dr. P. Bonato for their
`valuable collaboration and Mr. G. Mora for his help in design and in the
`experimental work.
`
`173
`
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`
`