IEEE TRANSACTIONS ON COMMUNICATIONS, VOL. COM-27, NO. 12, DECEMBER 1979
`
`1953
`
`Design and Performances of a 200 Mbit/s 16 QAM
`Digital Radio System
`
`IZUMI HORIKAWA. TAKEHIRO MURASE, AND YOICHI SAITO
`
`Abstract—A novel long haul 5 GHz 16 QAM digital radio system, which
`has 200 Mbit/s transmission capacity within the 40 MHz interleaved
`channel allocation. is proposed and described. It is designed to be overbuilt
`on existing FDM-FM routes with an approximately 50 km repeater
`spacing. To achieve the 5 bit/s/Hz RF spectral efficiency, the 16 QAM
`modulation and Nyquist cosine roll-off spectral shaping techniques
`( on = 0.5) are investigated. Then a new signal shaping filter, differential
`encoding and carrier recovery techniques are presented. Finally. the effects
`of TWT amplifier nonlinearity on a 16 QAM signal are experimentally
`investigated.
`
`1. INTRODUCTION
`
`“
`
`IGITAL Radio” is an attractive transmission system in
`
`digitalizing a communication network. In designing a
`microwave digital ratio system as a large capacity trunk trans-
`mission system, particular consideration should be taken of
`the efficient use of the radio frequency spectrum, and 0f the
`compatibility with existing analog FM systems, such as trans-
`mission capacity, frequency allocation. repeater spacing, re-
`peater station facilities, etc. Several digital radio systems have
`already been developed or are now being investigated in the
`microwave frequency band [l]-[6]. They employ the 4, 8
`PSK or QPRS modulation techniques with around 4 bit/s/Hz
`RF spectral efficiency. Transmission capacities are less than
`100 Mbits/s, except for systems in 18-20 6112 bands [5] , [6] .
`In this paper, a novel
`long haul 5 GHZ 16 QAM digital
`radio‘ system, which has 200 Mbit/s transmission capacity
`within the 40 MHz interleaved channel allocation, is proposed
`and its system design considerations as well as the hardware
`performances are described. This system is designed to be
`overbuilt on existing FDM-FM routes with approximately
`50 km repeater spacing.
`To achieve the 5 bit/s/l-lz RF spectral efficiency,
`following three important tasks have to be completed.
`(1) The developing of a 200 Mbit/s capacity system design
`within the 40 MHZ bandwidth.
`
`the
`
`(2) The developing of 16 QAM modulation/demodulation
`and Nyquist cosine roll-off spectral shaping techniques.
`(3) The investigation of multipath fading effects on a multi-
`level modulated signal and of its countermeasure techniques.
`Specifically,
`the following problems are discussed, which
`are essential to the realization of this high capacity, high effi-
`ciency system.
`
`Manuscript received January 16, 1979; revised June 29, 1979. This
`paper was presented at the International Conference on Communica-
`tions, Boston, MA, June 1979.
`The authors are with the Yokosuka Electrical Communication Lab-
`oratory, Nippon Telegraph and Telephone Public Corporation, Yoko—
`suka-shi, Kanagawa 238-03, Japan.
`
`(1) Modulation technique. (2) Simple differential encoding
`for multilevel QAM systems. (3) New binary transversal filters.
`(4) Carrier recovery with selective gated PLL. Finally,
`the
`effects of TWT amplifier nonlinearity on a 16 QAM signal are
`reported.
`
`11. SYSTEM DESIGN
`
`A. Modulation Technique
`
`To obtain competitive transmission capacity with large
`capacity FM systems, a 200 Mbit/s bit rate is needed for the
`digital systems, which is equivalent to 2880 voice channels.
`This implies that a highly spectral efficient modulation tech-
`nique, higher than 2.5 bits/Hz, would have to be developed.
`This efficiency can be achieved by multilevel modulation
`with the Nyquist spectral shaping technique. The conventional
`8 PSK with 76 Mbaud symbol rate and rolloff factor a of 0.2
`might be one of the possible modulating schemes. However,
`severe Nyquist transmission, such as oz = 0.2. is quite sensitive
`to many kinds of distortions. Besides,
`the power spillover
`into the out of guard band in the existing analog system fre-
`quency allocations exceeds
`1% of
`the total
`transmission
`power. These considerations lead to the 16 level modulation
`schemes, or = 0.5.
`The QAM signal constellation is very close to the best per-
`formance achievable by the optimum packing of the 16 signal
`points into two dimensions [7], [8]. Also, the QAM tech-
`nique is quite suitable for a 200 Mbit/s digital microwave
`system, taking into account the easy implementation of high
`speed MODEM circuitry.
`
`B. System Configuration
`
`Figure 1 and Table 1 show a functional block diagram and
`system parameters for the 200 Mbit/s 16 QAM digital radio
`system. The system is composed of three functional blocks:
`terminal interface, 16 QAM MODEM and radio equipment.
`The transmitting terminal
`interface translates two 100
`Mbit/s AMI signals into four 50 Mbit/s NRZ signals. In addi-
`tion, the terminal interface provides redundunt pulse insertion
`for monitoring errors, scrambling and differential encoding.
`The
`16 QAM MODEM is
`regenerative equipment
`that
`carries 200 Mbits/s in a 40 MHz bandwidth at 140 MHz carrier
`frequency. In order to conserve the 200 Mbit/s signal within
`40 MHz, the 16 QAM modulation and the Nyquist channel
`(or = 0.5) techniques are used. The transmitting and receiving
`filters are adopted at
`the baseband, since two dimensional
`modulation systems, such as 8 PSK, QPRS or 16 QAM, are
`sensitive to asymmetrical transmission responses, and since it
`is difficult to realize the accurate required filter at IF or RF.
`
`0090-6778/79/1200~1953$00.75 © 1979 IEEE
`
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`
`

`

`1954
`
`IEEE TRANSACTIONS ON COMMUNICATIONS, VOL, COM-27, NO. 12, DECEMBER 1979
`
`
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`Fig.
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`1 Block diagram of 16 QAM system.
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` e M
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`l-l
`
`TABLE I
`
`200 Mbit/s 16 QAM DIGITAL RADIO SYSTEM PARAMETERS
`
`Fig. 2 Relation 0n 2M X 2’” rectangular signal set between 2’" X 2’" level
`converter output signals and 4 PSK signals.
`
`Frequency band
`
`4.4 ~ 5.0 GHz
`
`Capacity
`Modulation
`
`200 Mbit/s.sys
`16 QAM
`
`Demodulation
`
`Coherent detection
`
`Symbol rate
`
`50 MB
`
`RF channel spacing 40 MHz
`
`Repeater spacing
`
`50 km
`
`Regeneration
`
`Every repeater
`
`the same functions as
`The radio equipment has almost
`those of high capacity FM systems. In addition, the receiver is
`equipped with an equal-gain in-phase combiner and a dynamic
`equalizer to improve frequency distortion caused by selective
`fading.
`
`III. CIRCUIT DESCRIPTION
`
`A. Differential Encoding for Multiple QAM systems
`
`A problem arises at the demodulator in the practical hard-
`ware design phase. This results from the fact that the demodu-
`lator can recognize the pattern of signal points but cannot
`distinguish between the various symmetric phase orientations
`of the signal set. This ambiguity has to be resolved in some
`way. The use of differential encoding has the advantages of
`no redundunt information insertion and easy implementation.
`The rectangular signal sets in two-dimension are, generally,
`constructed from two 2 -2M level converters and a quadrature
`amplitude modulator. Another new technique, which gener-
`ates the signal sets by summing the outputs of two multiphase
`modulators, has been recently proposed [9]. Generalizing
`these two generating techniques and considering the relations
`between the two methods, a simple differential encoding for
`multilevel signal sets can be derived. Figure 2 illustrates the
`relation between the QAM technique and the new “superim-
`posed modulation” technique for 2M X 2M rectangular signal
`sets. As illustrated in Fig. 2, the pair of i-th figure digital sig-
`
`nals, at the 2 — 2M level converter Zi—ibxi and 2i—1byi, is
`equivalent to the 4PSK signal (4PSK)i—1.
`From these discussions,
`it becomes clear
`
`the i-th
`
`that
`
`figure signals of D/A converter output are the same as the
`figure signals of the 2M -leve1 decision circuit at the demodu-
`lator, and also that each pair of the same figure signals forms
`individual 4 PSK signals.
`Therefore, the differential encoder and decoder for 2’" X
`2’" QAM systems can be implemented withM pairs of module-
`4 summing and subtracting logic circuits by processing the
`latter with their i-th figure signals in pairs of D/A and A/D .
`converter output, as shown in Fig. 3. These differential encod-
`ing and decoding logic equations are shown Eq. (I).
`
`Encoder
`
`Yin(bxin ’ byin) : Xinmxin a ayin)
`
`® Yin—Kbxin—‘l , byin‘l) mod _ 4
`
`(1)
`
`Decoder
`
`Xi"(axi" : ayi") = Zin(cxin: Cyr")
`
`©Zi"_l(cxi"_l,cyi"_l) mod "_4
`
`where Xi"(ax,-", ayi"), Yi"(bx,-", byi") andZi"(cxi",cy,-")ex-
`press their quaternary numbers, which consist of each pair of
`binary signals, axi”, ayi" , etc.
`Using this coding procedure along with thelencoding for
`PSK systems,
`the bit error is increased twofold. This corre—
`sponds to 0.3 dB C/N degradation. However, the difference in
`C/N degradation between this procedure and another possible
`procedure [10] is only 0.1 dB.
`Thus,
`the differential encoding procedure for multilevel
`QAM signals is quite simple, and the implementation of the
`encoding scheme can easily be constructed by adopting the
`same logic circuitry.
`
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`
`

`

`HORIKAWA et al.: 200 Mbit/s 16 QAM DIGITAL RADIO SYSTEM
`
`1955
`
`(2)
`
`(3),
`
`.
`(4)
`
`(5)
`
`CLQEK
`NRZ
`
`
`
`N-STAGE
`SHIFT REGISTER
`-- /
`
`A(f)=ZSQ‘) 2 [cos|:2k+lrrfT:|]
`
`
`
`2K
`
`'K
`
`k=,1
`
`=50) 'H2K(f)-,
`
`And,forL = 2K+ 1,
`
`_
`
`a(t)—s(t)+k§1ls<t
`
`K
`
`_
`
`
`t— 2"
`2K+i
`
`+
`
`s
`
`T
`
`
`2k
`
`2K+1
`
`
`I
`2
`
`
`
`(b) DIFFERENTIAL DECODING
`
`Fig. 3 Differential encoding for 2’" X 2’" QAM.
`
`A(f)=S(f) 1+2 f
`k=1 cos
`
`
`2k
`2K+ l1rfT
`
`B. Binary Transversal Filters
`
`= 5U) ' H2K+1(f)
`
`The transmitting spectral shaping filter controls the overall
`
`transmission characteristic for the Nyquist channel. Binary
`transversal filters (BTF’s), are effective for accurate spectral
`shaping, since the BTF’s directly generate the required wave-
`form through the design procedure of the time domain. This
`section describes the new design principle for the binary trans-
`versal filter (BTF).
`A conventional BTF [11], [12] processing a return-to-zero
`type pulse stream has to be equipped with an NRZ-RZ con-
`verter. As a result, the timing margin decreases, andthe dis-
`crete frequency component at the symbol rate increases.
`' This new BTF accepts an NRZ pulse stream without an
`RZ signal processing stage, as shown in Fig. 4. NRZ pulses of
`width “T,” generally, can be considered as L RZ pulses of
`width “T/L”, superposed in the time difference of T/L, where
`L is an integer.
`
`This means that this new BTF is equivalent to combining
`the L conventional BTF with the time difference of T/L. From
`these facts, the following equations are given in both time and
`frequency domain.
`ForL = 2K
`
`l‘ = E
`
`K
`
`
`
`2k+1
`
`g
`
`‘
`
`._ +
`
`T
`
`
`
`
`
`where s(t) is the conventional BTF output signal and S0) is
`its Fourier transform. Figure 5 illustrates these relations for
`L = 2. Thus,
`the new BTF, which accepts an NRZ pulse
`stream, has the same effect as the conventional BTF’s trans‘
`mission function H2k(f), or H2k+1(f). Therefore, with this
`new BTF,
`the eXpected transmitting spectral S(f) can be
`designed by substituting Sm/HZkU) or, S(f)/H2k+1(f) for
`50‘) in the conventional BTF design method.
`I
`The number of stages of shift register is determined by con-
`sidering the residual
`intersymbol interference, out of band
`energy and circuit size. The eleven stage shift registers are
`selected and three MECL 10 K IC’s are driven by a 100 MHZ
`clock signal. Figure 6 shows the theoretical and experimental
`BTF output spectral density for or = 0.5. The new BTF gener-
`ates no discrete symbol frequency component, and effectively
`reduces the residual out of band energy in the f/fp =_ 1.0
`vicinity. The measured spectral performance agrees exactly
`with the theoretical value. The out of band spectrum was
`reduced by fifth order Chebyshev LPF of 60 MHz cutoff
`(0.01 dB ripple). Figure 7 shows eye diagrams of the BTF,
`comparing it with conventional and new type BTF’s. At the
`LPF output, an accurate Nyquist pulse waveform with less
`than 3% of intersymbol interference is achieved, and is shown
`in Fig. 8.
`
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`Page 3
`
`

`

`1956
`
`IEEE TRANSACTIONS ON COMMUNICATIONS, VOL. COM-27, NO. 12, DECEMBER 1979
`
`sun/4)
`
`-3T/4
`4,40
`
`7/4
`
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`v
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`Zl/T 4R/T'
`
`Fig. 5 Waveform and spectral function for new BTF; L = 2.
`
`
`
`1
`(p) New BTF '
`Fig. 7 Eye diagrams of th BTF comparing with conventional and new type
`BTF‘s.
`
`
`
`Fig. 8
`
`4-level eye diagram for BTF followed, by an LPF. horizontal 5 ns/div.
`
`ance. A novel carrier recovery with a selective gated phase-
`locked loop (PLL) has been developed for this system. Figure
`9 shows a functional block diagram of the carrier recovery
`circuit for 16 QAM signal.
`»
`Half of the 16 QAM signal points is different from the 4
`PSK signal phases. This causes undesirable false lock phenom—
`ena and a severe phase jitter. Therefore, in this carrier tracking
`loop, first, an error signal, which is proportional to the phase
`error between the incoming and the locally generated carrier,
`is generated by nonlinear processing of 4 ,PSK signals. Next,
`the obtained error signal is selectively gated With the control
`signal, which is driven by a phase recognition circuit. Thus,
`this loop’is controlled by only the error signal derived from
`the same phases of 4 PSK signal. In this experimental carrier
`tracking loop, Costas processing with EXCLUSIVE-OR circuits
`is employed for nonlinear processing, and an ECL Master-Slave
`Flip Flop is used for a sample and hold circuit.
`
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`
`
`
`—— NEW TYPE BTF
`
`------CONVENTIONAL BTF
`
`l
`
`l - STAGE SHIFT REGISTER
`
`
`
`2.0
`
`f/fp
`
`IO
`
`O
`
`en 0
`
`05O
`
`
`
`POWERSPECTRALDENSITY(dB)
`
`L .0
`
`
`
`
`
`l
`
`I
`
`‘1 O
`
`Fig.6 Measured and calculated power spectral density: a = 0.5.
`
`C. Carrier Recovery
`
`Highly coherent quadrature references are required for
`synchronous demodulation of the QAM signal. A number of
`methods for generating carrier references from a suppressed
`carrier signal, such as PSK signals, have been established [13].
`Similarly,
`these carrier
`tracking loops are applicable [14].
`However, problems were encountered in the 16 QAM carrier
`recovery. The carrier tracking loop for the QAM signal has to
`hold only four stable phase points as well as 4PSK systems,
`and simultaneously has to have'a good phase jitter perform-
`
`

`

`HORIKAWAetaI.: 200 Mbit/s 16 QAM DIGITAL RADIO SYSTEM
`
`1957
`
`FULL
`WAVE RECT
`
`
`
`ERROR SIGNAL
`’ SELECTIVE GATE
`SIGNAL
`
`
`
`E}
`
`l6 QAM
`
`IF INPUT
`
`
`
`
`
`‘W
`
`VC 0
`
`LOOP FILTER
`
`Fig. 9 Block diagram of the carrier recovery with selective gated PLL.
`
`|.2
`
`._W]TH SELECTIVE GATE
`----— WITHOUT SELECTIVE GATE
`
`T-R BACK TO BACK ERROR
`
`RATE
`
`-7 VALqut-‘IDEAL
`I0
`l6 QAM'
`
`‘
`
`'68 -66
`~70
`-72
`-74
`- --76
`RECEIVE SIGNAL LEVEL (dBm)
`_1_L__..._.L_l_L___.—l_
`l6
`I8
`20' 22
`24
`26
`AVERAGE C/N
`(dB)
`
`Fig.
`
`ll
`
`A
`co
`
`3
`5
`l:
`i
`2
`3
`
`Bit error rate performances. MODEM back to back and overall
`repeater back to back.
`'
`
`8
`
`o
`
`8 a
`3
`
`6 5.U)
`a
`4 >
`a2
`
`.
`E
`
`i
`0 .<,
`51
`
`o
`
`to
`5
`I
`OUTPUT aACK OFF (dB)
`Two TWT‘s nonlinear characteristics.
`
`Fig.
`
`|2
`
`ROLL-OFF FACTOR
`(1 =0. 5
`
`(d8)
`EQUIVALENTC/NDEGRADATICN
`
`OUTPUT BACK OFF ((18)
`l3 Equivarent C/N degradation due toTWT nonlinearity.
`
`Fig.
`
`measured equivalent C/N degradations are shown in Fig. 13.
`From Figs. 12 and 13, it
`is shown that AM-PM conversion
`mainly dominates the error degradation caused by TWT non-
`linearity. From the above results, TWT nonlinear distortions
`on a 16 QAM signal can be evaluated, and it is concluded that
`a TWT should be operated at the power level at which the
`AM:PM conversiOn is less than 2 deg/dB, in order to suppress
`C/N degradation to less than 0.5 dB.
`‘
`‘
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`Page 5
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`
`
`
`NORMALIZEDPHASE
`
`UHECTEROUTPUTVOLTAGE o
`
`.0
`
`
`
`
`
`.0m
`
`.0A
`
`
`o_m \’
`
`ON
`
`_1_i
`5|015202530354045
`
`PHASE ERROR {D (DEG)
`
`Fig. [0 Equivalent phase detector characteristiCs ofthe loop.
`
`Figure 10 shows an equivalent phase detector characteristic
`of this carrier recovery. Note that the loop exhibits only the
`
`desired stable lock point at (1) = O and, simultaneously, im-
`proves detector sensitivity twice as much as the conventional
`4 PSK nonlinear processing.
`The measured rms carrier jitter is about 0.4 degrees, and
`the pull-in frequency range is more than i300 kHz without
`sweep acquisition in 200 Mbit/s 16 QAM systems.
`
`IV. ERROR RATE PERFORMANCE
`
`Figure l 1 shows the theoretical and experimental error rate
`performances of a 200 Mbit/s 16 QAM repeater. The 16 QAM
`MODEM, with about 1.5 dB C/N degradation, is achieved as
`shown in Fig. 11. Main error'degradation sources are 1) imper-
`fection of ring modulator and demodulator, 2) residual inter-
`symbol interference due to delay and amplitude distortions,
`and 3) differential encoding. The difference in C/N degrada-
`tions, between MODEM’s back to back and T—R’s back to
`back, is caused by TWT amplifier nonlinearity.
`
`The effects of TWT amplifier nonlinearity for the 16 QAM
`signal have not been reported. In order to clarify the effects
`due to the nonlinear distortions (AM-PM conversion and gain
`deviation), experimental investigations have been carried out.
`' Two kinds of TWT amplifier with different nonlinear char-
`acteristics and saturation power (45 dBm saturation power for
`LD4217 and 34.5 dBm for LD4007) are used. Figure 12 shows
`the measured nonlinear characteristics with respect to output
`back off of TWT’s. Note that AM-PM conversions prevail, even
`in the smaller output power region. For these TWT’s,
`the
`
`

`

`1958
`
`IEEE TRANSACTIONS ON COMMUNICATIONS, VOL. COM-27, NO. 12, DECEMBER 1979
`
`V. CONCLUSION
`
`System considerations, repeater design and some hardware
`properties of 16 QAM digital microwave system, which are
`capable of 200 Mbit/s signal transmission within 40 MHz RF
`bandwidth, are described and system feasibility is demon-
`strated.
`
`The optimum modulation technique and filtering strategy
`are given for the high efficiency signal processing of a 5 bit/s/
`Hz system. The proposed three distinctive circuit techniques
`are shown to have satisfactory performances for differential
`encoding, binary transversal
`filtering, and carrier recovery.
`Finally, the influence of nonlinear distortion caused by TWT
`amplifier nonlinearity, is clarified experimentally.
`
`ACKNOWLEDGMENT
`
`The authors wish to express their thanks to T. Aratani and
`M. Shinji for fruitful discussions. Thanks are also due to M.
`Araki for his cooperation.
`
`[141
`
`[13] W. C. Lindsey. Synchronization Systems in Communications and
`Control. Englewood Cliffs. NJ: Prentice-Hall. 1972.
`M. K. Simon and J. G. Smith “Carrier Synchronization and Detection of
`QASK Signal Sets." IEEE Trans. Commun.. vol. COM—22. No. 2. pp.
`98-106. Feb. 1974.
`
`if
`
`
`
`Izumi Horikawa was born in Shiga. Japan. on July
`2. 1944. He received the BS. from the University of
`Osaka Prefecture. Japan. in 1968.
`He
`joined
`the Yokosuka Electrical Com-
`munication Laboratory. NTT.
`Japan in
`1968.
`From 1968 to 1974. he was primarily concemed with
`the design and development of demodulator circuits.
`such as a 1.7 GHz carrier recovery circuit. a 200
`Mbaud timing recovery circuit.
`and a digital
`equalizer. for the 20 GHz 400 Mbit/s digital radio
`relay system. Since 1975. he has ben engaged in the
`research and development on a high capacity 16 QAM digital radio system in the
`Radio Transmission Section. His responsibilities have been in the field of
`synchronization. modulation and filtering techniques for bandwidth reduction.
`and phase—locked loops. He is currently an assistant chief in the Radio Device
`and Propagation Section.
`He is a member of the Institute of Electronics and Communications Engineers
`of Japan.
`
`*
`
`
`
`Takehiro Murase was born in Ehime. Japan. on
`October 23. 1944. He received his BS. degree in
`electrical engineering and'M.S. degree in control
`engineering from the Tokyo Institute of Technology
`in 1968 and 1972. respectively.
`'
`Since
`joining the Electrical Communication
`Laboratory. Nippon Telegraph and Telephone Public
`Corporation. Tokyo in 1972. he had been engaged in
`research on interference characteristics on 20 GHZ
`digital
`radio system. He is currently involved in
`studies related to the transmission characteristics of
`'
`1‘
`16 QAM Digital Radio Systems.
`Mr. Murase is a member ofthe Institute of Electronics and Communications
`Engineers ofJapan.
`'
`
`*
`
`
`
`Yoichi Saito was born in Chiba. Japan. on February
`7. 1949. He received his 8.5. degree in electro-
`physics engineering from the Tokyo Institute of
`Technology in 1972.
`Since
`joining the Electrical Communication
`Laboratory. Nippon Telegraph and Telephone Public
`Corporation. Tokyo. in 1972. he had been engaged in
`study on composite—modulation for 20 GHz Digital
`Radio System supervisory and control circuit. He is
`currently developing a demodulator of 16 QAM
`Digital Radio System.
`_
`Mr.‘ Saito is a member of the Institute of Electronics and Communications
`Engineers of Japan.
`
`PMC Exhibit 2040
`PMC Exhibit 2040
`Apple v. PMC
`Apple v. PMC
`|PR2016-00755
`IPR2016-00755
`Page 6
`Page 6
`
`[I]
`
`110]
`
`REFERENCES
`P. R. Hartman. “A 90 MBS Digital Transmission System at 11 GHZ
`Using 8 PSK Modulation."
`lEEE international Conference on
`Communications. Philadelphia. June 1976. pp. 18—8. 18—13.
`[2] M. Ramadan. “Practical Considerations in the Design of Minimum
`Bandwidth. 90 MB.
`8PSK Digital Microwave System."
`IEEE
`International Conference on Communications. Philadelphia. June 1976.
`pp. 29—1 . 29-6.
`[3] B. W Godfrey. Q. S. Chow and F. T. Halsey. “Practical Considerations
`in the Design of th DRS-8 Digital Radio." IEEE International
`Conference on Communications. Chicago. June 1977. pp. 55-106.
`554 10.
`[4! C. W. Anderson and S. Barber. “Modulation Considerations for a 91
`Mbit/s Digital Radio". IEEE Trans. Commun.. Vol. COM—26. No. 5.
`pp. 523-528. May 1978.
`[5| H. Yamamoto. K. Kohiyama and K. Morita. “400 Mb/s QPSK Repeater
`for 20—GHz Digital Radio-Relay System." lEEE Trans. MTT. Vol.
`MTT-23. No. 4. pp. 334—341. April 1975.
`[6| A. C. Longton. “DR-18 A High Speed QPSK system at 18 GHz.“ IEEE
`International Conference on Communications. Philadelphia. June 1976.
`pp. 18~I4.18—17.
`[7! V. N. Campopiano and B. G. Glazer. “A Coherent Digital Amplitude
`and Phase Modulation Scheme.“ IRE Trans. Com. Sysr.. Vol. CS—IO.
`pp. 90-95. Mar. 1962.
`'
`[8] K. Kawai. S. Shintani and H. Yanagidani. "Optimum Combination of
`Amplitude and Phase Modulation Scheme and its Application to Data
`Transmission MODEM."
`lEEE
`International Conference
`on
`Communications. Philadelphia. June' 1972. pp. 29-6. 2911.
`[9| K. Miyauchi. S. Seki and H. Ishio. “New Technique forGenerating and
`Detecting Multi-Level Signal Formats.“ IEEE Trans. Commun.. vol.
`COM—24. No. 2. pp. 263-267. Feb. 1976.
`W. J. Weber. “Differential Encoding for Multiple Amplitude and Phase
`Shift Keying Systems." IEEE Trans. Commun.. vol. COM-26. No. 3.
`pp. 385—391.Mar. 1978.
`H. B. Voelcker. "Generation of Digital Signaling Waveforms.“ IEEE
`Trans. Commun.. vol. COM—16. pp. 81-93. Feb. 1968.
`K. Feher and R. D. Cristofaro. “Transversal Filter Design and
`Application in Satellite Communications.” IEEE Trans. Commun. vol..
`COM-24. pp. 1262-1267. Nov. 1976.
`
`Ill]
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`1121
`
`