United States Patent [19]
`Tuttle et al.
`
`[54] MODULATED SPREAD SPECTRUM IN RF
`IDENTIFICATION SYSTEMS METHOD
`
`[75] Inventors: John R. Tuttle, Corrales, N.M.;
`Eugene H Hoyt’ Colorado Springs,
`Colo; James C. Springett, La
`Crescenta, Calif.
`
`[73] Assignee: Micron Technology, Inc., Boise, Id.
`
`[21] Appll NO; 32,334
`_
`122] Fli?dl
`[51] Int Cl 6
`[52] U S Ci """"""""""""""
`
`Mar- 17, 1993
`
`[58] Field of Search
`
`[56]
`
`_
`375/203’ 204’ 208’ 367’ 370/18’ 19’ 107
`.
`References cued
`US, PATENT DOCUMENTS
`
`H04B 1/69
`375/367_
`37O/18
`375/200 202
`
`llllllllllllllIllllllllllllllllllllllllllllllllllllllllllllllllllllllllllll
`[11] Patent Number:
`5,539,775
`[45] Date of Patent:
`Jul. 23, 1996
`
`I USOO5539775A
`
`.
`
`5,170,411 12/1992 Ishigaki ................................. .. 375/200
`.
`.
`Primary Examzner—Stephen Chm
`Assistant Examiner —_D on V0 _
`Attorney, Agent, or Fzrm~Henr1 J. A. Charmasson; John D.
`Buchaca
`[57]
`
`ABSTRACT
`
`A method for RF communication between transceivers in a
`radio frequency identi?cation system that improves range,
`decreases multipath errors and reduces the effect of outside
`RF source interference by employing spread spectrum tech—
`niques. By pulse amplitude modulating a spread spectrum
`carrier before transmission, the receiver can be designed for
`simple AM detection, suppressing the spread spectrum car
`rier and recovering the original data pulse code waveform.
`The data pulse code waveform has been further encrypted by
`a direct sequence pseudo-random pulse code. This addi
`tional conditioning prevents the original carrier frequency
`components from appearing in the broadcast power spectra
`and provides the basis for the clock and transmit carrier of
`the transceiver aboard an RFID tag. Other advantages
`include high resolution ranging, hiding transmissions from
`eavesdroppers, and Selective addressing
`
`3 Claims, 10 Drawing Sheets
`
`1/1989 Lundquist et a1. .................... .. 375/200
`4,817,113
`7/1990 Iwasaki .............. ..
`375/200
`4,941,150
`3/1992 Mikoshiba et a1.
`......... .. 375/202
`5,099,495
`5,157,688 10/1992 Dell-Imagine ........................ .. 375/200
`
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`US. Patent
`
`Jul. 23, 1996
`
`Sheet 1 of 10
`
`5,539,775
`
`FIG. 1
`
`2
`
`

`
`US. Patent
`
`Jul. 23, 1996
`
`Sheet 2 0f 10
`
`5,539,775
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`US. Patent
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`Jul. 23, 1996
`
`Sheet 3 of 10
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`5,539,775
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`

`
`U.S. Patent
`
`Jul. 23, 1996
`
`Sheet 4 of 10
`
`5,539,775
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`
`5
`
`

`
`U.S. Patent
`
`Jul. 23, 1996
`
`Sheet 5 of 10
`
`5,539,775
`
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`
`US. Patent
`
`Jul. 23, 1996
`
`Sheet 6 of 10
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`5,539,775
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`
`7
`
`

`
`US. Patent
`
`Jul. 23, 1996
`
`Sheet 7 of 10
`
`5,539,7 75
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`CARRIER MODULATION SPECTRUM
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`
`8
`
`

`
`U.S. Patent
`
`Jul. 23, 1996
`
`Sheet 8 of 10
`
`5,539,775
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`I
`
`Jul. 23, 1996
`
`Sheet 9 of 10
`
`5,539,775
`
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`
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`
`FIG. 12
`
`10
`
`

`
`US. Patent
`
`Jul. 23, 1996
`
`Sheet 10 0f 10
`
`5,539,775
`
`FIG. 15
`
`FIG. 14
`
`FIG. 15
`
`FIG. 16
`
`11
`
`

`
`5,539,775
`
`1
`MODULATED SPREAD SPECTRUM IN RF
`IDENTIFICATION SYSTEMS METHOD
`
`FIELD OF THE INVENTION
`
`This invention relates to data communication with low
`power radio frequency (RF) transceivers, speci?cally to data
`communication between RF identi?cation (RFID) tags and
`an operator (interrogator), and more particularly, to using
`spread spectrum techniques to simplify the tag receiver
`design, reduce tag cost, increase the range and reduce
`interference on the RF communication channel.
`
`BACKGROUND OF THE INVENTION
`
`An RFID tag is a small radio transceiver that can be
`attached to a movable article to help keep track of its
`whereabouts and status. A small, light-weight inexpensive
`tag is desirable. To miniaturize the tag, the size of the
`circuitry is reduced by using an integrated circuit (IC)
`design. The simpler the circuit, the smaller the resulting IC
`becomes thereby reducing the cost.
`Operational environment factors can disrupt the reliability
`of these weak communication links. Re?ective and refrac
`tive properties of the environment introduce possible mul
`tipath errors, and outside RF sources introduce interference
`in the received signal.
`
`SUMMARY OF THE INVENTION
`
`The principle object of this invention is to provide a
`method by which an RFID tag can communicate more
`information over much greater distances through harsher RF
`environments than previously available, while reducing the
`size and cost of the tag IC.
`The secondary objects of this invention are to provide
`simple AM reception by the receiver circuit without having
`to use complex synchronization schemes, external frequency
`references such as quartz crystals and complex circuits like
`Costas loops;
`interrogator transmission power to be less than or equal to
`1 Watt, allowing unlicensed operation within FCC
`guidelines;
`extension of the range from interrogator to RFID tag;
`limitation of eavesdropping of signals by outside parties,
`and;
`selective addressing of particular tags or communicating
`with more than one tag simultaneously in any given
`transmission.
`These and other objects are achieved by modulating a
`spectrally spread carrier with a pulse code waveform rep
`resenting information pertaining to the article to which the
`tag is attached.
`One advantage of using a spread spectrum modulated
`signal is the enhanced interference rejection obtained during
`the demodulation process. The effect is a signal to noise gain
`over traditional narrow band broadcasting techniques.
`Using spread spectrum techniques gives good range with
`relatively little complexity on the tag. The high complexity
`is contained in the interrogator design.
`Other advantages are: selectively addressing a particular
`receiver, pulse code multiplexing allowing addressing of a
`plurality of receivers in any given transmission, broadcast
`transmissions with low density power spectra for signal
`hiding, message encryption to discourage eavesdroppers,
`and high resolution ranging between transmitter and
`receiver.
`
`2
`Although this invention was speci?cally designed for
`RFID tag communication, there are applications for it in
`hand-held walkie-talkies, pagers, mobile phones, cordless
`telephones, cordless microphones and musical instruments,
`cordless computer network communication links, and inter
`coms.
`
`BRIEF DESCRIPTION OF THE DRAWINGS
`
`FIG. 1 is a simpli?ed drawing of the type of radio
`frequency identi?cation system to which this invention
`applies.
`FIG. 2 is a block diagram of the transmitter part of the
`interrogator.
`FIG. 3 is a series of time domain plots representing binary
`phase shift keying (BPSK) the carrier with a direct sequence
`pseudo-random pulse code.
`FIG. 4 is a frequency domain plot of an ideally spectrally
`spread carrier.
`FIGS. SA-SB are a realistic time domain plot and fre
`quency domain plot of the BPSK spectrally spread carrier.
`FIG. 6 is a series of time domain plots representing pulse
`amplitude modulating the spectrally spread carrier with a
`pulse code waveform.
`FIG. 7A-7B are a realistic time domain plot and fre
`quency domain plot of a pulse amplitude modulated non
`spectrally spread carrier.
`FIG. 8A-8B are a realistic time domain plot and fre
`quency domain plot of a pulse amplitude modulated spec
`trally spread carrier.
`FIG. 9 is a block diagram of the transceiver in the
`REMOTE RFID tag.
`FIG. 10 is a time domain plot representing one data bit
`period of the received signal l+PN1(t)d(t) after the carrier
`has been removed.
`FIG. 11 is a time domain plot representing one data bit
`period of the received signal without carrier that has been
`gated by PN1(t) in sync.
`FIG. 12 is a time domain plot representing one data bit
`period of the received signal without carrier that has been
`gated by PN1(t) that is 3 chips out of sync.
`FIG. 13 is a time domain plot representing one data bit
`period of the received signal without carrier that has been
`gated by the logical compliment of PN1(t) in sync when
`d(t)=l.
`FIG. 14 is a time domain plot representing one data bit
`period of the received signal without carrier that has been
`gated by PN1(t) in sync when d(t)=l.
`FIG. 15 is a time domain plot representing one data bit
`period of the received signal without carrier that has been
`gated by the logical compliment of PN1(t) in sync when
`d(t)=—l.
`FIG. 16 is a time domain plot representing one data bit
`period of the received signal without carrier that has been
`gated by PN1(t) in sync when d(t)=—l.
`
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`DESCRIPTION OF THE PREFERRED
`EMBODIMENT OF THE INVENTION '
`
`65
`
`Referring now to FIG. 1, a typical arrangement of an
`RFID system includes a plurality of operator controlled
`interrogators A that communicate via RF links with a
`plurality of RFID tags B which are attached to articles C
`located somewhere in the vicinity of the interrogators.
`
`12
`
`

`
`3
`An interrogator transmits a signal to a particular tag
`requesting or updating the status of the article associated
`with that tag. Status information can include the article’s
`name, owner, address, destination, pertinent dates, weight,
`etc.
`The tag receives the signal and awakens from its power
`conserving quiescent sleep state. The tag interprets the
`request and decides on a response which is then transmitted
`back to the requesting interrogator. A receiver in the inter
`rogator analyzes the return signal, determining the status of
`the article. Its location can be determined by triangulation
`with multiple interrogators or multiple antennas. This infor
`mation is then displayed to the operator.
`Referring now to FIG. 2, a block diagram of the trans
`nritter part of the interrogator transceiver, a crystal con
`trolled oscillator 1 generates a constant amplitude sinusoidal
`carrier with a frequency of 2441.75 MHZ.
`This carrier is modulated in the balanced modulator 2 by
`a pseudo-random direct sequence pulse code, PN2, gener
`ated by the pseudo-noise (PN) generator 3. PN2 has a chip
`rate of about 40 Mega-chips per second.
`The resulting spectrally spread carrier is then modulated
`4 by another pulse code waveform generated by combining
`a data waveform 5 and another pseudo-noise waveform,
`PN1(t) 6. The resulting signal to be transmitted is sent to a
`power ampli?er 7, and then to the antenna 8.
`The data waveform, d(t) represents information to be
`transmitted to the REMOTE and has a data rate of about 2
`megabits per second. PH1(t) has a chip rate equal to or less
`than PN2. In this embodiment PN1(t) and d(t) are multiplied
`together to form the modulating unipolar waveform,
`l+PN1(t)d(t).
`The waveforms of FIG. 3 refer in detail to operation of the
`?rst balanced modulation of the carrier. Modulation occurs
`by binary phase shift keying (BPSK) the original constant
`amplitude carrier 9, with the PN2 pseudo-random direct
`sequence pulse code 10. During the time when PN2 is low,
`the carrier is keyed with a 180 degree phase shift from when
`PN2 is high 11. For illustration purposes, the sinusoidal
`carrier is shown at a much lower frequency than has been
`suggested for this embodiment.
`The resulting waveform has a power spectrum which is
`ideally spread according to the (sin(x)/x)**2 function shown
`in FIG. 4. The center of the main lobe is at the original
`carrier frequency Fe. The mainlobe bandwidth (null to null)
`is now twice the clock frequency of the PN2 waveform Fg.
`A more realistic representation of the time domain signal
`and its power spectra appears in FIG. 5a and FIG. 511
`respectively.
`FIG. 6 illustrates in detail the operation of the modulation
`of the spectrally spread carrier by the pulse code waveform
`1+PN1(t)d(t). The spectrally spread carrier 11 of FIG. 3 is
`viewed over a much greater time period 12. One must keep
`in mind that there are over 150 cycles of the original carrier
`for every bit (amount of time between possible changes,
`high-to-low or low-to~high) of the l+PN1 (t) d (t) modulat
`ing waveform 13.
`Pulse amplitude modulation involves multiplying the
`spread spectrum canier by the modulating waveform 14. It
`is important to note that pulse amplitude modulating a
`non-spectrally spread carrier results in a power spectrum
`with an undesirable spike 15 at the carrier frequency as
`shown in FIG. 7a and FIG. 7b. By spectrally spreading the
`carrier, the resultant power spectra is more acceptable as
`shown in FIG. 8a and FIG. 81;.
`Referring now to FIG. 9, the receiver portion of the
`remote RFID tag receives signals via an antenna 16. In order
`
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`4
`to sense the existance of a signal meaningful to the
`REMOTE, the signals are sent to a lowpass ?lter 17. This
`voltage is compared to a value V0 in a comparator 18. If the
`voltage is greater, the comparator outputs a true, informing
`the sequential mode logic 19 that a signal is present. The
`receiver is then initialized, awakening from its quiescent
`sleep state and the PN1(t) waveform generation circuitry is
`set to acquisition mode through the PN epoch controller 20.
`In an awakened REMOTE, the signal arriving at the
`antenna 16 is sent to several stages of RF (S-band) ampli
`?cation and ?ltering 21. As much RF ampli?cation as
`attainable is provided. The bandpass ?lter characteristics are
`a center frequency of the original carrier (2441.75 MHZ) and
`a bandwidth of around 84 MHz, which corresponds to the
`main lobe of the transmitted signal. Although this bandpass
`?ltering is shown as a single block 22, the circuitry accom
`plishing this ?ltering will likely be distributed throughout
`the RF ampli?er circuits.
`The resulting signal is sent to a full wave envelope
`demodulator 23, which recovers the amplitude modulating
`waveform as 1+PN1(t)d(t). Some added lowpass ?ltering 24
`and baseband ampli?cation 25 further increases the voltage
`level of the waveform to a more useful value (approx. 0.5
`Volts peak-to-peak).
`As seen in FIG. 10 the waveform l+PN1(t)d(t) is essen
`tially two-level in nature, exhibiting positive and negative
`transitions between the limits of 0 and Vs which correspond
`to the PN1(t) waveform, and a small amount of additive
`noise or interference. Here, PN1(t) is represented by the chip
`sequence: 1 l l—1—l l—l—1, and spans exactly one data bit
`period, Td. It can be assumed during this time period that
`d(t)=l; if it were a —l, the waveform would be inverted. In
`order to facilitate synchronization, d(t) is equal to l and thus
`non-transitioning during a preamble portion of the transmit
`ted signal.
`Referring back to FIG. 9, the transition pulse generator 26
`acts to differentiate the 1 +PN1(t)d(t) waveform, producing
`“spikes” at the transition times, which are then recti?ed
`(absolute value) to furnish a unipolar train of transition
`derived pulses. Spectrally, this waveform is rich in the
`PN1(t) code clock frequency. It’s injected into a free
`running multivibrator 27 (or a phase-locked loop oscillator)
`which produces a synchronized pulse for every chip of the
`received signal. In effect the waveform produced is a rela
`tively stable PN1 clock waveform used by the PN1 genera
`tor 28. Once synchronization to the incoming PN1 sequence
`is achieved, the locked multivibrator 27 is used as the tag’s
`clock for processing the incoming data and commands. If a
`return transmission is requested, the locked multivibrator 27
`becomes the source for the tag transmitter carrier frequency.
`The PN1 generator 28, running from the PN-clock wave
`form, generates the tag’s version of the PN1(t) chip
`sequence. This will initially be time offset by an integral
`number of chips (11) from the sequence that makes up the
`received waveform 1+PN1(t)d(t). The PN Sync Detector 29
`eleminates this offset.
`The PN Sync Detector accomplishes its objective by
`doing a sequential correlation comparison. A MOSFET
`switch 30 gates the l+PN1(t)d(t) code received using the
`PN1 code generated. If the two codes are in sync, the output
`of the gate is a waveform FIG. 11, having an average value
`around Vs/2. If the two codes are unaligned, say offset by
`three chips, the gate output is a waveform FIG. 12, having
`an average value around Vs/4 or less. By usual spread
`spectrum standards, correlation discrimination by a factor of
`two would be considered poor, but under very high signal to
`
`13
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`
`5,539,775
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`5
`noise ratio (SNR) conditions, which are present in the
`forward link, it is acceptable.
`The proper, in-sync starting point for the PN1 code
`sequence is discovered by generating PN1 at all possible
`starting points and recording the results. Each starting point
`represents a waveform which is a possible candidate for
`being the correct insync PN1 waveform. The candidate
`having the largest correlation value will be the one used. A
`more detailed explanation of this process follows.
`Upon initialization, the Sequential Mode Logic 19, in
`conjunction with the timing events T1 and T2 31, initializes
`the sample-and-hold (S&H) memories 32 and 33. T2 occurs
`later than T1. The event of T1 causes the output of LPF4 34
`to be stored in S&H #1 as value El. Likewise, T2 results in
`E1 being stored in S&H #2 as value E2. LPF4 is responsible
`for smoothing the correlation voltage producing a more
`accurate representation of the correlation than what is avail
`able from the broader LPF2 35 which is matched to the bit
`period. Initially, the sequential mode logic sets the PN epoch
`controller 20 to produce a shift-register word which is
`loaded into the PN generator’s shift-register directing the
`generator to produce the PN1 code sequence at a particular
`staring point called epoch-1. If the code sequence is made up
`of L chips, there are L possible starting points and L possible
`epochs. (In the example waveform in FIG. 4, L=8). The PN
`sync detector must ?nd epoch-n corresponding to the largest
`correlation, with l<=n<=L. Once the generator is producing
`PN1 based on epoch-1, the correlation value output from
`LPF4 is transferred to S&H #1 at T1 and then to S&H #2 at
`T2.
`Next, the chip sequence based on epoch-2 is generated.
`The output from from LPF4 is again sent to S&H #1 at T1.
`If El is greater than E2, the correlation at epoch-2 is greater
`than the correlation at epoch-1. This is communicated by the
`output of the comparator 36 to the sequential mode logic
`which transfers El to S&H #2 and makes note of the epoch
`value It (in this case 2) for which the transfers occurred. If
`El had been less than E2, Then E2 remains unchanged. The
`acquisition search continues with epoch-3 and so on until all
`epoch values have been tried. In the end E2 will represent
`the largest correlation value found, and the sequential mode
`logic will know which epoch value to use.
`i
`There is the possibility that proper alignment was not
`achieved because a correlation value mistake was made due
`to noise. To avoid this, the entire aquisition process is
`repeated two more times. PN synchronization success is
`declared if the epoch chosen is the same for all three tries (or
`two out of three tries).
`Once PN1 has been synchronized with the received
`signal, the REMOTE is ready to detect data. Remember,
`throughout the acquisition period the LOCAL interrogator
`has been transmitting with d(t) equal to 1. This period will
`last for several hundred data bit periods. The start of a
`message packet will be indicated by a data transition from 1
`to —l, followed by a block of unique bits which the data
`management logic will recognize as packet synchronization.
`Data detection itself takes place as the received signal
`l+d(t)PN1(t) is gated 30 by the tag’s generated PN1 and sent
`to LPF2 35. Likewise, it is gated by PNl’s logical compli
`ment 37 and sent to LPFl 38. When d(t) is equal to l, the
`input to LPFl is the waveform seen in FIG. 13, and the
`corresponding input to LPF2 is seen as FIG. 14. The average
`value of the waveform sent to LPF2 is clearly greater. When
`d(t) is equal to —l, the inputs to LPFl and LPFZ are as shown
`in FIG. 15 and FIG. 16 respectively. Here, the average value
`of the waveform sent to LPFl is greater.
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`The two low pass ?lters, LPFl and LPF2 are approxima
`tions to matched ?lters (if a simple RC ?lter is used, then RC
`=l.3/(2 pi*Td)). They function to maximize the SNR at the
`end of the bit period, Td, for input to the comparator 39. End
`of bit period timing is provided by a state AND gate 40
`operating from the PN1 generator. This gate toggles a J—K
`?ip-flop 41 which temporarily stores the value of d(t) until
`it can be transferred into data management logic 42 and
`RAM 43.
`Each sequential data bit provides its own detection ref
`erence to the comparator. Also, as the received signal level
`varies due to movement of the tag or interrogators or
`changes in the RF environment, the data detection decision
`reference is self-adjusting, giving rise to the designation
`“per-bit” reference for data detection.
`Once the forward link data has been detected, the data
`management logic 42 (executes required actions) and
`assembles the return link data packet message.
`The transmitter within the tag will generate the return link
`carrier by multiplying the PN1 clock frequency produced by
`the locked multivibrator 27 by a factor M using a frequency
`multiplying circuit 44. This carrier is then ampli?ed 45 and
`pulse amplitude modulated 46 by the return data packet
`waveform. Although an amplitude modulation scheme is
`shown, other modulation schemes using spread spectrum
`techniques are possible and desirable. This signal is then
`sent to the antenna 47. The power ampli?er stage may be
`less than 0.5 milliwatts due to the power available within the
`tag. Because of this low radiated power, spectrally spreading
`the return link carrier may not be necessary, and an addi—
`tional pseudo-noise waveform analogous to PN1 need not
`further encrypt d(t). When higher powers are desired, a
`spectrally spread scheme would be used to satisfy FCC
`unlicensed rules. To extend the range of the system, spread
`spectrum techniques are desirable in the tag transmitter.
`While the preferred embodiments of the invention have
`been described, modi?cations can be made and other
`embodiments may be devised without departing from the
`spirit of the invention and the scope of the appended claims.
`What is claimed is:
`1. A method for communicating pulse coded information
`between low power transceivers which comprises:
`spectrally spreading a carrier; and
`modulating the spectrally spread carrier with a data pulse
`code waveform comprising information to be transmit
`ted, to form a modulated spectrally spread (MSS)
`signal;
`wherein said modulating the spectrally spread carrier
`comprises pulse amplitude modulating and said MSS
`signal comprises a pulse amplitude modulated spec
`trally spread signal;
`wherein said method further comprises:
`modulating said data pulse code waveform with a
`second direct sequence pseudo-random pulse code
`waveform;
`transmitting said MSS signal,
`receiving said MSS signal, and
`extracting said data pulse code waveform from said
`MSS signal;
`wherein said extracting comprises:
`removing said spectrally spread carrier from said MSS
`signal resulting in a received pulse code waveform
`comprising said data pulse code waveform and said
`second direct sequence pseudo-random pulse code
`waveform;
`generating a third direct sequence pseudo-random
`pulse code waveform substantially similar to and in
`
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`7
`synchronization with said second direct sequence
`waveform;
`gating said received pulse code waveform with said
`third direct sequence waveform; and
`?ltering the output of said gating step resulting in an
`average value representing one of two possible logi
`cal values for said data pulse code waveform;
`wherein said generating a third direct sequence psuedo
`random pulse code waveform (DSPPCW) comprises:
`transmitting a preamble portion of said MSS signal
`wherein said data pulse code waveform is non
`transitioning,
`producing a clock waveform in-sync with said second
`DSPPCW,
`performing a sequential correlation comparison
`between said received pulse code waveform and
`each of a plurality of candidate waveforms,
`recording the results of each comparison, and
`choosing the candidate with a highest correlation value
`as said third DSPPCW.
`2. The method of claim 1 which comprises:
`using said clock waveform as a return carrier frequency
`source.
`3. An apparatus for identifying and tracking the where
`abouts of moving bodies around a de?ned area which
`comprises:
`an interrogating station including a ?rst transceiver;
`said ?rst transceiver comprising:
`means for generating a carrier;
`means for spectrally spreading said carrier;
`means for generating a data pulse code waveform; and
`means for modulating said carrier with said data wave
`form resulting in a modulated carrier waveform;
`at least one tag associated with one of said bodies,
`said tag including a second transceiver,
`said second transceiver comprising:
`means for extracting said data pulse code waveform
`from said modulated carrier waveform;
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`wherein said apparatus further comprises means for
`modulating said data waveform with a second direct
`sequence pseudo-random pulse code waveform;
`wherein said means for extracting comprises:
`means for removing said spectrally spread carrier from
`said modulated carrier waveform resulting in a
`received pulse code waveform comprising:
`said data pulse code waveform and
`said second direct sequence pseudo-random pulse
`code waveform;
`means for generating a third direct sequence pseudo
`random pulse code waveform substantially similar to
`and in synchronization with said second direct
`sequence waveform;
`means for gating said received pulse code waveform
`with said third direct sequence waveform;
`means for ?ltering the output of said means for gating
`resulting in an average value representing one of two
`possible logical values for said data pulse code
`waveform;
`wherein said means for generating a third direct sequence
`psuedo-random pulse code waveform (DSPPCW) com
`prises:
`means for transmitting a preamble portion of said MSS
`signal wherein said data pulse code waveform is non
`transitioning,
`means for producing a clock waveform in-sync with said
`second DSPPCW,
`means for performing a sequential correlation comparison
`between said received pulse code waveform and each
`of a plurality of candidate waveforms,
`means for recording the results of each comparison,
`means for choosing the candidate with a highest correla
`tion value as said third DSPPCW.
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`15