On the Direct Conversion Receiver -- A Tutorial
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`On the Direct Conversion Receiver -- A Tutorial
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`A detailed look at the characteristics of the direct conversion receiver
`
`Ashkan Mashhour, William Domino, Norman Beamish
`June 1, 2001
`
`TUTORIAL
`
`On the Direct Conversion Receiver -- A Tutorial
`
`Increased pressure for low power, small form factor, low cost and reduced bill of materials in such radio
`applications as mobile communications has driven academia and industry to resurrect the direct conversion
`receiver. Long abandoned in favor of the mature superheterodyne receiver, direct conversion has emerged over
`the last decade or so thanks to improved semiconductor process technologies and astute design techniques.
`This article describes the characteristics of the direct conversion receiver and the issues it raises.
`
`Ashkan Mashhour,
`William Domino
`and Norman Beamish
`Conexant Systems
`Newport Beach, CA
`
`Very much like its well established superheterodyne receiver counterpart, first introduced in 1918 by
`Armstrong,1 the origins of the direct conversion receiver (DCR) date back to the first half of last century when
`a single down-conversion receiver was first described by F.M. Colebrook in 1924,2 and the term homodyne
`was applied. Additional developments in 1947 led to the publication of an article by D.G. Tucker,3 which first
`coined the term synchrodyne, for a receiver which was designed as a precision demodulator for measurement
`equipment rather than a radio. Another paper by Tucker in 19544 reports the various single down-conversion
`receivers published at the time and clarifies the difference between the homodyne (sometimes referred to as
`coherent detector) and the synchrodyne receivers -- the homodyne receiver obtains the LO directly (from the
`transmitter, for example), whereas the synchrodyne receiver synchronizes a free-running LO to the incoming
`carrier.
`
`Over the last decade or so, the drive of the wireless market and enabling monolithic integration technology
`have triggered research activities on direct conversion receivers, which integrated with the remaining analog
`and digital sections of the transceiver, have the potential to reach the "one-chip radio" goal. Besides, it favors
`multi-mode, multi-standard applications and thereby constitutes another step towards software radio.
`
`The present article refers to several recent publications5,6 which provide a thorough survey and insight, and
`display renewed interest in direct conversion receivers. Overcoming some of the problems associated with the
`traditional superheterodyne and being more prone to integration, DCR has nevertheless an array of inherent
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`challenges. After a brief description of alternative and well-established receiver architectures, this article
`presents the direct conversion reception technique and highlights some of the system level issues associated
`with DCR.
`
`TRADITIONAL RECEPTION TECHNIQUES
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`The Superheterodyne Receiver
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`The superheterodyne or heterodyne receiver is the most widely used reception technique and finds numerous
`applications from personal communication devices to radio and TV tuners. It has been used extensively and is
`well understood. It comes in a variety of combinations,7,8,9 but essentially relies on the same principle -- the RF
`signal is first amplified in a frequency selective low noise stage, then translated to a lower intermediate
`frequency (IF) with significant amplification and additional filtering, and finally down-converted to baseband
`with either a phase discriminator or straight mixer, depending on the modulation format. This technique is
`illustrated in the schematic of Figure 1.
`
`The use of a superheterodyne technique entails several trade-offs. Image rejection is a prevailing concern in
`this architecture. During the first down-conversion to IF, any unwanted activity at a frequency spaced at fIF
`offset from the LO frequency (fLO ) on the opposite side of fLO from the desired RF channel, will produce a
`mixing product falling right into the down-converted channel at fIF . In practice, a RF bandpass filter, usually a
`surface acoustic wave (SAW) device, is utilized to perform band selection ahead of the low noise amplifier
`(LNA), while a second filter follows the LNA to perform image rejection. If these filters are identical they
`share the burden of the two functions. But some amount of image rejection must follow the LNA, for without
`it, the LNA's noise figure will effectively double due to the mixing of amplified image noise into the IF
`channel. Instead of the RF SAW filter, other passive filtering technologies such as dielectric or ceramic
`resonators can also be featured. The higher the IF, the more relaxed the requirements on the cut-off frequency
`of the image reject filter. Once at the IF, the presence of an interfering signal in the vicinity of the channel
`mandates sharp filtering around the channel; this filtering is performed after the first mixer by the channel
`select filter, which is also often an IF SAW filter. Figure 2 shows this filtering process. Essentially, the exercise
`is that of a carefully engineered balance among several variables, including the rejection provided by the
`various filters, frequency planning and linearity of the active stages. Dual IFs provide additional room to
`maneuver with filter selectivity, but somewhat complicate the frequency planning.
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`The selectivity required of the two aforementioned filters (in terms of fractional bandwidth) makes them
`unsuitable candidates in the foreseeable future for integration, due to the low Qs of current silicon processes,
`and have to be implemented by bulky off-chip components. The IF channel filter in particular requires high Q
`resonators for its implementation -- the higher the IF, the lesser the filter's fractional bandwidth (that is, its ratio
`of bandwidth to center frequency), necessitating ever-higher Q. This high Q requirement is most commonly
`met by the use of piezoelectric SAW and crystal filters. This introduces additional constraints, as those filters
`often require inconvenient terminating impedances, and matching may impinge on such issues as noise, gain,
`linearity and power dissipation of the adjoining active stages. The narrower the fractional bandwidth, the more
`likely the filter's passband shape will exhibit an extreme sensitivity to variations in matching element values.
`Additionally, the specificity of the IF filter to the bandwidth of the signal and hence the standard used makes
`superheterodyne receivers unsuitable for multi-standard operation. Nonetheless, superheterodyne is known for
`its high selectivity and sensitivity.
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`Image-reject Receivers
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`Alternatively, by smart use of trigonometric identities, the image can be removed without the need of any post-
`LNA image-reject filtering. This is the principle of image-reject receivers8,10 , the first of which is the Hartley
`architecture, introduced in 192811 . It makes use of two mixers with their local oscillators in a quadrature phase
`relationship; this separates the IF signal into in-phase (I) and quadrature (Q) components. It then shifts the Q
`component by 90° before recombining the two paths, where the desired signal, present in both paths with
`identical polarities, is reinforced, while the image, present in both paths with opposite polarities, is cancelled
`out. The dual of the Hartley architecture, known as the Weaver image-reject receiver,12 achieves the relative
`phase shift of one path by 90° by the use of a second LO enroute to another IF or to baseband. The same result
`is achieved. However, the reliability of these receivers heavily depends on the accuracy of the I/Q paths, that is,
`the gain and phase imbalance between the two branches. Figures 3 and 4 show diagrams of the Hartley and
`Weaver image-reject architectures, respectively (high frequency mixing products are removed by low-pass
`filtering -- not shown on figures).
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`Low IF Single Conversion Receiver
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`Low IF single conversion, shown in Figure 5, is an offspring of the DCR. Its main purpose is to protect the
`receiver from all the DC-related problems that pertain to DCR, while retaining the DCR's benefit of
`elimination of high Q IF filters. As its name indicates, instead of directly converting the signal to baseband, the
`LO is slightly offset from the RF carrier, typically one to two channels. The low IF means that the fractional
`bandwidth of the IF bandpass filtering is large, making it possible to implement it with low Q components. The
`IF SAW or crystal filter needed in the high IF case can be replaced with an active RC filter or other filter
`suitable for low frequency operation, that is also conducive to silicon integration. The low IF signal may be
`translated to baseband through another mixer, or preferably, in the digital domain following analog-to-digital
`(A/D) conversion. Of course, this comes at the expense of faster and higher resolution A/D converters. If the IF
`frequency is equal to only one or two channel widths, then it is not possible to provide image rejection at RF,
`as the RF filter must be wide enough to pass all channels of the system. In this case, all image rejection must
`come from the quadrature down-conversion to the low IF, which itself resembles the Hartley architecture, once
`the baseband conversion is added.
`
`Wideband IF with Double Conversion
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`This architecture, shown in Figure 6, is very similar to the superheterodyne configuration. In this case, the first
`mixer utilizes an LO that is at a fixed frequency, and all channels in the RF band are translated to IF, retaining
`their positions relative to one another. The second mixer utilizes a tunable LO, thus selecting the desired
`channel to be translated to baseband. A subsequent lowpass filter suppresses adjacent channels.
`
`DIRECT CONVERSION RECEIVERS
`
`Direct conversion reception, shown in Figure 7, and also referred to as homodyne, or zero-IF, is the most
`natural solution to receiving information transmitted by a carrier. However, it has only been over the past
`decade or so that this type of reception has found applications other than pagers.13 Direct conversion reception
`has several qualities which makes it very suitable for integration as well as multi-band, multi-standard
`operation, but there are severe inherent obstacles that have for a long time kept it in the shadow of the
`superheterodyne technique.
`
`First, the problem of the image has been eliminated, since the IF is zero and the image to the desired channel
`(for all but single-sideband signals) is the channel itself. Then, only one local oscillator is required, which
`means only one phase noise contribution. The need for the bulky off-chip filters is consequently removed.
`Filtering now only occurs at low frequencies (baseband) with some amplification, which means less current
`consumption than at higher frequencies (to drive device parasitics), fewer components and lower cost.
`Practically, however, strong out-of-band interference or blocking signals may need to be removed prior to
`down-conversion in order to avoid desensitizing the receiver by saturating subsequent stages, as well as
`producing harmonics and intermodulation terms which will then appear in the baseband. Such a filter may be
`placed after the LNA for example. DCR, however, brings its own set of issues.
`
`DC Offsets
`
`In direct conversion, as the signal of interest is converted to baseband very early in the receive chain, without
`any filtering other than RF band-selection, various phenomena contribute to the creation of DC signals, which
`directly appear as interfering signals in the band of interest, as shown in Figure 8.
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`The LO may be conducted or radiated through an unintended path to the mixer's RF input port, thus effectively
`mixing with itself, producing an unwanted DC component at the mixer output. Worse still, this LO leakage
`may reach the LNA input, producing an even stronger result. This effect presents a high barrier against the
`integration of LO, mixer and LNA on a single silicon substrate, where numerous mechanisms can contribute to
`poor isolation. These include substrate coupling, ground bounce, bond wire radiation, and capacitive and
`magnetic coupling.
`
`Conversely, a strong in-band interference signal, once amplified by the LNA, may find a path to the LO-input
`port of the mixer, thus once again producing self-mixing.
`
`Some amount of LO power will be conducted through the mixer and LNA (due to their non-ideal reverse
`isolation) to the antenna. The radiated power, appearing as an interferer to other receivers in the corresponding
`band, may violate emissions standards of the given system. It is important to note that since the LO frequency
`is inside the receive band, the front-end filters do nothing to suppress this LO emission. Additionally, the
`radiated LO signal can then be reflected by buildings or moving objects and re-captured by the antenna. This
`effect, however, is not of significant importance compared to the aforementioned LO self-mixing and blocking
`signal self-mixing.
`
`The leakage of LO or RF signals to the opposite mixer port is not the only way in which unwanted DC can be
`produced. Any stage that exhibits even-order nonlinearity will also generate a DC output. This is covered in
`more detail later.
`
`Whether or not the DC product will desensitize the receiver depends on the type of system. Obviously it is
`preferable to AC couple at the mixer output to eliminate the DC. Some modulation schemes, such as frequency
`shift keying (FSK) used in paging applications, show little degradation if low frequency components of the
`spectrum are filtered out, as shown in Figure 9. However, other modulation schemes present a peak at DC, and
`capacitive AC coupling will lead to significant information loss, hence considerably degrading the bit error rate
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`(BER). In TDMA systems such as GSM, there is no significant low frequency spectral peak, but it still
`becomes impossible to AC couple. This is because of the conflicting requirements on an AC coupling capacitor
`in a TDMA system -- the capacitor must be large enough to avoid causing a wide notch at DC, but it must be
`small enough that all transients settle out upon power-up of the receiver (every frame) before data reception
`begins.
`
`In TDMA receivers that cannot be AC coupled, the idle timeslot (just before reception) can still be put to good
`use by storing the value of the offset in a capacitor and then subtracting it from the signal path during the burst.
`This is exactly the same method that is normally used to correct DC offsets occurring at the second mix of
`superheterodyne TDMA receivers, where this mix goes to baseband (in that case the only problem causing DC
`is LO self-mixing). In this method the value of DC produced by the receiver is obtained in a pre-measurement
`prior to the receive burst. It is important when using this method that the signal path prior to the mixer be
`opened during the DC pre-measurement to prevent any large blocking signals from affecting the result.
`Variable or wandering offsets are most often induced by blocking signals, which can appear at any time. These
`offsets cannot be corrected by the measurement-and-subtraction process, because the blocking signals may
`appear during the measurement and not during the burst, or vice-versa. For blocking-induced DC, the most
`effective measures are the elimination of self-mixing paths and the maximizing of linearity to prevent the DC
`to begin with. Failing these, there is still the possibility of DC correction after-the-fact in the digital signal
`processing occurring at baseband.
`
`Digital signal processing (DSP) techniques can be used to remove the DC offset in TDMA systems in a way
`that cannot be duplicated in the analog domain -- a full timeslot of the received signal can be buffered, the
`mean of which is determined and then removed from each data point of the signal. The resulting signal has
`zero mean. For systems such as GSM, an unwanted result of this is that any DC that is part of the signal will be
`lost, but the typical effect of this is minimal. Figure 10 illustrates the use of such a method for a typical GSM
`receiver. This technique can be further refined by tracking the mean over portions of the burst, allowing the
`detection of sudden interferers or blockers and cancelling their DC product only where it occurs. Careful layout
`can also improve isolation.
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`Nonlinearities
`
`As mentioned previously, another problem for the DCR is nonlinearity. Just as with the superheterodyne
`receiver, the DCR exhibits spurious responses. For the superheterodyne these occur at RF input frequencies
`where N(RF) ± M(LO) = IF, while for the DCR they occur where N(RF) M(LO) = 0. When a blocking signal's
`carrier falls on one of these spurious frequencies, the signal is translated to baseband with an attendant shift in
`its bandwidth, dependent on the spurious order.
`
`More importantly, however, large blocking signals also cause DC in the direct conversion receiver, whether on
`a spurious frequency or not. The DC is produced at the mixer output and amplified by the baseband stages. It is
`due primarily to second-order nonlinearity of the mixer, characterized by the second-order intercept point (IP2)
`and second-order intermodulation (IM2). It can be alleviated by extremely well-balanced circuit design.
`However, the mixer and LNA used to require a single-ended design because the antenna and a hypothetical
`preselect filter were usually single-ended.
`
`In most systems, the third-order intermodulation is of importance as it usually falls in-band, in the vicinity of
`the signals of interest, and is characterized by the third-order intercept point (IP3). In direct conversion,
`second-order nonlinearity becomes critical, as it produces baseband signals, which now appear as interfering
`signals in the down-converted desired signal. IM2 is measured by the IP2. IP2 is defined in the same manner as
`IP3, as shown in Figure 11. Either a two-tone, or single tone test can be performed, and the IP2 is defined by
`extrapolating the low frequency beat tone in the former or the DC component in the latter, until it intercepts the
`fundamental curve. To illustrate the case of a single tone test, the input signal is
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`x(t) = Acost(ωt).
`
`Assuming a nonlinearity modeled by a polynomial
`
`It can be seen that the DC component due to the second-order nonlinearity is growing with twice the slope of
`the fundamental on a logarithmic scale. At the intercept point,
`
`Due to the doubled slope of the second-order product,
`
`IIP2 = Pin + Δ with Δ = Pout IM2
`
`Noise
`
`Low frequency noise14 becomes a great concern in a DCR, as significant gain is allocated to baseband stages
`after the mixer. Weak signal levels of a few millivolts in baseband are still very vulnerable to noise. This
`requires stronger RF stage gain to alleviate the poor noise figure of baseband blocks, but of course this must be
`traded against the linearity problems, just described, that accompany higher RF gain.
`
`Flicker noise, or 1/f noise, is the major baseband noise contributor. Associated with a flow of direct current, it
`has a spectral response proportional to 1/f. In RF circuits, 1/f noise tends to be modulated onto the RF signal,
`and in the case of a mixer with baseband output, 1/f noise sees especially high conversion gain. In practice,
`flicker noise becomes an issue for MOS devices more than bipolar, and is modeled as a voltage source in series
`with the gate. 1/f noise complicates the use of MOS transistors for RF circuits, since the main method of
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`reducing it in MOS is to increase the transistor's size, which increases the device capacitance, adversely
`affecting RF gain. For this reason it is preferable to use bipolar transistors for DCR mixer designs. In the first
`baseband stages after the mixer, it becomes possible to use MOS devices, as the transistor-size trade-off is
`feasible at low frequencies.
`
`I/Q Mismatches
`
`Due to the high frequency of the LO, it is not possible to implement the IQ demodulator digitally. An analog
`IQ demodulator exhibits gain and phase imbalances between the two branches, as well as the introduction of
` and
` are the amplitude and
`DC offsets. Such imperfections distort the recovered constellation. Assuming
`phase mismatch, respectively, between the quadrature ports of the demodulator, and the complex signal
`incident upon it have in-phase and quadrature components I and Q, then
`
`Iout = (Icos(ωt) + Qsin(ωt)) 2cos(ωt)
`
`Qout = (Icos(ωt) + Qsin(ωt)) 2(1 +
`
` )sin(ωt +
`
` )
`
`Filtering out the high frequency terms yields
`
`Iout = I
`
`Qout = (1 +
`
` )(Isin + Qcos )
`
`Figure 12 shows how this affects a given constellation diagram. In DCR systems, however, the IQ matching is
`not as critical as in image-rejection architectures. Rather, it is only important insofar as the accuracy of the
`modulation is concerned.
`
`Analog and digital (DSP based) calibration and adaptation methods have been described so as to correct for
`these imbalances.15
`
`CONCLUSION
`
`The direct conversion receiver is an attractive yet challenging receiving technique. It has been successfully
`applied to devices such as pagers, mobile phones, PC and internet wireless connectivity cards, and satellite
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`receivers, etc. in a variety of process technologies and increasing integration levels. It is poised to appear in
`many more applications in the near future.
`
`ACKNOWLEDGMENTS
`
`The authors would like to thank Darioush Agahi and Morten Damgaard of Conexant Systems for their valuable
`input to this article. *
`
`References
`
`1. L. Lessing, "Man of High Fidelity: Edwin Howard Armstrong, a Biography," Bantam Books, New York,
`1969.
`
`2. F.M. Colebrook, "Homodyne," Wireless World and Radio Rev., 13, 1924, p. 774.
`
`3. D.G. Tucker, "The Synchrodyne," Electronic Engng, 19, March 1947, pp. 7576.
`
`4. D.G. Tucker, "The History of the Homodyne and the Synchrodyne," Journal of the British Institution of
`Radio Engineers, April 1954.
`
`5. A.A. Abidi, "Direct-conversion Radio Transceivers for Digital Communications," IEEE Journal of Solid-
`state Circuits, Vol. 30, No. 12, December 1995.
`
`6. B. Razavi, "Design Considerations for Direct-Conversion Receivers," IEEE Transactions on Circuits and
`Systems-II: Analog and Digital Signal Processing, Vol. 44, No. 6, June 1997.
`
`7. S.J. Franke, "ECE 353 Radio Communication Circuits," Department of Electrical and Computer
`Engineering, University of Illinois, Urbana, IL, 1994.
`
`8. B. Razavi, "RF Microelectronics," Prentice Hall, Upper Saddle River, NJ, 1998.
`
`9. J.C. Rudell, et al., "Recent Developments in High Integration Multi-standard CMOS Transceivers for
`Personal Communication Systems," International Symposium on Low Power Electronics and Design, 1998.
`
`10. J.C. Rudell, "Issues in RFIC Design," lecture notes, University of California Berkeley/National
`Technological University, 1997.
`
`11. R. Hartley, "Single-sideband Modulator," U.S. Patent No. 1666206, April 1928.
`
`12. D.K. Weaver, "A Third Method of Generation and Detection of Single Sideband Signals," Proceedings of
`the IRE, Vol. 44, December 1956, pp. 17031705.
`
`13. I.A.W. Vance, "Fully Integrated Radio Paging Receiver," IEE Proc., Vol. 129, No. 1, 1982, pp. 26.
`
`14. P.R. Gray and R.G. Meyer, "Analysis and Design of Analog Integrated Circuits," Third edition, John Wiley
`& Sons, New York, 1993.
`
`15. J.K. Cavers and M.W. Liao, "Adaptive Compensation for Imbalance and Offset Losses in Direct
`Conversion Transceivers," IEEE Transactions on Vehicular Technology, Vol. 42, November 1993, pp. 581588.
`
`Ashkan Mashhour received his Diplôme d'ingénieur from ENST Bretagne, France, and his MSc from
`University College London, UK, both in 1997. He then joined Nokia Networks, Camberley, UK, where he
`worked as a RF design engineer. His research involved the development of new RF/DSP technologies and
`linear transceiver architectures for future generation base stations. He is currently with Conexant Systems,
`Newport Beach, USA. He can be reached at: ashkan.mashhour@conexant.com.
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`William Domino received his BSEE degree from the University of Southern California in 1979, and his MEng
`degree from the California State Polytechnic University, Pomona, CA, in 1985. He joined the Collins Radio
`business of Rockwell International, Newport Beach, USA, in 1979, where he developed electromechanical IF
`filter models, design techniques and production processes. Currently a principal RF systems engineer with the
`Wireless Systems business of Conexant Systems, also in Newport Beach, USA, he has been involved in the
`design and development of integrated transceiver architectures for IS-136, Mobitex packet radio and GSM
`handsets. He can be reached at: william.domino@conexant.com.
`
`Norman Beamish earned his BEng and his PhD from University College Dublin, Ireland, in 1989 and 1994,
`respectively. His PhD research was in the field of DSP and digital communications with a particular interest in
`the equalisation of channels containing nonlinearities. He held a research engineering position at Teltec,
`Ireland, from 1994 to 1995. He is currently a principal engineer with Conexant Systems, Newport Beach, USA,
`where his primary interests are in the area of wireless baseband systems, particularly for GSM, 3G cellular
`systems and spread spectrum communications. He can be reached at: norman.beamish@conexant.com.
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