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`U.S. Patent
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`Sep. 10, 1996
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`Sheet 1 of 20
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`5,555,190
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`U.S. Patent
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`Sep. 10, 1996
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`Sheet 3 of 20
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`Sep. 10, 1996
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`Sheet 8 of 20
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`Sep. 10, 1996
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`Sep. 10, 1996
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`Sheet 13 of 20
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`Sheet 14 of 20
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`U.S. Patent
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`Sep. 10, 1996
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`Sheet 15 of 20
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`Sep. 10, 1996
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`Sheet 16 of 20
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`Sheet 17 of 20
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`DONE
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`DONE
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`FIG. 1 7
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`
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`A / D INTERRUPT
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`
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`1700
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`1702
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`READ A/D
`SAMPLE INTO
`CIRCULAR
`BUFFER
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`1 704
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` 8 NEW
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`SAMPLES READ
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`SINCE LAST
`CONVOLUTION
`7
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`
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` DETERMINE THE
`
`
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`CONVOLUTION OF SAMPLES
`AND STORE IN SECOND
`CIRCULAR BUFFER
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` 1706
`
`
`
`
`SINCE LAST
`CONVOLUTION
`
`?
`
`
`
` DETERMINE THE
`
`
`
`
`
`CONVOLUTION
`VALUES IN SECOND
`BUFFER AND STORE IN
`SAMPLE BUFFER
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` 1710
`
`
`DONE
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`18
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`18
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`U.S. Patent
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`Sep. 10, 1996
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`Sheet 18 of 20
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`5,555,190
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`FIG. 18
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`START
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`1800
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`1302
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`1304
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`INITIALIZE CIRCULAR BUFFERS FOR
`A/ D DECIMATION AND
`ENABLE A/ D INTERRUPTS
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`WAIT FOR DECIMATED
`SAMPLE AVAILABILITY
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`ACCUMULATE SIGNAL
`AND NOISE VALUES
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`APPLY NOTCH FILTER TO PRODUCE
`ENHANCED SAMPLE
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`UPDATE GOERTZEL
`FILTER TO ACCUMUIATE
`C0MpLEX NUMBEQ
`FOR PHASE
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`1 806
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`UPDATE FILTER PARAMETERS
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`
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`DETERMINE SNR VALUE
`FROM ACCUMULATED
`SIGNAL AND NOISE VALUES
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`
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`1814
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`RESET FILTER
`ADAPTATION
`COMPUTATIONS
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`1820
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`
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`DETERMINE At FOR PREVIOUS WINDOW AND
`APPLY UTILIZATION MEANS AND
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`DETERMINE GOERTZEL FILTER WEIGHTS
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`19
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`U.S. Patent
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`Sep. 10, 1996
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`Sheet 19 of 20
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`5,555,190
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`FIG. 19
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`1 806
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`DETERMINE
`UPDATED FORGETTING
`FACTOR
`
`DETERMINE
`UPDATED GAIN
`FACTOR
`
`-
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`DETERMINE
`UPDATED DEBIASING
`PARAMETER
`
`DETERMINE
`UPDATE COVARIANCE
`MATRIX
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`DETERMINE
`UPDATED NOTCH FILTER
`WEIGHTS
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`UPDATED
`WEIGHTS
`STABLE
`?
`
`APPLY UPDATED
`WEIGHTS TO
`NOTCH FILTERS
`
`20
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`20
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`U.S. Patent
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`Sep. 10, 1996
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`Sheet 20 of 20
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`5,555,190
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`1
`IVIETHOD AND APPARATUS FOR ADAPTIVE
`LINE ENHANCEMENT IN CORIOLIS MASS
`FLOW METER MEASUREMENT
`
`FIELD OF THE INVENTION
`
`The present invention relates to mass flow rate measure-
`ment and in particular to the use of digital signal processing
`adaptive filtration methods and apparatus in Coriolis mass
`flow meters.
`
`PROBLEM
`
`It is known to use Coriolis mass flowmeters to measure
`
`mass flow and other information for materials flowing
`through a conduit. Such flowmeters are disclosed in U.S.
`Pat. Nos. 4,109,524 of Aug. 29, 1978, U.S. Pat. No. 4,491,
`025 of Jan. 1, 1985, and Re. 31,450 of Feb. 11, 1982, all to
`J. E. Smith et al. These flowmeters have one or more flow
`
`tubes of straight or curved configuration. Each flow tube
`configuration in a Coriolis mass flowmeter has a set of
`natural vibration modes, which may be of a simple bending,
`torsional or coupled type. Each flow tube is driven to
`oscillate at resonance in one of these natural modes. Material
`flows into the flowmeter from a connected conduit on the
`inlet side of the flowmeter, is directed through the flow tube
`or tubes, and exits the flowmeter through the outlet side. The
`natural vibration modes of the vibrating, fluid filled system
`are defined in part by the combined mass of the flow tubes
`and the material within the flow tubes.
`
`When there is no flow through the flowmeter, all points
`along the flow tube oscillate about a pivot point with
`identical phase due to an applied driver force. As material
`begins to flow, Coriolis accelerations cause each point along
`the flow tube to have a different phase. The phase on the inlet
`side of the flow tube lags the driver, while the phase on the
`outlet side leads the driver. Sensors are placed on the flow
`tube to produce sinusoidal signals representative of the
`motion of the flow tube. The phase difference between two
`sensor signals is proportional
`to the mass flow rate of
`material through the flow tube.
`A complicating factor in this measurement is that the
`density of typical process material varies. Changes in den-
`sity cause the frequencies of the natural modes to vary. Since
`the flowmeter’s control system maintains resonance,
`the
`oscillation frequency varies in response to changes in den-
`sity. Mass flow rate in this situation is proportional to the
`ratio of phase difference and oscillation frequency.
`The above-mentioned U.S. Pat. No. Re. 31,450 to Smith
`discloses a Coriolis flowmeter that avoids the need for
`
`measuring both phase difference and oscillation frequency.
`Phase difference is determined by measuring the time delay
`between level crossings of the two sinusoidal sensor output
`signals of the flowmeter. When this method is used, the
`variations in the oscillation frequency cancel, and mass flow
`rate is proportional to the measured time delay. This mea-
`surement method is hereinafter referred to as a time delay or
`At measurement.
`Measurements in a Coriolis mass flowmeter must be made
`with great accuracy since it is often a requirement that the
`derived flow rate information have an accuracy of at least
`0.15% of reading. The signal processing circuitry which
`receives the sensor output signals measures this phase
`difference with precision and generates the desired charac-
`teristics of the flowing process material to the required
`accuracy of at least 0.15% of reading.
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`In order to achieve these accuracies, it is necessary that
`the signal processing circuitry operate with precision in
`measuring the phase shift of the two signals it receives from
`the flowmeter. Since the phase shift between the two output
`signals of the meter is the information used by the process-
`ing circuitry to derive the material characteristics,
`it
`is
`necessary that the processing circuitry not introduce any
`phase shift which would mask the phase shift information
`provided by the sensor output signals. In practice,
`it is
`necessary that this processing circuitry have an extremely
`low inherent phase shift so that the phase of each input
`signal is shifted by less than 0.001“ and, in some cases, less
`than a few parts per million. Phase accuracy of this magni-
`tude is required if the derived information regarding the
`process material is to have an accuracy of less than 0.15%.
`The frequencies of the Coriolis flowmeter output signals
`fall in the frequency range of many industrially generated
`noises. Also, the amplitude of the sensor output signals is
`often small and, in many cases, is not significantly above the
`amplitude of the noise signals. This limits the sensitivity of
`the flowmeter and makes the extraction of the useful infor-
`mation quite diflicult.
`There is not much a designer can do either to move the
`meter output signals frequency out of the noise band or to
`increase the amplitude of the output signals. Practical Corio-
`lis sensor and flowmeter design requires compromises that
`result in the generation of output signals having a less than
`optimum signal to noise ratio and dynamic range. This
`limitation determines the flowmeter characteristics and
`specifications including the minimum and maximum flow
`rates which may be reliably derived from the flowmeter’s
`output signals.
`The magnitude of the minimum time delay that can be
`measured between the two Coriolis flowmeter output signals
`at a given drive frequency is limited by various factors
`including the signal to noise ratio, the complexity of asso-
`ciated circuitry and hardware, and economic considerations
`that limit the cost and complexity of the associated circuitry
`and hardware. Also, in order to achieve a flowmeter that is
`economically attractive, the low limit of time delay mea-
`surement must be as low as possible. The processing cir-
`cuitry that receives the two output signals must be able to
`reliably measure the time delay between the two signals in
`order to provide a meter having the high sensitivity needed
`to measure the flowing characteristics of materials having a
`low density and mass such as, for example, gases.
`There are limitations regarding the extent to which con-
`ventional analog circuit design can, by itself, permit accurate
`time delay measurements under all possible operating con-
`ditions of a Coriolis flowmeter. These limitations are due to
`
`in any electronic equipment
`the inherent noise present
`including the imperfections of serni-conductor devices and
`noise generated by other circuit elements. These limitations
`are also due to ambient noise which similarly limits the
`measurement can be reduced to some extent by techniques
`such as shielding, guarding, grounding, etc. Another limi-
`tation is the signal to noise ratio of the sensor output signals
`themselves.
`
`Good analog circuit design can overcome some of the
`problems regarding noise in the electronic equipment as well
`as the ambient noise in the environment. However, an
`improvement in the signal to noise ratio of the output signals
`carurot be achieved without the use of analog filters. But
`analog filters alter the amplitude and phase of the signals to
`be processed. This is undesirable, since the time delay
`between the two signals is the base information used to
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`derive characteristics of the process fluid. The use of filters
`having unknown or varying amplitude and/or phase charac-
`teristics can unacceptably alter the phase difference between
`the two sensor output signals and preclude the derivation of
`accurate information of the flowing material.
`The flowmeter’ s drive signal is typically derived from one
`of the sensor output signals after it is conditioned, phase
`shifted and used to produce the sinusoidal drive voltage for
`the drive coil of the meter. This has the disadvantage that
`harmonics and noise components present in the sensor signal
`are amplified and applied to the drive coil to vibrate the flow
`tubes at their resonant frequency. However, an undesirable
`drive signal can also be generated by unwanted mechanical
`vibrations and electrical interferences that are fed back to the
`meter drive circuit and reinforced in a closed loop so that
`they create relatively high amplitude self-perpetuating dis-
`turbing signals which further degrade the precision and
`accuracy of the time delay measurement.
`There are several well known methods and circuit designs
`which seek to reduce the above problems. One such suc-
`cessful technique to reduce some of the above problems is
`described in U.S. Pat. No. 5,231,884 to M. Zolock and U.S.
`Pat. No. 5,228,327 to Bruck. These patents describe Coriolis
`flowmeter signal processing circuitry that uses three identi-
`cal charmels having precision integrators as filters. A first
`one of these channels is permanently connected to one
`sensor signal, say, for example,
`the left. The other two
`channels (second and third) are alternately connected, one at
`a time, to the right sensor signal. When one of these, say the
`second channel, is connected to the right sensor signal, the
`third channel is connected, along with the first channel, to
`the left sensor signal. The inherent phase delay between the
`. first and third channel is measured by comparing the time
`difference between the outputs of the two channels now both
`connected to the left signal. Once this characteristic delay is
`determined, the role of this third charmel and the second
`charmel connected to the right sensor signal is switched. In
`this new configuration,
`the second charmel undergoes a
`calibration of its delay characteristics while the third cali-
`brated channel is connected to the right sensor signal. The
`roles of second and third charmels are alternately switched
`by a control circuit approximately once every nrinute. Dur-
`ing this one-minute interval (about 30 to 60 seconds), aging,
`temperature, and other effects have no meaningful influence
`on the phase-shift of the filters and therefore their phase
`relationship is known and considered defined.
`The precisely calibrated integrators used by Zolock pro-
`vide a signal to noise ratio improvement amounting to about
`6 db/octave roll-ofi” in the amplitude transfer function of the
`integrator. Unfortunately, this 6 db/octave improvement is
`not enough under many circumstances in which Coriolis
`flowmeters are operated (such as light material or exces-
`sively noisy environments). The reason for this is that a
`single-pole filter, such as the Zolock integrator, has a rela-
`tively wide band width. As a result, noise signals generated
`by unwanted flow tube vibration modes, noisy environment,
`material flow noise, electromagnetic or radio frequency
`interference and disturbances are not removed to the extent
`necessary for high meter sensitivity required for precision.
`Depending on their frequency, their amplitude is reduced
`somewhat, but they can still interfere with the precision time
`delay measurement between the two sensor output signals
`when measuring low mass materials such as gases.
`There is another source for errors in the Zolock and Bruck
`systems. The integrator time delay measurements are made
`at three (3) certain well defined points of the sinusoidal
`sensor signals. The two sensor signals are ideal only when
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`their shape is the same and is symmetrical around their peak
`values. However, when the two magnetic circuits (sensors)
`that generate the sensor signals are not identical, the result-
`ing non-ideal wave forms contain different amounts of
`harmonics with possibly undefined phase conditions which
`can alter their shape and potentially change their symmetri-
`cal character. The result of such variations is such that when,
`during normal operations, a Zolock integrator is calibrated
`with one wave form and is subsequently used to measure
`another wave form,
`the difl“erence in wave forms may
`produce an undefined and unknown amount of error due to
`its harmonic content and its undefined and varying phase of
`its harmonics.
`
`Other analog circuit design techniques suffer from similar
`problems of complexity,
`insuflicient noise immunity, or
`insufficient harmonic rejection.
`There are techniques currently available, such as digital
`signal processing (hereinafter referred to as DSP) and asso-
`ciated digital filtering,
`to overcome the above-discussed
`problems and simultaneously improve the signal to noise
`ratio of the signals being processed. However, these alter-
`natives have been more complicated and expensive than
`conventional analog circuit designs. In addition, these prior
`DSP designs have shown only modest improvement over
`conventional analog circuit designs with respect to noise
`immunity and harmonic rejection. U.S. Pat. No. 4,934,196,
`issued Jun. 19, 1990 to Romano, teaches a DSP design for
`computing the phase difference, At, and correlated mass flow
`rate. Romano’s design alters the sampling frequency of an
`A/D converter in an attempt to maintain an integral number
`of sample times within each periodic cycle of the vibrating
`flow tubes. This need for variable frequency sampling
`complicates Romano’s DSP design. Although this DSP
`design is structurally quite distinct from prior discrete ana-
`log circuit designs, it has proven to provide only modest
`improvements over analog designs in measurement accu-
`racy because it provides significant improvement in filtration
`only at integer multiples of the fundamental frequency.
`However, many signal components result from mechanical
`vibration modes of the flow tubes whose resonant frequen-
`cies are not integer multiples of the fundamental frequency
`and are therefore poorly rejected by the prior DSP designs.
`Neither prior approach (analog nor prior DSP) effectively
`rejects non-harmonic or broadband noise. From the above
`discussion, it can be seen that there is a need for an improved
`method and apparatus for measuring mass flow rate in a
`Coriolis mass flow meter.
`
`SOLUTION
`
`The present invention solves the above identified prob-
`lems and achieves an advance in the art by applying digital
`filtering and digital signal processing (DSP) methods and
`apparatus to improve the accuracy of mass fiow measure-
`ments in a Coriolis mass flow meter. The present invention
`comprises a DSP design which includes adaptive notch
`filters to improve the accuracy of frequency and phase
`measurements used in the computation of mass flow rate.
`The use of adaptive notch filtration in the present invention
`is one application of the technology commonly referred to as
`Adaptive Line Enhancement (ALE).
`In the present invention, the signal from each vibrating
`flow tube sensor is sampled, digitized, and then processed by
`a digital adaptive notch filter which passes all noise signals
`outside a narrow frequency band (a notch) around the
`fundamental frequency. This digitized filtered signal is then
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`subtracted from the original digitized signal to produce an
`enhanced signal representing the sensor output signal wave-
`form at the fundamental frequency with virtually all noise
`signals eliminated. This method and apparatus eliminates
`harmonic as well as non-harmonic noise signals. Initially the
`width of the notch filter’s “notch” is wide and is adapted
`over time to narrow as it converges on the fundamental
`frequency. Adaptation algorithms rapidly adapt the notch
`frequency of the adaptive filter to track changes over time in
`the fundamental frequency of the vibrating flow tubes.
`The DSP design of the present invention uses a fixed
`sampling frequency as distinct from Romano’s variable
`frequency design. This fixed sampling frequency approach
`permits rapid convergence of the adaptive notch filters on
`the fundamental frequency of the vibrating flow tubes and
`simplifies the total circuit design. The fixed sampling rate
`eliminates the need exhibited in Romano to provide addi-
`tional circuitry to vary the sampling rate. The present design
`performs computational adjustments to compensate for
`spectral leakage between the fixed sampling frequency and
`the variable fundamental frequency of the vibrating flow
`tubes. Despite this added computational complexity,
`the
`present invention is simpler than prior designs exemplified
`by Romano and provides better noise immunity due to the
`use of adaptive notch filtration.
`The present invention provides superior noise immunity
`and harmonic rejection as compared to all known designs
`and simplifies aspects of the DSP design disclosed by
`Romano. This permits improved accuracy of the flow rate
`measurements even in particularly noisy environments as
`well as applications with low density flow materials (such as
`gas).
`Since the flow tubes vibrate at the same fundamental
`
`frequency, adaptation of the notch filters is determined by
`samples from only one of the two notch filters. The adap-
`tation weights so determined are applied to both notch
`filters. Heuristics applied to the computations by the present
`invention prevent the notch filters from diverging from the
`fundamental frequency due to instability in the computa-
`tions. Other heuristics restart convergence computations for
`the adaptation when the signal to noise ratio measured by the
`notch filter is too small. A small signal
`to noise ratio
`indicates that the adaptive notch filter is not converged on
`the fundamental frequency. This may be due to a shift in the
`fundamental frequency of the vibrating flow tubes.
`In a first embodiment of the present invention, the output
`signal from each vibrating flow tube sensor is sampled at a
`fixed frequency by a corresponding A/D converter. The
`sampled value generated by each A/D converter is then
`applied to a corresponding decimation filter to reduce com-
`putational complexity by reducing the number of samples
`used in subsequent computations. The decimation filters also
`provide a degree of anti-aliasing filtration to smooth the
`sampled analog signals. The decimated signals are then each
`applied to a corresponding adaptive notch filter to further
`enhance the signal from each sensor. The enhanced output
`signal from each sensor, after being filtered of most noise
`and harmonics, is then applied to a corresponding phase
`computation element
`to determine the phase difference
`between the two enhanced signals. The output of each phase
`computation element is applied to a computation element to
`determine the time difference between the enhanced sensor
`signals and hence the proportional mass flow rate.
`In a second embodiment of the methods of the present
`invention, four adaptive notch filters are utilized,
`two in
`series on each of the left and right channel signals. The two
`
`filters on each of the left and right channels are “cascaded”
`in that the first filter utilizes a low-Q (wide notch) filter to
`supply limited signal enhancement but the ability to rapidly
`converge on changes in the fundamental frequency of the
`vibrating flow tubes. The signal output from the first cas-
`caded notch filter is then applied to a second cascaded notch
`filter. The second notch filter utilizes a high-Q (narrow
`notch) filter to provide superior noise and harmonic rejection
`over that of previous solutions or over that of the first
`embodiment described above. Despite the narrow notch
`(high-Q) of the second notch filter, it can still rapidly adapt
`to changes in the fundamental frequency of the vibrating
`flow tubes due to the limited enhancement (filtration) per-
`formed by the first notch filter. The reduced noise and
`harmonic levels in the signal applied to the second notch
`filter allow it to also rapidly converge on changes in the
`fundamental frequency of the vibrating flow tubes.
`An additional notch filter (fifth filter) having a notch
`shape even wider than that of the first cascaded notch filter
`is used to provide an estimate of the fundamental frequency
`of the vibrating flow tubes. This estimate is used by weight
`adaptation computations to set the frequency parameter of
`the first cascaded notch filters for both the left and right
`channels. The output from the second cascaded notch filters
`is used by weight adaptation computations to adjust the
`frequency parameter of the second cascaded notch filters.
`This combination of two (or more) cascaded adaptive
`notch filters to enhance the output signal from each sensor
`further enhances both the rejection characteristics of the
`filtration and the speed with which the adaptive filters
`converge on changes in the fundamental frequency of the
`vibrating flow tubes.
`The term “adaptive notch filter” as used herein refers
`broadly to a filter with variable parameters. This definition
`contrasts with a more widely accepted definition which
`combines a variable parameter filter with a mechanism for
`automatically tuning the parameters of the filter based on the
`filter’s own inputs and outputs. As used herein, the adapta-
`tion of some notch filters is computed based on the operation
`of other filters rather than each filters own inputs and
`outputs. In other words, some notch filters in the present
`invention are slaved to the operation of other notch filter
`computations. For this reason, the detailed discussions of the
`filters and the adaptation mechanisms are separated. One
`adaptation computation may adjust the parameters for mul-
`tiple notch filters based on inputs from a single filter.
`The above and other aspects of the present invention will
`become apparent from the following description and the
`attached drawing.
`
`BRIEF DESCRIPTION OF THE DRAWING
`
`FIG. 1 shows a typical Coriolis mass flow meter attached
`to meter electronics which embody the apparatus and meth-
`ods of the present invention;
`FIG. 2 shows a block diagram of the computational
`elements within the meter electronics which determine mass
`flow rate through the flow meter in accordance with the
`present invention;
`FIG. 3 shows additional detail of a first embodiment of the
`present invention shown in FIG. 2 wherein a single adaptive
`notch filter is used in conjunction with each sensor signal;
`FIGS. 4-12 show additional detail of the computational
`elements of the first embodiment of the present invention
`shown in FIG. 3;
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`FIG. 13 shows additional detail of a second embodiment
`of the present invention shown in FIG. 2 wherein two
`cascaded adaptive notch filters are used in conjunction with
`each sensor signal;
`FIGS. 14-16 show additional detail of the computational
`elements of the second embodiment of the present invention
`shown in FIG. 13; .
`
`FIG. 17 is a flowchart of a software implementation of the
`first embodiment of the present invention and depicts inter-
`rupt processing for servicing of an AID converter and
`associated decimation of the samples;
`FIG. 18 is a flowchart of a software implementation of the
`first embodiment of the present invention and depicts pro-
`cessing of decimated samples for purposes of filtering and
`determination of At phase difference;
`FIG. 19 is a flowchart depicting additional detail of an
`element of FIG. 18 which determines updated filter param-
`eters after each decimated sample is processed; and
`FIG. 20 is a block diagram of digital signal processing
`electronics suitable to perform the software methods of the
`present invention.
`
`DETAILED DESCRIPTION OF THE
`PREFERRED EMBODIMENT
`
`A typical Coriolis mass flowmeter 10 is illustrated in FIG.
`1 as having two cantilever mounted flow tubes 12, 14 aflixed
`to a manifold body 30 so as to have substantially identical
`spring constants and moments of inertia about their respec-
`tive out-of-phase bending axes W—W and W’—W‘.
`A drive coil and magnet 20 are mounted at a midpoint
`region between the top portion 130 and 130' of flow tubes
`12, 14 to oscillate flow tubes 12, 14 out of phase about axes
`W—W and W’—W‘. Left sensor 16 and right sensor 18 are
`mounted near the respective ends of the top portions of flow
`tubes 12, 14 to sense the relative movement of flow tubes 12,
`14. This sensing may be done in many ways including by
`measuring the movement of the top ends of the flow tubes
`12, 14 through their zero crossings or some other pre-defined
`point. Flow tubes 12 and 14 have left side legs 131 and 131'
`and right side legs 134 and 134'. The side legs converge
`downwardly toward each other and are aflixed to surfaces
`120 and 120' of manifold elements 121 and 121'. Brace bars
`140R and 140L are brazed to the legs of flow tubes 12, 14
`and serve to define the axes W—W and W'——W' about which
`the flow tubes oscillate out of phase when driver 20 is
`energized over path 156. The position of axes W—W and
`W—W‘ is determined by the placement of brace bars 140R
`and 140L on flow tube side legs 131, 131' and 134, 134’.
`Temperature detector 22 is mounted on side leg 131 of
`flow tube 14 to measure the flow tube’s temperature and the
`approximate temperature of the material flowing therein.
`This temperature information is used to determine changes
`in the spring constant of the flow tubes. Driver 20, sensors
`16 and 18 and temperature detector 22 are connected to mass
`flow instrumentation 24 by paths 156, 157, 158 and 159,
`respectively. Mass flow instrumentation 24 includes at least
`one microprocessor which processes the signals received
`from sensors 16, 18, and 22 to determine the mass flow rate
`of the material flowing through flowmeter 10 as well as other
`measurements, such as material density and temperature.
`Mass flow instrumentation 24 also applies a drive signal
`over path 156 to driver 20 to oscillate tubes 12 and 14
`out-of-phase about axes W—W and W‘-—W'.
`Manifold body 30 is formed of casting 150, 150'. Casting
`elements 150, 150' are attachable to a supply conduit and
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`exit conduit (not shown), by flanges 103, 103'. Manifold
`body 30 diverts the material flow from the supply conduit
`into flow tubes 12, 14 and then back into an exit conduit.
`When manifold flanges 103 and 103‘ are connected via inlet
`end 104 and outlet end 104' to a conduit system (not shown),
`carrying the process material to be measured, the material
`enters manifold body 30 and manifold element 110 through
`inlet orifice 101 in flange 103 and is connected by a channel
`(not shown) having a gradually changing cross-section in
`casting element 150 to flow tubes 12, 14. The material is
`divided and routed by manifold element 121 to the left legs
`131 and 131' of flow tubes 14 and 12, respectively. The
`material then flows through the top tubes elements 130, 130'
`and through the right side legs 134 and 134‘ and is recom-
`bined into a single stream within flow tube manifold element
`121'. The fluid is thereafter routed to a channel (not shown)
`in exit casting element 150' and then to exit manifold
`element 110'. Exit end 104' is connected by flange 103'
`having bolt holes 102' to the conduit system (not shown).
`The material exits through outlet orifice 101' to return to the
`flow in the conduit system (not shown).
`Mass flow instrumentation 24 analyzes signals received
`on paths 157, 158, and 159 and generates standard output
`signals on path 155 to indicate mass flow rates utilized by a
`control system or operator for monitoring and control of the
`mass flow rate through the associated conduit system (not
`shown).
`
`OVERVIEW
`
`The present invention comprises digital signal processing
`methods operable within a digital signal processor (DSP)
`chip to perform the computational functions within mass
`flow instrumentation 24. Discrete samples are taken of the
`analog signals generated as output from each of the flow
`tube sensors. The discrete samples from the left and right
`sensors are digitized by use of standard analog to digital
`conversion (A/D) devices. Once digitized, further process-
`ing of the samples is performed by digital signal processing
`methods within the DSP chip. The processing of the digi-
`tized signal samples is expressed herein in two forms. In one
`form of expression, the DSP software flowcharts and equa-
`tions used for the various filtering and processing functions
`are presented. To aid in the explanation of the methods of the
`present invention, a second form of expression is utilized
`which depicts the computation of the various equations as
`pseudo-circuits (e.g. block diagrams representing summing
`junctions, multiplication junctions, delay circuits, registers,
`multiplexors, etc.). Certain more complex mathematical
`operations are left as high level elements in the pseudo-
`circuit diagrams and are typically referred to herein as
`“computational elements”. The two forms of explanation of
`the present invention are intended as equivalent descrip-
`tions, either of which fully specifies the methods and func-
`tion of the present invention.
`
`OVERVIEW- PSEUDO CIRCUITS
`
`FIG. 2 depicts the general structure of, and associated
`flow of information in, the flow meter electronics of the
`present
`invention. The meter electronics of the present
`invention is comprised of two essentially identical “chan-
`nels”: a first channel for processing the left flow tube sensor
`output signal and a second channel for processing the right
`flow tube sensor output signal. The two “channels” are
`identical except with respect to the weight adaptation of the
`notch filters as discussed below.
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`The description presented belowis discussed in terms of
`a typical Coriolis flowmeter application in which the fun-
`damental frequency of the vibrating flow tubes is approxi-
`mately lOO Hz. It will be readily recognized that
`the
`apparatus and methods of the present invention may be
`applied to any common flowmeter fundamental vibrating
`frequency.
`Many of the computational elements discussed below
`operate synchronously with clock signals associated with
`various samplings of